Applied Science, Faculty of
Engineering, School of (Okanagan)
DSpace
UBCO
Rafi, Md. Abdur
Master of Applied Science - MASc
University of British Columbia
This thesis presents an innovative and practical approach to developing planar resonant-based spectroscopy in microwave sensor technology. Resonant based microwave sensors have drawn attention for decades now in terms of sensing and measuring materials dielectric properties because of demonstrating a higher accuracy, precision, and sensitive measurement compared to non-resonant based sensors. This method has been mostly used on binary solvents due to their ability to display a highly delicate response to the materials under test. This thesis presents a comprehensive analysis of the sensor design and fabrication, sensor characterization, and testing samples to achieve the materials' frequency selective response. Microwave resonator sensors are limited in their current form to a single fixed resonant frequency, leading to issues in selectivity among materials or mixtures with similar dielectric properties. The first study presents a successful effort of achieving a frequency-tunable resonator circuit by utilizing the liquid metal and 3D printed microchannel with a modified split ring resonator. The resonator utilizes liquid metal in a microfluidic channel to change the resonant frequency of the resonator over a wide, continuous spectrum by changing the effective capacitance of the sensor. In the second study, highly sensitive interdigitated capacitive traces have been added with the designed sensor structure for material sensing. The designed and fabricated sensor demonstrated an operating resonant frequency that could be tuned between 2.1 and 2.8 GHz for both solid and liquid sensing. The sensitivity of the designed sensor structure is investigated by sensing standard solid samples, and sensitivity variation was noticed due to the increased reactive loading introduced by the liquid metal. Experimental validation of the sensor and standard liquid mixtures confirmed that the frequency tunability enables similar dielectric materials to be distinctly identified using multiple frequency spectra. Specific attention was made towards materials with identical dielectric properties at specific frequencies, simulated with the Debye model, and matched experimentally with dielectric probe validation. The device enables multi-phase material detection with the ability to investigate dielectric properties over a frequency spectrum rather than a single point.
Tunable Microwave Resonator UsingLiquid Metal And 3D Printed FluidicChannel For The Development OfMicrowave SpectroscopybyMd. Abdur RafiB.Sc. Engg., Islamic University of Technology (IUT), 2016A THESIS SUBMITTED IN PARTIAL FULFILLMENT OFTHE REQUIREMENTS FOR THE DEGREE OFMASTER OF APPLIED SCIENCEinTHE COLLEGE OF GRADUATE STUDIES(Electrical Engineering)THE UNIVERSITY OF BRITISH COLUMBIA(Okanagan)January 2021© Md. Abdur Rafi, 2021The following individuals certify that they have read, and recommendto the College of Graduate Studies for acceptance, a thesis/dissertation mi-croelectronics:Tunable Microwave Resonator using Liquid Metal and 3Dprinted Fluidic Channel for the Development of MicrowaveSpectroscopysubmitted by Md. Abdur Rafi in partial fulfilment of the requirements ofthe degree of Master of Applied ScienceDr. Mohammad Hossein Zarifi, School of EngineeringSupervisorDr. Ian Foulds, School of EngineeringSupervisory Committee MemberDr. Kevin Golovin, School of EngineeringSupervisory Committee MemberDr. Sina Kheirkhah, School of EngineeringUniversity ExamineriiAbstractThis thesis presents an innovative and practical approach to developingplanar resonant-based spectroscopy in microwave sensor technology. Reso-nant based microwave sensors have drawn attention for decades now in termsof sensing and measuring materials dielectric properties because of demon-strating a higher accuracy, precision, and sensitive measurement comparedto non-resonant based sensors. This method has been mostly used on bi-nary solvents due to their ability to display a highly delicate response to thematerials under test. This thesis presents a comprehensive analysis of thesensor design and fabrication, sensor characterization, and testing samplesto achieve the materials’ frequency selective response.Microwave resonator sensors are limited in their current form to a singlefixed resonant frequency, leading to issues in selectivity among materialsor mixtures with similar dielectric properties. The first study presents asuccessful effort of achieving a frequency-tunable resonator circuit by uti-lizing the liquid metal and 3D printed microchannel with a modified splitring resonator. The resonator utilizes liquid metal in a microfluidic channelto change the resonant frequency of the resonator over a wide, continuousspectrum by changing the effective capacitance of the sensor. In the secondstudy, highly sensitive interdigitated capacitive traces have been added withthe designed sensor structure for material sensing. The designed and fab-ricated sensor demonstrated an operating resonant frequency that could betuned between 2.1 and 2.8 GHz for both solid and liquid sensing. The sen-sitivity of the designed sensor structure is investigated by sensing standardsolid samples, and sensitivity variation was noticed due to the increasedreactive loading introduced by the liquid metal.Experimental validation of the sensor and standard liquid mixtures con-firmed that the frequency tunability enables similar dielectric materials tobe distinctly identified using multiple frequency spectra. Specific attentionwas made towards materials with identical dielectric properties at specificfrequencies, simulated with the Debye model, and matched experimentallywith dielectric probe validation. The device enables multi-phase material de-tection with the ability to investigate dielectric properties over a frequencyiiiAbstractspectrum rather than a single point.ivLay SummaryThis thesis demonstrates a successful approach to detecting materialsin different phases using a resonant based spectroscopy method. The de-signed structures were employed in multiple independent studies in order toachieve material identification in different resonant frequencies. The initialstudy discusses the frequency tuning capability of the designed structureby utilizing the potential application of liquid metal in a 3D printed mi-crofluidic channel. The second study shows a modification in the designedstructure and includes performance characterization of the designed sensorby sensing standard solid samples in different tunable resonant frequencies.In the third study, coaxial probe measurements indicate the liquid samplesthat exhibit similar dielectric properties at a frequency, which defines theinability to sense those samples using a single fixed resonant frequency. Inthe final study, the designed tunable resonant structure was successfullyutilized to identify samples of identical dielectric properties in microwavesensor technology.vPrefaceThis work has been performed under the guidance of Dr. MohammadHossein Zarifi at the School of Engineering at The University of BritishColumbia. The fourth chapter of this thesis contains materials that havebeen published in the following journal:• Wideband Tunable Modified Split Ring Resonator Structure UsingLiquid Metal and 3-D Printing, IEEE Microwave and Wireless Com-ponents Letters, vol. 30, no. 5, pp. 1–4, 2020.The chapter five of this thesis has also been presented in the followingjournal for review:• Continuously Tunable Resonator Development utilizing Liquid Metalto generate Spectroscopic Data for Multi-phase Material Identification,IEEE Transactions on Microwave Theory and Techniques.viTable of ContentsAbstract . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . iiiLay Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . vPreface . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . viTable of Contents . . . . . . . . . . . . . . . . . . . . . . . . . . . viiList of Tables . . . . . . . . . . . . . . . . . . . . . . . . . . . . . xList of Figures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . xiAcknowledgements . . . . . . . . . . . . . . . . . . . . . . . . . .xviiDedication . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .xviiiChapter 1: Introduction . . . . . . . . . . . . . . . . . . . . . . . 11.1 Motivation . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11.2 Objective . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31.3 Thesis Organization . . . . . . . . . . . . . . . . . . . . . . . 3Chapter 2: Introduction to microwave technology, planar trans-mission lines and measurement techniques . . . . 52.1 Introduction to microwave technology . . . . . . . . . . . . . 52.2 Planar transmission lines . . . . . . . . . . . . . . . . . . . . 72.2.1 Microstrip . . . . . . . . . . . . . . . . . . . . . . . . . 72.2.2 Coplanar waveguide . . . . . . . . . . . . . . . . . . . 82.2.3 Coplanar strip and differential line . . . . . . . . . . . 82.2.4 Stripline . . . . . . . . . . . . . . . . . . . . . . . . . . 92.2.5 Embedded differential line . . . . . . . . . . . . . . . . 92.3 Microwave technology in material detection . . . . . . . . . . 9viiTABLE OF CONTENTS2.4 Different measurement techniques in microwave technology . 132.4.1 Transmission/reflection (TR) line method . . . . . . . 132.4.2 Open-ended coaxial probe method . . . . . . . . . . . 142.4.3 Free space method . . . . . . . . . . . . . . . . . . . . 162.4.4 Resonant method . . . . . . . . . . . . . . . . . . . . . 17Chapter 3: Planar microstrip resonator, tuning mechanismsand microwave spectroscopy . . . . . . . . . . . . . 203.1 Planar Microstrip Resonator . . . . . . . . . . . . . . . . . . . 203.1.1 Theory of operation . . . . . . . . . . . . . . . . . . . 203.1.2 Resonant frequency and quality factor . . . . . . . . . 243.1.3 Split ring resonator . . . . . . . . . . . . . . . . . . . . 273.2 Tuning Mechanisms . . . . . . . . . . . . . . . . . . . . . . . 303.2.1 External devices . . . . . . . . . . . . . . . . . . . . . 303.2.2 Microfluidic-based tuning . . . . . . . . . . . . . . . . 323.3 Review of Microfluidic Applications . . . . . . . . . . . . . . . 333.3.1 Challenges and limitations of microfluidic based tuning 353.3.2 Microfluidic Channel . . . . . . . . . . . . . . . . . . . 383.4 Microwave Spectroscopy . . . . . . . . . . . . . . . . . . . . . 40Chapter 4: Adjustable Microwave Resonator . . . . . . . . . . 444.1 Design and Simulation . . . . . . . . . . . . . . . . . . . . . . 444.1.1 Design methodology of the resonator structure . . . . 454.1.2 Design methodology of the fluidic channel . . . . . . . 494.1.3 Structural simulation . . . . . . . . . . . . . . . . . . . 514.2 Fabrication and Experimental Setup . . . . . . . . . . . . . . 564.2.1 Fabrication of highly coupled resonator structure . . . 564.2.2 3D printed fluidic channel . . . . . . . . . . . . . . . . 574.2.3 Experimental setup . . . . . . . . . . . . . . . . . . . . 604.3 RF Characterization . . . . . . . . . . . . . . . . . . . . . . . 604.3.1 Scattering (s) parameter measurement . . . . . . . . . 614.3.2 Equivalent circuit modeling . . . . . . . . . . . . . . . 644.3.3 Results comparison . . . . . . . . . . . . . . . . . . . . 654.4 Chapter summary . . . . . . . . . . . . . . . . . . . . . . . . 67Chapter 5: Resonant Based Spectroscopy for multi-phase ma-terial detection . . . . . . . . . . . . . . . . . . . . . 685.1 Problem statement . . . . . . . . . . . . . . . . . . . . . . . . 685.2 Operating principle of the designed resonator sensor . . . . . 705.3 Design methodology and simulation . . . . . . . . . . . . . . 71viiiTABLE OF CONTENTS5.3.1 Structure design and tunable operation . . . . . . . . 715.3.2 Sensitivity analysis of the designed sensor . . . . . . . 755.3.3 Sensor performance analysis . . . . . . . . . . . . . . . 795.4 Experimental setup and measurement discussion . . . . . . . 805.4.1 Characterizing the designed resonator sensor . . . . . 805.4.2 Standard probe measurement . . . . . . . . . . . . . . 855.4.3 Validation of the designed resonator sensor . . . . . . 905.5 Chapter summary . . . . . . . . . . . . . . . . . . . . . . . . 95Chapter 6: Conclusion and Future Investigations . . . . . . . 976.1 Thesis outcome and conclusion . . . . . . . . . . . . . . . . . 976.2 Future work . . . . . . . . . . . . . . . . . . . . . . . . . . . . 98Bibliography . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100ixList of TablesTable 3.1 Comparison of different switching techniques . . . . . 33Table 3.2 Comparison of physical properties of Copper, Galin-stan, EGaIn and Mercury © 2015 IEEE . . . . . . . . 36Table 4.1 Design parameters of the designed modified resonatorstructure . . . . . . . . . . . . . . . . . . . . . . . . . . 48Table 4.2 Defining the values of the parameters of the equivalentcircuit model based on the simulation results. . . . . 65Table 5.1 Defining the values of the parameters of the designedsensor . . . . . . . . . . . . . . . . . . . . . . . . . . . 72Table 5.2 Measured effective permittivity of the sample liquidsolutions . . . . . . . . . . . . . . . . . . . . . . . . . . 87Table 5.3 Measured frequency difference due to sample liquidspresence in microfluidic channel in different resonantfrequencies. . . . . . . . . . . . . . . . . . . . . . . . . 94xList of FiguresFigure 2.1 Electromagnetic frequency spectrum and associatedwavelengths. . . . . . . . . . . . . . . . . . . . . . . . 6Figure 2.2 Cross sections of different planar transmission linestructures where (a) microstrip, (b) coplanar waveg-uide, (c) coplanar strip and differential line, (d) striplineand (e) embedded differential line. . . . . . . . . . . . 8Figure 2.3 A dielectric with polar molecule: (a) in the absenceof an external electric field Eext , (b) in the presenceof an external electric field Eext and (c) the inducedelectric field Eind inside the dielectric produced bythe induced surface charge Qi. . . . . . . . . . . . . . 10Figure 2.4 Complex dielectric permittivity spectrum over a rangeof frequencies where ′ and ′′ denote the real andimaginary part, respectively. . . . . . . . . . . . . . . 12Figure 2.5 Measurement using TR method with a waveguide . . 13Figure 2.6 Experimental setup of an open-ended coaxial probemethod. . . . . . . . . . . . . . . . . . . . . . . . . . . 15Figure 2.7 Measurement of sample under test using free spacemethod. . . . . . . . . . . . . . . . . . . . . . . . . . . 16Figure 2.8 Measurement of a thin rod using resonant method. . 18Figure 3.1 Lumped element representation of an infinitesimalpart ∆Z of a microstrip transmission line where (a)the actual microstrip and (b) the modelled microstrip. 21Figure 3.2 Electric (E-field) and magnetic (H-field) field distri-bution of a microstrip line. . . . . . . . . . . . . . . . 22Figure 3.3 Lumped element model of a microstrip resonant circuit. 23Figure 3.4 Quality Factor of a resonant circuit. . . . . . . . . . . 26Figure 3.5 A simple split ring resonator structure coupled withfeedline. . . . . . . . . . . . . . . . . . . . . . . . . . . 27xiLIST OF FIGURESFigure 3.6 (a) Resonator sensor structure implemented in HFSSfor the purpose of early detection breaches in pipelinecoatings and (b) open split ring resonator structure(OSRR) . . . . . . . . . . . . . . . . . . . . . . . . . . 28Figure 3.7 Some geometries of split ring resonator (SRR). (a)Double split ring resonator, (b) Double split squareresonator, (c) Singly split ring resonator, (d) Two-turn spiral (circular) resonator, (e) Two-turn spiral(rectangular) resonator and (f) Modified ring resonator. 29Figure 3.8 Tuning by the external devices. (a) SRR with capac-itor loaded, (b) Varactor loaded S-shaped resonatorand (c) RF-MEMS loaded microstrip patch antenna. 31Figure 3.9 Microfluidic based tuning. (a) Implemented proto-type of the microfluidic waveguide switch, (b) Recon-figurable CPW folded slot antenna. . . . . . . . . . . 34Figure 3.10 A liquid metal plug sandwiched between the carrierfluid inside the microfluidic channel. . . . . . . . . . . 36Figure 3.11 (a) A mm syringe and (b) 11 Pico Plus Elite pro-grammable syringe to control the liquid flow. . . . . . 37Figure 3.12 (a) EGaIn plug inside a PDMS and (b) 3D printedstructure utilized as a microfluidic channel. . . . . . . 39Figure 3.13 Dielectric spectra of soybean oil at 25°C temperatureover a broadband where (a) real part and (b) imagi-nary part. . . . . . . . . . . . . . . . . . . . . . . . . 40Figure 3.14 CPW folded slot antenna integrated with liquid metalto achieve three different frequency bands of 2.4, 3.5and 5.8 GHz. . . . . . . . . . . . . . . . . . . . . . . . 41Figure 4.1 Pictorial depiction of the device development process. 44Figure 4.2 Simple SRR integrated with (a) no microfluidic chan-nel (MC), (b) one MC, (c) two MC, (d) three MCand (e) four MC filled with liquid metal, and (f) sim-ulation result of different configurations of MC filledliquid metal. . . . . . . . . . . . . . . . . . . . . . . . 46Figure 4.3 Frequency spectra of a simple SRR with varying liq-uid metal inside the fluidic channel. . . . . . . . . . . 47Figure 4.4 Designed highly coupled modified SRR structure withdimensions. . . . . . . . . . . . . . . . . . . . . . . . . 48xiiLIST OF FIGURESFigure 4.5 (a) The cross-section of the designed fluidic channel,(b) channel height (h=) 1.5 mm, width (w=) 2.0 mmand channel separation (d1=) 50 µm and (c) channellength (L=) 3 cm. . . . . . . . . . . . . . . . . . . . . 50Figure 4.6 Resonant frequency shift at different widths when liq-uid metal (LM) length is 32mm, 26mm and 20mm. . 51Figure 4.7 Layout of the designed resonator integrated with flu-idic channel. . . . . . . . . . . . . . . . . . . . . . . . 52Figure 4.8 Cross-sectional sideview of the designed resonator in-tegrated with fluidic channel. . . . . . . . . . . . . . . 53Figure 4.9 Transmission spectra (S21) of the designed resonatorstructure for (a) different gap sizes, g and (b) differentcoupling distances, d. . . . . . . . . . . . . . . . . . . 53Figure 4.10 Transmission spectra (S21) of the designed resonatorstructure. . . . . . . . . . . . . . . . . . . . . . . . . . 54Figure 4.11 EM field distribution plot of the designed resonatorstructure from HFSS at a resonance of 3.52 GHz. . . 55Figure 4.12 Fabricated resonator structure on Rogers RT/duroid5880 high-frequency substrate and SMA connectorsare soldered with the feedlines. . . . . . . . . . . . . . 57Figure 4.13 (a) Fabricated PDMS microchannel and (b) the de-signed resonator structure integrated with the PDMSmicrochannel. . . . . . . . . . . . . . . . . . . . . . . 58Figure 4.14 Fabricated resonator integrated with 3D printed flu-idic channel. . . . . . . . . . . . . . . . . . . . . . . . 58Figure 4.15 Surface profile of the 3D printed fluidic channel alongthe length A to C. . . . . . . . . . . . . . . . . . . . . 59Figure 4.16 Surface profile of the embedded channel inside the 3Dprinted structure. . . . . . . . . . . . . . . . . . . . . 60Figure 4.17 Experimental setup used to measure the S21 responseof the tunable resonator. The fabricated resonatorfilled with liquid metal and microchannel implemen-tation is shown in the top-left inset. . . . . . . . . . . 61Figure 4.18 Frequency spectra from structure simulation for dif-ferent liquid metal amounts. Few of the instances ofliquid metal inside the channel is shown on the leftside. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62xiiiLIST OF FIGURESFigure 4.19 Resonant S21 spectra recorded by the VNA as thevolume of metal inside the channel is varied. Few ofthe instances of liquid metal inside the channel areshown on the left side. . . . . . . . . . . . . . . . . . 63Figure 4.20 Equivalent circuit model overlaying the substrate, res-onator and liquid metal. . . . . . . . . . . . . . . . . 64Figure 4.21 Comparison between the simulated (HFSS and ADS)and measured S21 frequency spectra for the resonatorwith the fluidic channel in a (a) empty state and (b)filled state. . . . . . . . . . . . . . . . . . . . . . . . . 66Figure 4.22 Resonant frequencies recorded for different metal lengthswith selected images included. . . . . . . . . . . . . . 66Figure 5.1 The calculated dielectric complex permittivity of ma-terial A and material B at 23.7°C modelled by Debyemethod. The parameters for material A were set tomatch water: s=76.58 ∞=4.73 τ=9.3 ps. MaterialB was defined by s=87.37 ∞=68.13 τ=90.9 ps. . . . 69Figure 5.2 Structural design of the proposed resonator sensorwith dimensions. . . . . . . . . . . . . . . . . . . . . . 71Figure 5.3 S21 response of the unloaded resonator from the HFSSsimulation software. . . . . . . . . . . . . . . . . . . . 72Figure 5.4 Field distribution plot (a) Electric field distributionat 2.57 GHz and (b) Magnetic field distribution at2.75 GHz. . . . . . . . . . . . . . . . . . . . . . . . . . 73Figure 5.5 (a) Layout of the sensor structure from the HFSSsoftware, (b) simulations results from HFSS whenthe channel is filled with liquid metal in the lengthof 18-32 mm of microfluidic channel referenced withunloaded resonator response, (c) change in resonantfrequency and (d) resonant amplitude with varyinglength of liquid metal. . . . . . . . . . . . . . . . . . . 74Figure 5.6 (a) Simulations results from HFSS when liquid metalis filled in the length of 25 mm of the microfluidicchannel, and different solid materials are tested and(b) scatter plot indicating the achieved resonant fre-quencies over materials with different permittivity. . . 75Figure 5.7 Frequency difference over different dielectric constantmaterials when liquid metal is in different locationsinside the microfluidic channel. . . . . . . . . . . . . . 76xivLIST OF FIGURESFigure 5.8 Simulation results from HFSS while varying loss tan-gent of test samples of permittivity 16.5, 10.0, and 5.7at different liquid metal settings inside the microchan-nel. Varying loss tangent of test samples varies theamplitude of the sensor response while liquid metalis at (a) 18 mm, (b) 21.5 mm, (c) 25 mm, (d) 28.5mm, and (e) 32 mm location. The magnification ofthe amplitude difference is shown in the bottom rightinset at every liquid metal setting. . . . . . . . . . . . 77Figure 5.9 Amplitude difference compared against the loss tan-gent for a MUT of permittivity 10 at different loca-tions of liquid metal. . . . . . . . . . . . . . . . . . . 78Figure 5.10 Simulated result from HFSS while testing the mate-rials at (a) 2.258 GHz and (b) 2.55 GHz of frequency. 79Figure 5.11 Experimental setup of the designed resonator usedfor testing standard samples (solid) at different res-onant frequencies. The magnification of the sensorwith MUT is inset in the top right corner. . . . . . . 81Figure 5.12 Measured response of liquid metal being at differentlocations inside the microchannel resulting in tunableresonant frequencies. . . . . . . . . . . . . . . . . . . 82Figure 5.13 Measured response of standard materials (r=2.20,6.15 and 10.20) influencing resonant frequency of 2.68GHz. . . . . . . . . . . . . . . . . . . . . . . . . . . . 83Figure 5.14 Measured response of standard material (r=6.15) in-fluencing different resonant frequencies. . . . . . . . . 84Figure 5.15 Sensitivity of the designed resonator sensor by test-ing samples of permittivity 2.20, 6.15 and 10.20 indifferent resonant frequency. . . . . . . . . . . . . . . 85Figure 5.16 Experimental setup of the coaxial probe measurement. 86Figure 5.17 Measured real (′) and imaginary part (′′) of the com-plex permittivity of different solutions where (a) 15%ethanol-water and 10% IPA-water, (b) 30% ethanol-water and 25% IPA-water, (c) 40% ethanol-water and35% IPA-water and (d) 65% ethanol-water and 55%IPA-water. . . . . . . . . . . . . . . . . . . . . . . . . 88Figure 5.18 Dielectric response of (a) 80% and 85% ethanol-watercompared with pure methanol and (b) 80% and 85%IPA-water compared with pure ethanol. . . . . . . . . 89xvLIST OF FIGURESFigure 5.19 Experimental setup of the designed sensor used forsensing liquid samples of pure methanol, pure ethanol,81% ethanol-water and 83% IPA-water solutions atdifferent resonant frequencies. . . . . . . . . . . . . . 91Figure 5.20 Measured frequency shift due to the presence of puremethanol and 81% ethanol-water at resonant frequen-cies of (a) 2.817 GHz, (b) 2.719 GHz, and (c) 2.685GHz. The corresponding liquid metal position andfrequency shifts are shown in the top right inset andbottom right inset, respectively. . . . . . . . . . . . . 92Figure 5.21 Measured frequency shift due to the presence of pureethanol and 83% ethanol-water at resonant frequen-cies of (a) 2.817 GHz, (b) 2.719 GHz and (c) 2.685GHz. . . . . . . . . . . . . . . . . . . . . . . . . . . . 93xviAcknowledgementsSome of the very important people need to be recognized here because oftheir unwavering support and mindful contribution to my thesis. I would liketo begin by thanking Prof. Mohammad Hossein Zarifi, my supervisor, whoencouraged me in the first place to pursue master’s in microwave resonatortechnology. His continuous assistance and tenacious encouragement alwayspush me beyond my level to achieve the desire. Every meeting with himbrings a new idea and inspires me to think outside the box. His mentorshiphas provided me with so many new prospects, and I have learned a greatdeal from him. While working with him, I have learned so many technicaldetails and developed a tremendous theoretical knowledge which will helpme further. His undiluted guidance and sincere advice help me develop manymore skills that I can utilize in my professional and personal life. I considermyself very fortunate to work under the supervision of such a generous andencouraging professor like him.I would also like to thank my committee members, Prof. Ian Foulds,Prof. Kevin Golovin, and Prof. Sina Kheirkhah, for their excellent guid-ance throughout the Master’s program. Besides, I would like to acknowledgethe contribution of the Natural Sciences and Engineering Research Councilof Canada (NSERC) Discovery Grant Program, Canada Foundation for In-novation program for partial financial assistance, and CMC Microsystemsfor the use of their software and equipment.I would like to extend my gratitude by thanking Benjamin Wiltshire,Ph.D. student, for mentoring me to the necessary software and hardwareskills and being actively involved in developing and validating the presentedwork. His previous experience and in-depth knowledge benefit me to trans-form the failures into success. My sincere thanks go to my colleagues atthe OMEGA (Okanagan Microelectronics & Gigahertz Application) Lab forall their support, constructive suggestions, and being open to questions atany time. Finally, I would like to appreciate my parents undivided attentionthroughout my post-graduate journey at the University of British Columbia.A special thanks to my wife, Nawshin Tabassum, for always being there asmy mental support.xviiDedicationTo my father, Abdul Wazed,who has always been my inspiration.To my mother, Abeda Begum,who has always shown unconditional love and support.To my loving wife, Nawshin Tabassum,who has always been there as a motivator and friend.xviiiChapter 1Introduction1.1 MotivationSensors play an important role in numerous modern industrial applica-tions. The term “Sensor” generally refers to a device that detects events orchanges in its environment and then provides a corresponding output, typi-cally electrical or optical signals, read by an observer or by an instrument.Most sensors are electronic, but some are simple, such as a mercury-in-glassthermometer, which provides visual data. The use of sensors has grownbeyond the most conventional fields of temperature, pressure, or flow mea-surement with developments in micro-machinery and easy-to-use microwaveplatforms. The preferred choice of sensors is determined by different as-pects such as sensor cost, size, complexity, and the form of measurement.Microwave sensors have gained popularity regarding detecting and charac-terizing different materials of every state, such as solid, liquid, semi-liquid,and gas. It utilizes the microwave portion of the electromagnetic spec-trum that occupies the wavelength range from approximately 1m to 1mm.Longer wavelength microwave radiation can pass through the cloud, wind,dust, snow, and this property enables microwave energy to be detected un-der almost all weather and environmental conditions. Moreover, microwavesensors are highly sensitive and can detect very small changes in the en-vironment. Due to the characteristics, microwave sensors are widely usedto detect and characterize different materials dielectric properties. Everymaterial shows distinctive dielectric behavior based on the operating fre-quency due to their unique dielectric properties, thus transmitting differentdielectric responses at different frequencies. Furthermore, microwaves havedemonstrated other interesting sensing properties, such as low-cost, mate-rial interaction at various scales, compatibility with planar technology, androbustness.Microwave sensors can be divided into resonant and non-resonant meth-ods for the identification and characterization of dielectric materials. Thenon-resonant methods are based on wave propagation and suitable for broad-band measurement. Resonant methods are more precise and appropriate11.1. Motivationfor low loss material characterization; however, only a single frequency or adiscreet number of frequencies may provide measurement results. The com-bination of sensor size and performance of a resonant-type sensor is hard toachieve with non-resonant methods. Compared to wide-band non-resonantstructures, resonance dependent sensors have higher resolution in their per-formance, even though limited in the monitoring of discrete finite frequen-cies. Resonant sensors have the advantages of providing higher sensitivity,simpler signal processing, and lower cost compared to non-resonant sensors.This inspires researchers to integrate resonators into sensing actions, often ata single frequency, and extract the material’s dielectric properties in variousenvironments using the resonators behavioral variation. Recent investiga-tions show a comprehensive versatile application scope of the microwaveresonators.Since microwave resonator sensors have already been successfully promi-nent in sensing solid, liquid, and gas, the innovation of tunable resonanceoptimizes the sensing technology more extensively. Tunable microwave res-onator sensors can lower the cost much more due to its wideband perfor-mance and reusability. It also reduces the complexities of a microwave sys-tem by minimizing the number of circuits required for different operatingconditions. Researchers have utilized different modes (harmonics) of res-onators to detect materials dielectric properties in multiple points insteadof one. This benefits in achieving more comprehensive data over multipleresonant points and understanding the material behavior at different fre-quencies. However, this technique limits the accuracy of the measurementwhen higher resonant modes are used to characterize the sample. In recentyears, researchers employed liquid metal filled fluidic channel in differentlocations of a microwave resonator to tune resonant frequency into multipleresonant points. Despite the advantage of highly accurate material char-acterization over multiple resonant points, this technique can only providetwo or three discrete resonant points over a wideband. Therefore, devel-oping a continuously tunable microwave resonator would be very attractivefor resonant-based spectroscopy development. Through this work, a proof ofconcept of developing a wideband continuously tunable microwave resonatorfor the development of resonant based microwave spectroscopy in microwavetechnology will be presented.21.2. Objective1.2 ObjectiveThis thesis aims to design a wideband continuously tunable planar mi-crowave resonator sensor to develop resonant-based microwave spectroscopyand utilize the resonant-based spectroscopy to sense materials of similar di-electric properties. The designed sensor should provide continuous tuning ofresonant frequency and detect materials in multiple operating points. Con-sidering the aim of the thesis, the objectives of this thesis work are presentedbelow:The first objective of this thesis is to design a planar microwave resonatorstructure to achieve user-controlled continuous tuning of resonant frequencyover a wideband. To achieve this objective, the influence of liquid metal onthe planar resonator has been investigated. Subsequently, the advantagesof 3D printing technology have been fully utilized by creating 3D printedmicrochannels to guide the fluid. The entire process contains a suitable res-onator structure design, use of the 3D printed fluidic channel, liquid metalintegration, and optimization of the overall model to attain the desired aim.The progress of this objective will be measured by analyzing and compar-ing the data collected from the experiment and structural simulation. Theachieved result should reflect continuous tuning of the resonant frequencyof the user’s interest over a wideband of frequency.The second objective of this thesis is to utilize the developed resonant-based spectroscopy by tuning the resonant frequency to detect sample ma-terials that exhibit similar dielectric properties. To achieve this objective,the designed resonant structure was modified and integrated with a newlyfabricated 3D printed fluidic channel to guide the liquid metal and the ma-terial under test (MUT). Initially, solid samples of different dielectric prop-erties were tested to characterize the designed sensor structure, followedby preparing liquid samples of similar dielectric properties. The progressof this objective was measured by analyzing the data collected from themeasurement. The achieved result would show successful implementationof resonant-based spectroscopy of tuning the resonant frequency to detectsamples of similar dielectric properties.1.3 Thesis OrganizationThis thesis contains six chapters where Chapter 1 discusses the mo-tivation and objectives of the work. Chapter 2 outlines an introductionto microwave technology and a discussion on different planar transmission31.3. Thesis Organizationlines. The discussion then continuous to different available measurementtechniques in microwave technology. Chapter 3 begins with the operatingprinciple of the planar microstrip resonator in addition to a backgroundinvestigation on different tuning mechanisms in microstrip resonator. Thebenefits of microwave spectroscopy in terms of material characterization isalso identified and mentioned in this chapter. Chapter 4 presents the designmethodology of the tunable microwave resonator and RF characterization ofthe designed resonator. The designed resonator’s performance was validatedby comparing the measured and simulated results and presented in this chap-ter. The developed tunable resonant structure was modified and utilized asa tool to successfully distinguish between samples of similar dielectric prop-erties and detailed in terms of the design idea, and RF characterization ispresented in chapter 5. Chapter 6 is devoted to the conclusion of the the-sis that summarizes the thesis achievements and outlines some future studydirections.4Chapter 2Introduction to microwavetechnology, planartransmission lines andmeasurement techniquesThis chapter starts with introducing microwave technology and is fol-lowed by presenting different planar transmission line structures. One ofthe significant applications of microwave technology is material detection indifferent environments, and the fundamentals of material detection in mi-crowave technology are presented in this chapter. There are different mea-surement techniques available in microwave technology, and choosing thebest one depends on the material structure and the frequency of interest.This chapter closes with a detailed discussion of different measurement tech-niques based on their operation, advantages and limitations, and frequencyrange.2.1 Introduction to microwave technologyElectromagnetic rays with frequencies between 300 MHz and 300 GHzin the electromagnetic spectrum, with a corresponding electrical wavelengthbetween 1m to 1mm, are referred to as microwaves. As compared with thewaves used in radio broadcasting, microwaves are smaller. Their spectrumranges from radio waves to infrared waves. Microwaves travel by line ofsight, and the troposphere can impact them slightly. They do not requireany medium to travel. Microwaves are reflected by metals and are partlytransparent to nonmetals such as glass and particles [1].Microwave is a portable communication line-of-sight platform that en-ables high-speed wireless connections to send and receive audio, video, anddata information using high-frequency radio wave beams. Microwave con-nections are commonly used for point-to-point communications because their52.1. Introduction to microwave technologyshort wavelength enables easily sized antennas to direct them into narrowbeams, directly guided at the receiving antenna. This allows microwave sys-tems nearby to use the same frequencies without interfering with each otheras radio waves of lower frequency do [2]. Another benefit is that the highmicrowave frequency has a very broad information-carrying capacity for themicrowave band; the microwave band has a bandwidth 30 times that of allthe rest of the radio spectrum below it. Figure 2.1 shows the electromag-netic spectrum to indicate the wavelength and frequency correlated with thevarious bands.Figure 2.1: Electromagnetic frequency spectrum and associated wave-lengths. Copyright © 2008 Taylor & Francis Group, LLC. All rights re-served [3].Over the years, microwave technology has experienced substantial growthin terms of power consumption, scale, and cost due to the advancement ofsolid-state devices, technology for printed circuit boards, and integrated cir-cuit technology. In satellite communications, radar signals, phones, andmarine applications, microwaves are used most widely. Due to the enor-mous improvement in microwave technology, it is found to be utilized invarious industries such as communication [4, 5], manufacturing, process-ing for drying and curing [6–10], healthcare [11–13] and food industry [14].The use of microwave technology in biomedical and pharmaceutical appli-cations widens the range of possibility of early detection of deadly diseases.Many important applications include detecting and monitoring food quality,drug research, human tissue characterization for tumor and cancer detection[15, 16], brain activities detection [17], concealed weapon detection [18] andmotion detection [19]. Researchers prefer microwave technology due to its62.2. Planar transmission lineslow cost, robust, compact, non-contact, and label-free detection properties.2.2 Planar transmission linesIn the early days, RF (radio frequency) and microwave systems dependon waveguides, two-wire lines, and coaxial lines for the low-loss transmissionof power in high frequencies. Waveguides can handle high-power applica-tions with low loss, but they are bulky and expensive. Two-wire lines arelow-cost, but they lack shielding [20]. Coaxial lines are shielded, but theyare a difficult medium to fabricate complex microwave components [20]. Analternative, in the form of strip lines, microstrip lines, slot lines, coplanarwaveguides, and many other forms of similar geometries, is given by planartransmission lines [21]. Because of the trend toward miniaturization and in-tegration, most modern microwave circuitry is fabricated using planar trans-mission lines such as microstrips and stripline rather than waveguides [20].These transmission lines are lightweight, cost-effective, and easily integratedinto active circuit devices to form integrated microwave circuits [20]. Themost fascinating feature of planar transmission lines operating in microwavefrequencies and beyond is that such lines can replace lumped elements, suchas capacitances and inductances, thus eliminating lumped components thatincrease the complexity and costs of the circuit [21].The configuration of the planar transmission line includes both the trans-mission line selection and the substrate. Each structure of the planar trans-mission line comprises a combination of metal and dielectric. In most cases,the dielectric primarily supports the metal pattern, serves as a substrate,and affects the wave’s propagation. In this section, different types of planartransmission lines will be discussed.2.2.1 MicrostripThe easiest structure to produce with a thin dielectric substrate withmetal on both sides is the microstrip. One metal sheet is held as the electri-cal ground plane, while the other is patterned by way of photolithography.The metal is chemically etched to form a microstrip transmission network.The cross-section of a microstrip transmission line is shown in Figure 2.2(a).Under the current-carrying conductor traces, the ground plane avoids exces-sive field leakage and thereby reduces the radiation loss [22]. The presence ofa ground plane in the backside of the substrate effectively isolates the struc-ture from the backside region and prevents it from potential interferenceeffects caused by other circuits or materials [21].72.2. Planar transmission linesFigure 2.2: Cross sections of different planar transmission line structureswhere (a) microstrip, (b) coplanar waveguide, (c) coplanar strip and differ-ential line, (d) stripline and (e) embedded differential line. Copyright ©2019 by M.B.Steer. All rights reserved [22].2.2.2 Coplanar waveguideCoplanar waveguide (CPW) can be formed with the active metallizationand the ground planes on the same side of the substrate. Each side-planeconductor is grounded, and the signal is carried by the center strip, so ascompared to the microstrip, much less field reaches the substrate. Due tothe center conductor’s presence and the proximity of the ground planes,coplanar waveguides are especially useful for making active circuitry [20].CPW’s may also be implemented with backside metallization (also knownas conductor backed CPW’s). However, due to the parasitic parallel-platewaveguides present on both sides of the CPW axis, this backside metalliza-tion can cause leaky-wave propagation [21]. The cross-sectional geometryof a CPW is shown in Figure 2.2(b). CPW does a good job of reducingradiation; it has low-frequency dispersion and is preferred to microstrip forlarge spatially distributed circuits at frequencies above 20 GHz or so. Thereare disadvantages to it, including the increased area required and the needto use field straps (compared with microstrips) [22].2.2.3 Coplanar strip and differential lineThis planar transmission line structure is simple and consists of two con-ductors in the same plane above the dielectric substrate. When one of theconductors is grounded, this form is called coplanar stripline or coplanarstrip (CPS). CPS is used as an area-effective variation of CPW in this con-figuration. If neither conductor is grounded and the line is operated, the82.3. Microwave technology in material detectioninterconnection is called a differential line. A differential line is commonlyused in high-speed optical networks with RFICs, and essential networks[22]. Both types have nearly similar electrical properties, with variationsfrom communicating with other metallic structures such as ground planes.The cross-section of the coplanar strip and differential line is shown in Figure2.2(c).2.2.4 StriplineStripline is a planar transmission line that is well suited for integrated mi-crowave circuitry, miniaturization, and photolithographic manufacture [21].The geometry of the stripline is shown in Figure 2.2(d). A thin conductingstrip is centered between the two conducting ground planes, and the re-gion between the ground planes is filled with a dielectric material. Striplineis typically built on a grounded dielectric substrate by etching the centerconductor and then covering it with another grounded substrate [20]. Incomparison with the microstrip line, the stripline manufacturing process iscomplicated because it requires multiple layers to support the embeddedtrace between two ground planes [22]. The selection of the transmission linedepends on the need of the design [23].2.2.5 Embedded differential lineThis simple transmission structure shown in Figure 2.2(e), is created byhaving only two conductors embedded in a substrate with no unique groundplane. It follows the same manufacturing procedure as stripline, exceptin embedded differential line, there are two conductors. This structure isincidental to the potential presence of ground planes, and preferably theseshould not impact the field pattern [21]. In multilayer boards, this tracearchitecture is more convenient and has less attenuation and radiation thanin microstrips at higher frequencies [24].2.3 Microwave technology in material detectionWith industrial advancement, the modern industry depends on data ob-tained from an immense number of sensors in today’s date. These sensorsnot only need to display sensitivity, selectivity, and high resolution in theirservice, but they should also be able to withstand environments that areharsh, inaccessible, and dangerous. Microwave technology can be easily92.3. Microwave technology in material detectionimplemented in sensing, and monitoring applications in unpleasant environ-ments [25]. Microwave devices have recently shown significant efficiency indetecting and observing solid, liquid, and gaseous materials [26–28]. Thesedevices have gained quite a popularity because of real time and highly ac-curate sensing.The operating principle of material sensing in microwave technology uti-lizes the electromagnetic wave interaction between an input signal and thematerial in the sensors local environment, which translates information intoa quantifiable electric signal [29]. This information can be used to charac-terize the material, allowing better processing and decision making of theinformation [30]. In this case, the dielectric properties of the materialsplay a vital role because of their unique characteristics. Coaxial probes,waveguides, dielectric resonators, and planar microstrip resonators are themicrowave sensors in use, each following the general working theory of trans-lating the dielectric properties (permittivity and conductivity) of materialinto factors of impedance and scattering, to be analyzed by readout circuitry[25, 31, 32].Figure 2.3: A dielectric with polar molecule: (a) in the absence of an externalelectric field Eext, (b) in the presence of an external electric field Eext and (c)the induced electric field Eind inside the dielectric produced by the inducedsurface charge Qi. Copyright © 2018 Rice University. All rights reserved[33].A material is categorized as “dielectric” if it can store energy while anexternal electric field is applied. When a DC voltage source is applied acrossa parallel plate capacitor, more charge is stored if there is a dielectric mate-rial compared to no material (vacuum) in between the plates. The dielectricmaterial between the plates increases the capacitor’s energy storage capacityby neutralizing the electrode charges, which would usually contribute to the102.3. Microwave technology in material detectionexternal field [33]. The effect of dielectric will be understood by gazing atits behavior at the molecular level. In the absence of any external electricfield Eext, the electrical dipoles of the molecules are randomly oriented, asshown in Figure 2.3(a). When the dielectric material is placed in an externalelectric field, the dipoles align with the external field, as shown in Figure2.3(b). Opposite charges on neighboring dipoles neutralize each other withinthe dielectric volume, so there is no net charge within the dielectric volume[33]. This, however, is not the case very close to the upper and lower surfacesthat border the dielectric (rectangular dash lines in Figure 2.3(b)), wherethe alignment produces a net charge. The induced surface charges (+Qi and−Qi) produce an additional electric field Eind which opposes the externalfield shown in Figure 2.3(c) [33].The ability of a material to resist the electric field is called the permittiv-ity of that material. The permittivity of free space is the resistance offeredby the vacuum between two conducting plates. Dielectric constant or rel-ative permittivity indicates how easily a material can become polarized bythe imposition of an electric field on an insulator. Relative permittivity isexpressed as the ratio of a substance’s permittivity to the permittivity offree space or vacuum.r =o(2.1)Where r refers to the relative permittivity and  is the permittivity ofthe substance (F/m) and o is the permittivity of the vacuum or free space(8.85×10−12 F/m). So, r > 1 indicates that a material can store moreelectric energy in a volume that can be stored in a vacuum.In general, permittivity is a complex number where the real part de-scribes the energy storage and the imaginary part describes energy dissi-pation between the electromagnetic field and the material. Energy storagerepresents the lossless component of energy exchange between the field andthe material, and energy dissipation occurs when electromagnetic energy isconsumed by the material [34]. The complex permittivity can be expressedas,r = ′ − j′′ (2.2)Where the real part (′) defines the dielectric constant and the imaginarypart (′′) accounts for the loss in the medium. Dielectric material losses areprimarily caused by dielectric damping (ω′′) and conduction loss (σ). Arelated quantity of the interest is loss tangent, which can be expressed as,112.3. Microwave technology in material detectiontanδ =ω′′ + σω′(2.3)Where ω′′ + σ is accounted for total effective conductivity. The termloss tangent (tanδ) is also called by terms such as tangent loss, dissipationfactor, or loss actor.Materials are usually characterized by these two properties (dielectricconstant and loss tangent), which are highly frequency dependent. Mi-crowave sensor devices utilize these frequency-dependent dielectric prop-erties to characterize the materials. Figure 2.4 quantitatively shows thebehavior of dielectric properties as a function of frequency [34].Figure 2.4: Complex dielectric permittivity spectrum over a range of fre-quencies where ′ and ′′ denote the real and imaginary part, respectively.All rights reserved [35].122.4. Different measurement techniques in microwave technology2.4 Different measurement techniques inmicrowave technologyThe measurement techniques involve measuring the scattering parame-ters (S-parameters) using a vector network analyzer (VNA). The particularmethod used depends on the frequency range of interest and the type oftarget material. There are different measurement techniques available inmicrowave technology. Four significant methods of material detection inmicrowave technology will be discussed here.2.4.1 Transmission/reflection (TR) line methodFigure 2.5: Measurement using TR method with a waveguide. All rightsreserved [36].In a transmission/reflection (TR) line method, the sample material isplaced inside a portion of an enclosed transmission line. The line is typicallya section of the rectangular waveguide or coaxial line. The VNA connectedto the ports of the transmission line measures the S-parameters and recordsboth the reflected (S11) and transmitted signal (S21). This method requiressample preparation (machining) so that the sample can fit tightly into the132.4. Different measurement techniques in microwave technologywaveguide or coaxial line. Figure 2.5 shows a typical measurement usingTR method with a waveguide.Calibration of the VNA must be taken out before making the measure-ment. After the calibration of the VNA, the prepared sample is placed ina sample holder. To reduce the uncertainty of the measurement caused byair gaps, the sample must be tightly fit in the sample holder. This methodis used widely for the extraction of complex permittivity of materials. Forinstance, Costa et al. utilized this technique for electromagnetic characteri-zation of materials because of wider bandwidth, but he also mentioned thesample preparation complexity [37]. On the other hand, Maleki et al. usedthe transmission line method to characterize the ferrite materials due toeasy implementation and no requirement to measure the phase [38].This technique benefits to measure samples with medium to high loss[36, 39, 40]. The disadvantage of this method is the limited measurementaccuracy due to air-gap effects. It also limits the accuracy when the lengthof the sample is the multiple of one-half wavelength in the material [36].2.4.2 Open-ended coaxial probe methodThe open-ended coaxial probe method is widely used as a non-destructivetesting method. It is a modification of the transmission line method. Thistechnique calculates the dielectric properties from the phase and amplitudeof the reflected signal at the end of an open-ended coaxial line inserted into asample to be measured [41]. The main component of this method is the high-temperature probe, which includes a circular flange with an aperture in thecenter. The properties of the sample material are measured by immersingthe probe into a liquid or touching a flat surface of a solid. The fields atthe end of the probe fringe into the substance and change when they comeinto contact with the sample material. The reflection coefficient is measuredusing a VNA. The setup for this type of measurement is shown in Figure2.6. This technique is best suited for liquids, and semi-liquids [42].Calibration at the tip of the probe must be performed before takingthe measurement. The calibration corrects the directivity, tracking, andthe source match that can be presented in the reflection measurement inthe VNA [42]. The standard calibration procedure requires a three-fold(open, short and match) calibration [36]. The three known standards areair, a short circuit, and distillate and de-ionized water [42]. As a calibrationnorm, the referenced liquid is used and must be a liquid with ”known”dielectric properties. The reference liquids are commonly selected as water,saline, and methanol. To eliminate the measurement’s systematic errors,142.4. Different measurement techniques in microwave technologyFigure 2.6: Experimental setup of an open-ended coaxial probe method. Allrights reserved [36, 42].the difference between the predicted and actual values is used. There areadditional sources of error (such as cable stability, air gaps and samplethickness) even after calibrating the probe that can impact the probe andthe measurement precision [42]. One needs to be very careful about cablestability and air gap before taking the measurement.The open-ended coaxial probe method is prevalent because it allows flex-ibility and permittivity measurements at high temperatures. Santos et al.utilized this technique for permittivity measurements of liquid at high tem-peratures [43]. The open-ended coaxial probe technique is used to measuresnow water content in [44] by Mavrovic et al. One of the benefits of thisprocedure is, after calibration, the dielectric properties of a large numberof samples can be measured within a very short time. This technique canbe utilized to measure the dielectric properties over a wide range of fre-quencies. For instance, the effect of different measurement probe on tissuecontact pressure was monitored in a wide frequency range from 0.5 to 2.5GHz in [45]. While using this technique, cautions must be exercised becauseerrors are introduced at very low frequencies and very high frequencies, aswell as with low dielectric constant and loss factor values [41]. Limited ac-curacy under certain conditions is a drawback of the coaxial probe systemcompared with the other methods such as transmission line, free space, orresonant [42].152.4. Different measurement techniques in microwave technology2.4.3 Free space methodFree space techniques use antennas concentrating microwave energy onor through a sheet of material or slab. As the sample is never touched or con-tacted, high-temperature measurements are simple to perform in free space.It generally operates in wideband frequencies [42]. Besides, this can easilybe used for continuous monitoring and control in industrial applications.For instance, Kraszewski et al. presented various aspects of measuring andcontinuous monitoring of moisture content in solids and liquids using freespace technique [46].Figure 2.7: Measurement of sample under test using free space method. Allrights reserved [36].In a free-space technique, the sample is placed between a transmittingantenna and a receiving antenna, and the attenuation and phase shift of thesignal is measured using a VNA, as shown in Figure 2.7. The measurementrequires the sample to be large and flat [42]. Initially, by positioning theempty sample holder halfway between the two antennas, the s-parameters ofan empty sample holder is determined. The sample is then placed betweenthe antennas on the sample holder, and the s-parameter measurement is162.4. Different measurement techniques in microwave technologyagain performed. The VNA’s de-embedding function is used to cancel thesample holder’s effect, and only the s-parameter of the sample can be cal-culated [36]. By utilizing this method, both the reflection and transmissioncoefficient of the s-parameters can be determined.Prior to measurements, calibration of the network analyzer is manda-tory. It is not easy to calibrate the network analyzer for free space as theyare connector-less. Moreover, there are a variety of methods of calibrationwhich can be used, such as through-reflect-line (TRL), through-reflect-match(TRM), and the line-reflect-line (LRL) [36]. However, The LRL calibrationmethod will generate the highest level of calibration [36].This measurement technique benefits the characterization of materials inhigh temperature, and hostile environment [47–51]. The free space methodcan also be used over a wide range of frequencies. Hakansson et al. utilizesthis technique to measure the dielectric properties of polymerized films onsynthetic textile substrates over a wide range of frequency from 1-18 GHz[52]. On the contrary, Trabelsi et al. accounted for multiple reflections,mismatches, and diffraction effects at the edges of the sample, which are thecommon drawbacks of this method [41]. Special attention must be paid tothe choice of the radiating elements, the sample holder’s configuration, andthe sample geometry and position between the two radiating elements toimprove the measurement accuracy [41].2.4.4 Resonant methodThe most accurate methods of obtaining the dielectric properties of ma-terials are resonant methods because the resonant cavities are high qualityfactor (Q) structures that resonate at specific frequencies [42]. The resonantfrequency and the Q-factor of the cavity are influenced by a piece of samplematerial inserted into the cavity. The complex permittivity of the materialcan be determined from these parameters at a single frequency.Perturbation method and low loss measurement method are the twotypes of resonant measurement methods which are widely used [36]. Pertur-bation methods are suitable for medium to high loss material measurementwhereas, and the low loss measurement method is a measurement of low lossmaterials using a larger sample [36, 53]. In Figure 2.8, a typical setup of theresonant method is shown where the perturbation method is utilized espe-cially using a transverse magnetic (TM) cavity geometry to determine thedielectric properties of a thin rod. Initially, the measurement of an emptycavity is performed by measuring the resonant frequency and Q-factor. Thesecond step is to place the sample thin rod along the center of the cavity172.4. Different measurement techniques in microwave technologyand record the new resonant frequency and the Q-factor in the VNA. Withthe sample in the cavity, any field shifts can result in the measured resonantfrequency or Q-factor changing. Using the frequency, volume, and Q-factor,the sample’s dielectric properties can then be computed [36].Figure 2.8: Measurement of a thin rod using resonant method. All rightsreserved [36].Resonant-based method is very popular for obtaining complex dielectricproperties of materials such as permittivity due to its non-contact, portable,label-free sensing, and highly accurate measurement. Saeed et al. presentedthe use of substrate integrated waveguide cavity resonators to accuratelycharacterize the material properties due to the attractive features such ashigh sensitivity and high Q-factor [54]. In [55], Tian et al. proposed a mi-crowave resonant cavity-based sensor for measuring the moisture content inwet granules. The variable moisture content in the cellulose granule impactsthe resonant frequency and Q-factor differently. The frequency difference isutilized as a signature to identify the different moisture levels.Among the discussed measurement techniques in this section, the reso-nant based technique uses less expensive devices [56]. Machining the samplesto fit properly in the cavity is an essential part of the measurement sincea limited air gap can cause an error in the measurement. Microstrip line182.4. Different measurement techniques in microwave technologysensor can be utilized because it offers the advantages such as low-cost fab-rication, portability, robustness, non-contact, small sample size, and ease ofintegration with microfluidic chips [54, 57–62].19Chapter 3Planar microstrip resonator,tuning mechanisms andmicrowave spectroscopyThis chapter aims to address the operating concept of a planar microstripresonator and explore the wide variety of versatility in design. It is accompa-nied by a detailed discussion of various ways of tuning the resonant frequencyused in the past and particular attention given to tuning based on microflu-idics. An inquiry into the advantages of microwave spectroscopy in materialdetection has been described and provided in this chapter.3.1 Planar Microstrip ResonatorPlanar microstrip resonator is one of a kind of microstrip transmissionlines that is characterized by a resonant frequency (ωo). This frequencygives a rate of periodic exchange of electric and magnetic energy in thesystem. Among the other transmission lines discussed in section 2.2, inengineering, microstrip-based technology is well known to transmit signalsin the microwave regime, and this is mainly due to the compact natureof the resonant microstrip circuits [63]. Planar microstrip resonators havegained quite popularity in different industries like bio-medical, healthcare,pharmaceuticals, food processing, agriculture, and communication due to itseasy fabrication, non-contact measurement, real-time response, robust andhigh sensitivity [57–62].3.1.1 Theory of operationMicrostrips are planar transmission lines that consist of a conductingstrip and a ground plane separated by a dielectric. In a transmission line,inductance (L), capacitance (C), resistance (R), and conductance (G) areevenly distributed along the length of the line [20]. It is convenient to203.1. Planar Microstrip Resonatormodel small segments of a transmission line in terms of lumped elements,as presented in Figure 3.1. It helps to understand the model elements asa representation of physical phenomena occurring in the transmission. Theinductance per unit length L′ describes the inductance associated with theground plane and the conductor. The capacitance per unit length C ′ origi-nates from the immediate vicinity of the conducting layers. The resistanceper unit length R′ is the conductors’ resistance, and the shunting conduc-tance per unit length G′ describes the losses in the dielectric layer.Figure 3.1: Lumped element representation of an infinitesimal part ∆Z ofa microstrip transmission line where (a) the actual microstrip and (b) themodelled microstrip. All rights reserved [63].In practical applications, transmission lines exhibit low loss [64]. Usingthe Telegraphers equations [20], the wavelength (λ) of the signals propagat-ing along the transmission line can be determined by the equation,λ =2piω√L′C ′(3.1)The phase velocity (vph) is given by the formula [20],vph =1√L′C ′(3.2)Two important quantities defining a transmission line are the character-istic impedance (Zo) and the propagation constant (β), which can be definedas,Zo =√L′C ′(3.3)213.1. Planar Microstrip Resonatorβ = ko√eff (3.4)Where ko is the propagation constant of the surrounding medium andeff is the effective permittivity of the surrounding medium.When a time-varying electromagnetic signal travels through a transmis-sion line, it generates electric and magnetic fields around it. The electromag-netic wave propagating on a microstrip line exists in two separate media: air(above the conducting trace) and substrate (between conducting strip andground plane). The fields have to follow electromagnetic boundary condi-tions at the air-substrate interface. These boundary conditions contributeto components of the longitudinal electrical and magnetic field. The elec-tric and magnetic field lines of the microstrip exist in the dielectric layer aswell as in the surrounding medium, as shown in Figure 3.2. The effectivepermittivity of the microstrip is given approximately by [20],Figure 3.2: Electric (E-field) and magnetic (H-field) field distribution of amicrostrip line. All rights reserved [65].when (W/h ≥ 1),eff ≈ r + 12+r − 12 1√1 + 12 hW (3.5)when (W/h < 1),eff ≈ r + 12+r − 12 1√1 + 12 hW+ 0.04(1− Wh)2 (3.6)Where W is the width of the conducting strip and h is the thickness ofthe dielectric substrate. The characteristic impedance (Zo) of a microstripline is determined by its geometry and the effective dielectric constant,223.1. Planar Microstrip Resonatorwhen (W/h ≥ 1),Zo ≈ 120pi√eff [W/h+ 1.393 + 0.667 ln(W/h+ 1.444)]Ω (3.7)when (W/h < 1),Zo ≈ 60√effln(8hW+W4h)Ω (3.8)Such a transmission line functions as a resonator, the length of which isequal to half of its ground mode wavelength [20, 63, 65]. It is called a λ/2resonator. It is convenient and common practice to model a resonator as aparallel connection of a resistor, an inductor, and a capacitor in the vicinityof the resonant frequency [20, 66–69] as shown in Figure 3.3. The lumpedelement model of a transmission line resonant circuit is mainly based on thebook by David Pozar [20], and article of Martin [67]. The input impedanceZin (Ω) of this circuit is given by,Zin =(1R+1iωL+ iωC)−1(3.9)Where ω is the frequency and i is the imaginary part. The complexinput power Pin at the input of the resonators is given [68],Figure 3.3: Lumped element model of a microstrip resonant circuit. Allrights reserved [63].233.1. Planar Microstrip ResonatorPin =12|V |2(1R+1iωL+ iωC)−1(3.10)Where V is the voltage, the 1/R term represents the energy (Wloss) dis-sipated in the resistor. The complex terms define the stored magnetic (Wm)and electric energy (We) in the inductor and the capacitor, respectively.Resonance occurs when the planar resonant structure is able to transformthe magnetic energy to electric energy and to fulfill this condition, Wm = Wemust hold. Therefore, the resonant frequency (ωo) of the structure shouldbe,ωo =1√LCrad/s (3.11)This is the frequency of the resonator’s fundamental mode. The resonantfrequency of the nth mode of the resonator is given by,ωn = nωo =1√LnC(3.12)where [67],Ln = Ln2 (3.13)and n≥1 is an integral number. Then the internal quality factor of theresonant circuit is given by,Qint = ωn2WmWloss=RωnLn= ωnRC (3.14)3.1.2 Resonant frequency and quality factorResonant frequency and quality factor are those parameters of a res-onator that determines the performance of the resonator. Resonant ampli-tude is the maximum value of amplitude when the resonance occurs in aresonator circuit. It is the maximum value of oscillation at which the reso-nant occurs in a resonant system. In the previous section, the basic theoryof calculating the resonant frequency and quality factor of a microstrip res-onator is described as a lumped element model. One can also easily estimatethe resonant frequency and Q-factor from the structural design, and this willbe presented here.243.1. Planar Microstrip ResonatorResonant frequencyResonant frequency is the oscillation of a system at its natural or un-forced resonance. Resonance occurs when a system can store and easilytransfer energy between different storage modes, such as kinetic energy orpotential energy. Most systems have one resonant frequency and multipleharmonic frequencies that get progressively lower in amplitude as they moveaway from the center. A system tends to oscillate at maximum amplitudeat certain frequencies, known as the system’s resonant frequency. Resonantfrequencies are essential in both mechanical and electronic/electrical sys-tems. Mechanical resonance can cause massive vibrations that can destroyobjects. On the other hand, electrical/electronic resonance is generally usedto tune a circuit. This may be parallel resonance used to maximize a signalone wants, to select a radio signal, or cause an oscillation. Series resonancemay be used to absorb a specific signal. Combinations are used to filtersignals selectively.For passive resonators, the length of the microstrip line plays a criticalrole in determining the resonant frequency of the system. The resonatorsusually are designed with the half-wave resonator concept; thus, the resonantfrequency can be estimated using the equation 3.15 [70].fr =c2l√effHz (3.15)Where l is the total length of the resonator microstrip line (m), eff is theeffective permittivity of the materials in the resonator ambient environment,c is the velocity of light (m/s) and fr is the resonant frequency (Hz). Sincethe speed of the light is constant and effective permittivity varies slightlywith the material, the resonant frequency mostly depends on the length ofthe resonator microstrip line. By varying the total length of the microstripline, the resonant frequency can be altered. So, it can be said from theequation 3.15 that the higher the length of the microstrip line, the lower theresonant frequency of that resonator.Quality factorA parameter which is closely related to the finesse is the quality factor(Q) of a resonator, which is a measure of the sharpness of the resonance[71]. For electronic circuits, the quality factor is defined as the ratio ofthe stored energy over the energy dissipated in one cycle. Therefore, anideal resonator with no loss has an infinite quality factor [72]. The quality253.1. Planar Microstrip Resonatorfactor of a resonator can also be determined by the resonant frequency tobandwidth ratio. The following equation is utilized to find the Q-factor ofa resonator.Q =fr∆f=f0f3dB(3.16)Where fr is the resonant frequency and ∆f is the resonant width, whichis the bandwidth over which the power of vibration is greater than halfpower at the resonant frequency. Under this definition, Q is the reciprocalof fractional bandwidth. The Figure 3.4 clarifies the formula to calculatethe Q-factor of a resonant circuit.Figure 3.4: Quality Factor of a resonant circuit. Copyright © electronics-notes.com. All rights reserved [73].When dealing with RF tuned circuits, there are many reasons why Qfactor is important. Usually, a high value of Q is beneficial, but in someapplications, a defined level of Q may be what is required. With increasingQ factor, the bandwidth of the tuned circuit is reduced. As the Q increases,the 3dB bandwidth decreases and the overall response of the tuned circuitincreases, and the losses decrease. So, the higher the resonator’s qualityfactor, the better the performance of the oscillator [74]. In many instances,263.1. Planar Microstrip Resonatora high Q factor is needed to ensure that the required degree of selectivity isachieved.3.1.3 Split ring resonatorThe split ring resonator (SRR) is merely a transmission line formed ina closed loop with a split. Resonator structures like the split ring resonator(SRR) are commonly used to shape metamaterials as building blocks [75–77].The SRR structure was first introduced by Pendry in 1999 [78]. Metamate-rial is an artificial material in which electromagnetic properties (permittiv-ity,  and permeability, µ) can be controlled. This is important because itmakes it possible for fields and waves to be controlled on a sub-wavelengthscale. In particular composite media, electromagnetic waves interact withthe inclusions, which produce electric and magnetic moments, affecting themacroscopic effective permittivity and permeability of the bulk compositemedium. Since metamaterials can be synthesized by embedding artificiallyfabricated inclusions in a specified host medium, this provides the designerwith an extensive collection of independent parameters such as properties ofhost materials, size, shape, and compositions of inclusions. All these designparameters play a significant role in getting the final result.Figure 3.5: A simple split ring resonator structure coupled with feedline.A simple split ring resonator structure is shown in Figure 3.5. The basiccircuit consists of the feed lines, coupling gap, and the split ring struc-273.1. Planar Microstrip Resonatorture. Energy is coupled into and out of the resonator through feed linesand coupling gaps. The coupling gap plays a vital role in achieving a goodresolution of the circuit. A larger coupling gap between the feedline andthe ring structure is considered a loose coupling, which manifests the neg-ligibly small capacitance of the coupling gap [79]. However, if the feedlineis pushed closer to the resonator, the coupling becomes tight, and the gapcapacitances are visible. This allows the circuit’s resonant frequencies todeviate from the ring’s intrinsic resonant frequencies [79]. Hence, design-ers need to consider the coupling gap effect based on the application whiledesigning the structure.Figure 3.6: (a) Resonator sensor structure implemented in HFSS for thepurpose of early detection breaches in pipeline coatings [57] and (b) opensplit ring resonator structure (OSRR) [80].Different configurations of split ring resonators were designed to per-form different applications. The basic configuration, which consists of a halfwavelength, open ring microstrip structure in Figure 3.6(a) is reported in[57]. The ring resonator was coupled to the input-output signal ports usingmicrostrip feed lines. The designed microwave resonance-based sensor wasused for detecting the cause of defects in pipelines, which is mainly drivenby coating failure. A double-ring resonator was proposed as the main com-ponent of getting real-time and hazard-free water quality monitoring in [81].The strong capacitance between the two concentric rings helps the flow ofcurrent along SRR configuration. S.S. Karthaikeyan et al. [80] designed283.1. Planar Microstrip Resonatoran open split ring resonator (OSRR) back in 2010, which has a geome-try of given kind in the Figure 3.6(b). The OSRR and wire arrangementconstructed the metamaterial cell. Scattering parameters determined thecomplex permittivity and permeability of the metamaterial cell. Dual-modesplit ring resonator was also designed in [82] to eliminate relative humidityimpact. One of the resonators resonates at 0.552 GHz, and another res-onator resonates at 1.03 GHz, and the latter one was used to calibrate thepotential measurement error.There are different geometries of SRR used to realize artificial magneticmaterials. In Figure 3.7, different geometries of SRR is shown to understandthe wide range of design ideas. It can also be seen from the Figure that thesplit ring resonator is not just only confined to the circular shape; instead,rectangular shapes of the ring are also found in different investigations. So,based on the application, designers have the flexibility to play with differentshapes and sizes of the resonators.Figure 3.7: Some geometries of split ring resonator (SRR). (a) Double splitring resonator, (b) Double split square resonator, (c) Singly split ring res-onator, (d) Two-turn spiral (circular) resonator, (e) Two-turn spiral (rect-angular) resonator and (f) Modified ring resonator. Copyright © Ali Kabiri2010. All rights reserved [83].293.2. Tuning Mechanisms3.2 Tuning MechanismsThe resonance of the constituent elements exhibits a narrowband fre-quency response in many electromagnetic devices, especially in metamate-rial applications [84]. In this case, materials are characterized or monitoredin one single resonant frequency. RF designers estimate this particular fre-quency based on the MUT’s behavior and utilize the design parameters todesign a compact resonant structure. This technique is beneficial in pro-viding higher precision in measurement but limits the measurement to onlyone single frequency. In order to broaden the frequency coverage, a popu-lar technique to resolve this issue is to add a tunable narrowband response.Several tuning methods have been suggested, and an analysis of these is pro-vided according to their overarching operating mechanism in the followingsection.3.2.1 External devicesIn many different investigations earlier, it was proposed to incorporateexternal devices such as capacitors, varactors, and micro-electromechanicalradio frequency systems (RF MEMS) into a resonator component to tune thefrequency response [84]. These variable components are often loaded withthe resonator structure, and by changing the values of the variable elements,the resonant frequency can be tuned [85, 86]. In Figure 3.8(a), a resonatorstructure is loaded with an external variable capacitor is presented. Thevariable capacitor element is placed in different positions such as the gapbetween two rings, the split of the outer ring and the split of the inner ring,and by changing the value of the capacitance, the change in the resonantfrequency is observed [85].Another way of tuning the resonant frequency is mounting varactordiodes across the gap of a resonator structure. The desired frequency tuningcan be achieve by adjusting the bias voltage of the varactor diode [87, 89–92]. This technique normally allows a large range of frequency tunability.In [87], Nesimoglu et al. utilized this technique on an S-shaped resonatorstructure, where 28% tunability was achieved by using reverse-biased varac-tor diodes at a critical location on the structure, the structure is presented inFigure 3.8(b). Zhu et al. [92] proposed a varactor loaded split ring resonatorantenna where the designed antenna’s operating frequency was tuned from365 MHz to 500 MHz by changing the bias voltage from 1 to 9 V. A relativefrequency change of 37% was observed during this investigation. However,Biasing circuitries are introduced into the structure by the presence of varac-303.2. Tuning MechanismsFigure 3.8: Tuning by the external devices. (a) SRR with capacitor loaded[85], (b) Varactor loaded S-shaped resonator [87] and (c) RF-MEMS loadedmicrostrip patch antenna [88]. Copyright © 2007 American Institute ofPhysics, 2007 and 2016 IEEE.tors in the resonator, resulting in additional losses/reflections/coupling [84].Besides, it will interfere with the RF signal and will restrict the resonator’sefficiency. Furthermore, this also enhances the complexity and scalabilitylimitations of manufacturing [84]. Ferroelectric varactors are found to main-tain a relatively low Q and temperature-sensitive and can create distortionin high Q-tunable filters [93–95]. This makes the varactors not suitable fordifferent environments.Components of RF-MEMS were integrated with EM systems to achievefrequency tunability in many applications [88, 96–99]. As shown in Figure3.8(c), Erdil et al. [88] presented a reconfigurable microstrip patch antennaintegrated with RF-MEMS capacitors for tuning the resonant frequency. Bychanging the actuation voltage from 0 - 11.9 V, the antenna resonant fre-313.2. Tuning Mechanismsquency can continuously be tuned from 16.05 GHz to 15.75 GHz. In [96],Bougye et al. proposed a combination of 4-pole bandstop filter and splitring resonator, which can digitally control the bandwidth due to frequencyshift by the on-off state of the RF-MEMS switches. By moving a cantileverRF-MEMS bridge capacitor up or down, a metamaterial filter is electro-statically actuated as in [97], causing a high-frequency tunability. AlthoughRF-MEMS switches provide very fast tuning speed, fabrication of the RF-MEMS is quite complex, especially while considering resonator arrays wherea huge number of RF-MEMS will be required [84].Other techniques were utilized for tuning the operating frequency, suchas semiconductor switches, p-i-n diodes, and Yttrium iron garnet (YIG).Semiconductor switches cannot handle much power, and it generates a lot ofdistortion [100–102]. P-I-N diodes need biasing for achieving good linearity,and as a result, it consumes high power [101]. YIG (Yttrium iron garnet)enabled switches can maintain a very high Q and relatively low distortion.However, due to high current consumption and magnet requirements, itcannot be integrated into planar fashion. Therefore, YIG is not appropriatefor low-cost portable solutions [101]. Due to the solid-solid contact in MEMSswitches, it creates contact wear and heat generation, which result in areliability issue [103–105].3.2.2 Microfluidic-based tuningThe appearance of microfluidics has led to the latest ongoing reputa-tion of utilizing microfluidic innovation for reconfigurable RF components.Ordinarily, RF component tunability is found to use either mechanical ro-tation, solid-state and varactor diodes, or RF MEMS, which experience theill effects of downsides of non-linearity, power losses, non-robustness, andfabrication complexity. The headway in microfluidics has as of late permit-ted researchers to defeat these disadvantages and execute RF devices. Bythe progress within the microfluidics fields, fluidic control can be utilizedto acknowledge highly linear devices and fluids/liquids to flow in channelsfit for dealing with high power RF inputs. Moreover, fluidic-based tuningdispenses with the plausibility of surface damage and can be utilized in flex-ible, wearable devices. Besides, liquid metal components are fit for holdingtheir physical changes subsequent to expelling the activation power, killingthe need for consistent actuation. Furthermore, it offers the simplicity offabrication contrasted with other tuning mechanisms.Table 3.1 shows a detailed analysis of different switching techniques avail-able in microwave technology.323.3. Review of Microfluidic ApplicationsTable 3.1: Comparison of different switching techniquesLiquidmetalSemiconductorswitchYIG MEMSTemperaturesensitivitylow low high lowTuning speed ms ns ms µsBiasing no need electric field magneticfieldelectricfieldOperatingfrequencyunlimited limited limited limitedCost low high high highLinearity linear non-linear non-linear linearMaintainingQ-factorpossible additional cir-cuitimpossible additionalcircuitPlanar yes yes no yes3.3 Review of Microfluidic ApplicationsAt current times, fluidic control for tuning microwave devices has pulledin much consideration attributable to the points of interest offered by fluidiccontrol. The fluidic control can be divided into two major parts based onabsorption and reflection switching techniques [106]. Both techniques havetheir merits and demerits. For high power applications, absorptive behavioris more important than the reflective switching technique since the reflectivesignal can adversely affect the circuit performance or lead to catastrophicfailures [107]. So, for low power applications, conventional reflective MEMSswitches are suitable.Broadband absorption can be achieved by injecting liquid metal [110],or water [111]. In microwave and sub-mm region, water is a relatively lossymaterial with a complex dielectric constant, and it requires careful consid-eration to avoid losses. On the other hand, fluidic based tunable devicestake part in the action and substitute the liquid elements with a type ofliquid metal to ensure adjustability. Initially, mercury has been constrainedto deformable body-wearable antennas [112], frequency selective surfaces,and metamaterials [113–115], since mercury is the only naturally occurringmetal that exists as a liquid at room temperature. Nevertheless, mercury333.3. Review of Microfluidic ApplicationsFigure 3.9: Microfluidic based tuning. (a) Implemented prototype of themicrofluidic waveguide switch [108], (b) Reconfigurable CPW folded slotantenna [109]. Copyright © 2015 and 2017 IEEE.was not favorable to improve the RF devices in later days due to its hightoxicity. Accordingly, inferable from their high conductivity, non-toxicity,and liquid state at room temperature, commercially available liquid metalalloys, for example, Eutectic Gallium-Indium (EGaIn) and Galinstan (com-position of Gallium, Indium and Tin) have pulled in much consideration formicrofluidic-based configurability.While some of the examples of such work include liquid metal as a switchto connect reactive loading elements to the circuit [108, 116, 117], liquidmetal itself as a variable reactive load [109, 118], use of liquid metal toconnect to short and open microstrip stubs [119]. In Figure 3.9(a), imple-mented prototype of the microfluidic waveguide switch is shown, which isdeveloped by Vahabisani et al. [108]. A slot antenna with a configurableresonant frequency by changing the length of a liquid metal bridge over thewaveguide structure is also reported in [109] and the implemented proto-type is shown in Figure 3.9(b). This approach did not enable continuous343.3. Review of Microfluidic Applicationsfrequency tuning but operated in multiple fixed modes based on the com-plete filling or emptying of the liquid metal channels. Frequency tuning oversuch a wide range can be utilized for frequency selective surfaces, blockingor transmitting signals in a controllable way [120], or developing tunable an-tennas [112]. A configurable microstrip open stub resonator was developedvia liquid metal to change the filter frequency [121]. It is worth mentioningthat, unlike the solid-state and most of the MEMS switches that consumepower all the time, the liquid metal could retain its position without theneed to apply DC power. This could reduce DC power utilization, particu-larly for the applications that the switch requires to remain in one state forquite a while.3.3.1 Challenges and limitations of microfluidic basedtuningMicrofluidic control over tuning a microwave circuit has its advantagesand disadvantages depending on the purpose of use. Since there are chal-lenges of employing fluidic control to achieve tunability, the challenges needto be addressed to better respond to the circuit. The choice of liquid metal isone of the very important selection considering its properties and microwaveperformance. Upon selection, it is vital to understand its behavior towardsthe employed fluidic channel and choose a suitable injection method. Beloware the three most significant challenges and limitations discussed to easethe use of liquid metal.Liquid selection:Table 3.2 shows the comparison among the physical properties of theavailable liquid metals with copper [122]. The only naturally available liq-uid metal mercury has a very low melting temperature. Due to highly toxicand hazardous to the health and environment, the use of mercury has beencompromised. On that note, low melting temperature and non-toxic behav-ior of liquid metal alloy, EGaIn and Galinstan make its place to substitutemercury. These liquid metal alloys not only avail high conductivity but alsostays liquid at room temperatures. In this thesis, Galinstan was used as liq-uid metal to incorporate the fluidic based tuning of the resonant frequency.Skin and residues:EGaIn and Galinstan also avail higher surface tension, which means itshows a high tendency to oxidize with oxygen whenever it comes in con-353.3. Review of Microfluidic ApplicationsTable 3.2: Comparison of physical properties of Copper, Galinstan, EGaInand Mercury © 2015 IEEEProperties Copper Galinstan EGaIn MercuryMelting point (°C) 1085 -19 15.5 -38.8Boiling point (°C) 2562 >1300 2000 357Density (kg/m3) 8960 6440 6280 1353DC conductivity (S/m) 5.96×107 3.45×106 3.4× 106 1.0× 106Viscosity (Pa.S) - 2.4×10−3 2.0×10−3 1.5×10−3Surface tension (N/m) - 0.718 0.624 0.487Figure 3.10: A liquid metal plug sandwiched between the carrier fluid insidethe microfluidic channel [123]. Copyright © 2014 IEEE.tact, bringing about the arrangement of a thin oxide film on the metalsurface [123]. The surface oxide layer made it regularly stick to the wallsof the tubing and the microchannel. While flowing liquid metal inside themicrochannel, this tends to leave skin residue and affect the device’s perfor-mance. Some carrier liquid is needed to eliminate the problem, which wouldclear the residue of liquid metal and can work as a potential barrier to liq-uid metal oxidation. The RF property, viscosity, and toxicity of the carrierliquid plays a vital role and have been studied [123, 124]. A strong acid orbase used as carrier liquid helps to remove the skin layer of the oxide [125].However, the use of such strong chemicals for many practical applications isundesirable. A safer alternative approach is to use either naphthenic base363.3. Review of Microfluidic Applicationsoil, Hydrocal 2400 [123, 126], and a low loss Teflon solution [127, 128]. In[123], the proper orientation of liquid metal inside the microchannel followedby the carrier liquid in both sides of the liquid metal is illustrated and shownhere in Figure 3.10. In this thesis, Hydrocal 2400 was used as a lubricator toeliminate skin residue of the oxide layer, and it also benefits as a potentialbarrier to liquid metal oxidation.Injection control:After the proper choice of liquid metal and the carrier liquid, it comesto pushing the liquid metal inside the microchannel. It is a straightforwardprocess but needs proper implementation following the steps. There aremultiple ways of injecting liquid metal through a microchannel. One way isto use a syringe and injecting the liquid metal inside the microchannel bycontrolling the pressure using hands. Initially, both the channel and tubeneed to be lubricated using the carrier liquid. The syringe should then befilled with the desired amount of liquid metal and pushed through the tubeinside the channel. Once the channel is filled with liquid metal, more carrierfluid should be pushed inside the tube. To encapsulate the liquid metal, onecan gently push or withdraw the carrier liquid by pushing in and out fromthe inlet or outlet side.Figure 3.11: (a) A mm syringe and (b) 11 Pico Plus Elite programmablesyringe to control the liquid flow [129]. Copyright © 2020 Darwin Microflu-idics.373.3. Review of Microfluidic ApplicationsAnother way is to ensure automatic injection and positional control overthe liquid metal inside the microchannel is to integrate the micro-pump [112]or electrochemical actuation [130]. In this thesis, liquid metal was pushedthrough the microchannel using an 11 Pico Plus Elite programmable syringeautomated pump from Harvard instruments. The standard infuse/withdrawmethod controlled the flow rate of liquid metal into the microchannel. InFigure 3.11(a), a mm syringe is shown which can be used manually to con-trol the flow of the liquid metal, and in Figure 3.11(b), an automated pro-grammable syringe from Harvard Instruments is shown to control the liquidmetal flow.3.3.2 Microfluidic ChannelMicrofluidic channels are utilized to manipulate and provide a closed en-vironment for the liquid sample exposed to electromagnetic fields. There aredifferent ways to implement microfluidic channels in microwave devices. Indifferent investigations, PDMS (Polydimethylsiloxane) is used to create themicrofluidic channel to flow different liquids. The microfluidics communityhas embraced PDMS for many reasons. It allows simple, planar systems tobe replicated and produced easily. Complex 3D structures and microchan-nel networks can be fabricated easily in PDMS by multilayer prototypingapproaches. A 3D printed microchannel embedded in PDMS is reported in[109] and [131] to realize a switchable dual-band slot antenna and recon-figurable CPW folded antenna, respectively. Fluid samples are injected tothe PDMS microchannel aligned with the resonator for dielectric charac-terization of the microfluidic samples [132]. Few more examples of recentwork include open stub resonator filter using pressure-responsive fluid in-side PDMS [121], PDMS structure including the microchannel is bonded toa filter to ensure reconfigurability of the SIW (substrate integrated waveg-uide) cavity filter [117], PDMS structure including one microchannel wasemployed to place the MUT to implement wideband microwave interferom-etry sensor [133]. Since PDMS does not break but bend, researchers found itmore useful compared to glass. There are always cautions trying things withglass as it is sharp and has disposal problems. Experts say that although thematerial is attractive for quickly testing the fluidics of the new devices, ithas problems since there is no perfect material. In Figure 3.12(a), a PDMSmicrochannel with the embedded tube is shown to transfer the liquid metalEGaIn.The fact that PDMS absorbs molecules is not a new discovery. The issuewith the PDMS includes absorption of organic solvents and small molecules,383.3. Review of Microfluidic ApplicationsFigure 3.12: (a) EGaIn plug inside a PDMS [123] and (b) 3D printed struc-ture utilized as a microfluidic channel [70]. Copyright © 2014 and 2018IEEE.its inherent hydrophobicity, and evaporation of water [134]. In some applica-tions, it is considered useful that PDMS absorbs the solvent and evaporateswater [134]. Since microwave devices are very much sensitive, even a smallalternation of the electromagnetic field can widely impact the device’s per-formance. PDMS is pressure-sensitive, making it hard to fabricate a largechannel using PDMS [134]. While flowing fluids, it needs constant pressurethroughout the channel. Pressure inequalities can create bending of thePDMS, which particularly can alter the response of the device. There isalways a risk of leakage due to channel-resonator bonding defects [135]. Anew approach not to substitute PDMS but can alternatively use for fluidmovement is 3D printing to overcome this problem. 3D printing is an emerg-ing technology and can be useful to be utilized as a microfluidic channel.In [70] and [136], a 3D printed microfluidic channel embedded with planarmicrowave resonator was realized for RFID and liquid detection. In Figure3.12(b), a 3D printed microchannel is presented [70]. In this thesis, a fluidicchannel was 3D printed and utilized to guide the fluids, allowing low cost,non-contact, sensitive, and easily reusable properties.393.4. Microwave Spectroscopy3.4 Microwave SpectroscopyMicrowave spectroscopy is the determination of those frequencies of themicrowave spectrum that are selectively absorbed by certain materials, pro-viding information about the atomic, molecular, and crystalline structure.This way, the complex dielectric properties (loss tangent and relative per-mittivity) of materials can be observed in a frequency spectrum. Since losstangent and relative permittivity are a function of frequency, the spectrumbrings out a clear picture of the material’s dielectric properties over a widerange of frequency.Figure 3.13: Dielectric spectra of soybean oil at 25°C temperature overa broadband where (a) real part and (b) imaginary part. Copyright ©Springer-Verlag London Limited 2011. All rights reserved [137].In microwave technology, dielectric property measurement can be broadlyidentified in two different methods such as resonant and non-resonant. Reso-nant methods are more reliable, accurate, and appropriate for characterizinglow-loss materials [138]. They can only provide measurement results in onesingle frequency, or a number of discrete frequencies [139]. The non-resonantmethods such as the TR line method, coaxial probe method, and free spacemethod, can enable material characterization over broadband; however, theyare less accurate than resonant methods [138]. Calibration errors, connectornon-reproducibility limit the accuracy of TR line methods, and mismatch-ing of impedance [140]. In [141], Narayanan et al. proposed permittivitymeasurement of the dielectric substrate over broadband utilizing microstriptransmission line method. The real (Figure 3.13a) and imaginary (Figure403.4. Microwave Spectroscopy3.13b) part of a dielectric spectra of soybean oil at 25°C temperature isshown in Figure 3.13 where author utilized the coaxial probe method tomeasure the complex dielectric property of soybean oil in different tem-perature setting [137]. The free space method limits the measurement ofdielectric properties of materials that are large and flat. It is best for flatsheets, powders and high temperature measurements [36, 41, 142].Resonant based spectroscopyIn case of a single frequency resonator sensor, the resonant frequency ofthe dominant mode highly depends on the material properties such that aminute change in the liquid can be monitored. However, in that case, thesensors do not provide any information on the material behavior across thefrequency and only act on the dominant mode and thus provide minimal in-formation about the material. Initially, investigations from the researchersfound multiple ways to employ a spectrum analysis of the materials. Har-monic based sampling of the permittivity spectrum is realized as a microwavedielectric spectrum to characterize material is reported in [139]. Authors in[143] have considered harmonics to extract the permittivity of dielectric slabsfor LTCC (low temperature co-fired ceramic) characterization.Figure 3.14: CPW folded slot antenna integrated with liquid metal toachieve three different frequency bands of 2.4, 3.5 and 5.8 GHz. Copyright© 2015 IEEE. All rights reserved [109].Microwave spectroscopy has been used for various purposes. Few exam-ples include the usage of microwave spectroscopy to discriminate between413.4. Microwave Spectroscopysub-populations of B lymphoma cell lines with a frequency of 40 MHz to 40GHz [144] and used a coplanar waveguide (CPW) transmission line for thedielectric characterization of HepG2 cells with a frequency range from 1 to40 GHz [145]. In [109], CPW folded slot antenna employing liquid-metal ascapacitive loading is realized to achieve reconfigurability, and three differentfrequency bands of 2.4, 3.5, and 5.8 GHz has been achieved by employingliquid metal at different positions. The response is shown in Figure 3.14.All these methods employ a discrete frequency spectrum where only a fewresonant points far from each other can be achieved. Frequency spectrumachieved by continuously tuning the resonant frequency enables more reso-nant points within a narrowband. In chapter 4, a modified planar microstripsplit ring resonator structure was designed and integrated with liquid metal,and 3D printed fluidic channel to achieve a continuous tunability of resonantfrequency. The continuous tuning of the operating frequency helps achievemore resonant points close to each other within a narrowband and providesa highly accurate resonant-based spectroscopy.The urgency for accurate identification increases as microwave detectionand sensing becomes more common. This is an ongoing challenge, and theunique ability of microwave sensors to sense and identify the dielectric prop-erties of solid, liquid or gaseous materials can be hindered when the materialunder test (MUT) is present in unknown or uncontrolled conditions. Thisissue has not been adequately addressed in many previous studies, whereliquid detection has been performed in controlled environments where theliquid and concentration are standardized [29, 146–149]. Similarly, selectiv-ity issues arise in other sensing operations. A larger material with a lowdielectric constant may affect the resonant response in the same way as asmall material with a high dielectric constant, leading to misidentification.One approach that can take advantage of the frequency dependency ofdielectric properties and avoid the overlap of dielectric properties is to takemultiple measurements of the same material and conditions at multiple fre-quencies, generating a unique spectrum of properties rather than a singulardata point. In this way, even mixtures that may have identical permittivityand loss tangent at one frequency can be separated by tuning the frequency.This idea of spectroscopy in the microwave frequency regime has been inves-tigated before [150, 151]: Albishi et al. [152] developed a multiple resonatorsystem to address the problem, but this required multiple resonators in alarger area, and measurements required a machine learning algorithm dueto complex coupling between them. The issue of selectivity has also beendealt with by developing new resonator designs and finding empirical algo-rithms to detect sample size and chemical makeup combinations. Hosseini423.4. Microwave Spectroscopyet al. [153] created a sensor that captures the concentration of mixtures byanalyzing multiple higher-order resonant modes, but this still measured atfixed frequencies and required wideband detection from 1.4 to 4.3 GHz.In this thesis, by combining the benefits of liquid metal, 3D printing,and planar microwave resonators, aiming to improve upon earlier designsto fully characterize the capabilities of a modified split ring resonator witha simple and compact design that enables continuous frequency tunablesensing and high selectivity for many types of materials. The developedresonant based spectroscopy technique is utilized to detect and distinguishbetween samples that exhibit similar dielectric properties in a frequency andpresented in chapter 4.43Chapter 4Adjustable MicrowaveResonatorThis chapter outlines the design and simulation process of the adjustablemicrowave resonator using a 3D printed fluidic channel and liquid metal. Inorder to verify the results from the simulation, a fabricated resonator hasbeen tested, and results have been compared through RF characterization.This work has been accepted for publication [154].4.1 Design and SimulationFigure 4.1 displays a pictorial diagram that explains the whole processstep by step from the design concept to result validation. The compari-son of results from structural simulation, equivalent circuit model, and theexperiment has verified the assessment of the whole operation.Figure 4.1: Pictorial depiction of the device development process.444.1. Design and Simulation4.1.1 Design methodology of the resonator structureSRR’s are small metallic rings or square loops with a split placed on topof a dielectric substrate. SRR’s are among the first microwave resonatorsfocused primarily on metamaterials whose geometries are smaller than thewavelength of exciting electromagnetic waves [155]. Up to now, extraor-dinary designs have been delivered within the discipline of metamaterials,and the most traditional ones are circular and rectangular split-ring res-onators. These have been investigated concerning distinctive performanceparameters, including sensitivity for sensing programs [156]. Rectangulardesigns are capable of imparting better sensitivity while they may be uti-lized for sensing applications [156]. Although, circular SRR has improvedbandwidth over rectangular SRR, rectangular structures are more suitablefor miniaturization and dense packing than circular structures [156, 157].A simple SRR gives maximum amplitude (resonant amplitude) at a fre-quency (resonant frequency) which can be determined by the geometry ofthe pattered microstrip lines on a substrate. The resonant frequency is dic-tated by the Equation 3.15. The formula itself describes that the higherthe resonator’s wavelength, the less the resonant frequency. The resonantfrequency (fr) can also be estimated in terms of capacitance and inductanceby the following formula.fr =12pi√LC(Hz) (4.1)Where C and L indicate the value of the capacitance and inductanceof the overall system, respectively. The reactive elements play a vital rolein producing a tunable resonance, as no modification can be made to themicrostrip structure after the fabrication of the resonator patch. To tunethe resonant frequency, reactive elements need to be changed significantly,which can be done by either external devices such as microfluidics. Manyconventional design ideas were simulated, and one of them is presented inFigure 4.2 with results to understand the operating principle.A simple SRR is presented in this design, and a channel filled with liquidmetal is added to the resonator consecutively to introduce a reactive loadto the device, as shown in Figure 4.2. It is realized that with each sub-sequent line of liquid metal added over the resonator, it provides anotherpath for the signal to travel. This causes the resonator to resonate at twoseparate frequencies, resulting in the Figure 4.2(f) presenting two resonantpeaks. One resonance occurs due to the signal transmission through thedesigned resonator structure and the other one is due to the added liquid454.1. Design and SimulationFigure 4.2: Simple SRR integrated with (a) no microfluidic channel (MC),(b) one MC, (c) two MC, (d) three MC and (e) four MC filled with liquidmetal, and (f) simulation result of different configurations of MC filled liquidmetal.metal filled channels. The resonant frequency decreases because of externalreactive loading added by the introduction of liquid metal filled channels.The discontinuity of the resonant frequency and less control over the fre-quency bandwidth makes this design not considered for tunable resonanceuse.The original intent was to design a single resonator structure. However,in the design and simulation stage, it became clear that this requirementmade it significantly more challenging to achieve our goals (specifically thewidely tunable bandwidth and consistent resonant amplitude) using a planarresonator’s constraints and a 2-port planar microfluidic channel. In fact,there are benefits to using a coupled-resonator design, such as the ability toperform differential sensing with one resonator acting as a reference and the464.1. Design and Simulationother as a material sensor [158].It is also believed that a single resonator approach should permit betterlinearity, sensitivity and perhaps enable liquid metal length to be extractedfrom either resonant amplitude or quality factor of the response. One ofour first design attempts at a tunable single resonator is shown in Figure4.3. The below Figure 4.3 shows that the continuity adjustment was notpossible for a single resonator structure. The additional length of the liquidmetal filled channel impacts on the wavelength of the resonator structure.However, our proposed design enables continuity because of the presence ofcoupling between the resonator’s.Figure 4.3: Frequency spectra of a simple SRR with varying liquid metalinside the fluidic channel.Instead of a single resonator, multiple resonators can be used by couplingthem electromagnetically [159]. Considering this idea, two resonator struc-tures of similar configurations are designed and coupled with each other. Asa result, a highly coupled resonant structure is developed. The initial intentwas to design a coupled-resonator structure because the coupling can be af-fected by the liquid metal to achieve frequency tuning. The geometry of theresonator consists of a half-wavelength, open ring microstrip structure. The474.1. Design and Simulationdesigned structure can be considered as a single resonator with two unequalsplits. The designed highly coupled modified resonator structure is shownin Figure 4.4 with dimensions. The corresponding values of the resonatorstructure are given in the table 4.1. The geometry of the structure wasdesigned primarily through simulation. The structure of the design can bemanipulated based on user’s interest.Figure 4.4: Designed highly coupled modified SRR structure with dimen-sions.Table 4.1: Design parameters of the designed modified resonator structureSymbol a b g s W1 W2 dValue (mm) 9.75 8.00 2.00 1.50 2.00 1.50 1.00The designed resonator structure was coupled with a 50 Ω transmissionline where the width (W1) of the transmission line was calculated using thefollowing equation:W1 = 7.48× h× e−(Zo√r+1.4187)(m) (4.2)Where Zo, h, and r define the characteristic impedance of the trans-mission line, substrate height and the relative permittivity of the dielectric,respectively. The splits in the structure were chosen and set to (s=) 1.5484.1. Design and Simulationmm and (g=) 2.0 mm based on the simulation results and are presentedin section 4.1.3. The simulation dictated that these should be the valuesselected.4.1.2 Design methodology of the fluidic channelAs discussed in section 3.3.2, microfluidic channels have become veryprominent for guiding fluids in different applications within the microwavetechnology realm. The appearance of 3D printing has seen a rapid increasein the manufacturing and fabrication technology of 3D structures [160]. Ithas garnered a high degree of acceptance among researchers as a promisingfabrication alternative. One of the most significant advantages of 3D print-ing technology is rapid prototyping. It is the ability to design, manufacture,and test a customized part in as little time and cost as possible. Addition-ally, the design can be modified without adversely affecting the speed ofthe fabrication process. Although it suffers from low resolution and highequipment cost, high-end 3D printing machines can improve the resolutionimmensely.The structure, geometry and dimensions of the fluidic channel corre-spond with the microwave resonator. The design is created in Solidworksdesign software. The designed microfluidic channel was aligned with the res-onator covering the gap (g) in the XY plane. In Figure 4.5, the cross-sectionof the designed channel is presented along with the channel width, length,and height. Channel height (h) and width (w) are chosen to 1.5 and 2.0mm, respectively. The length (L) of the channel is 3 cm, and the channelinterior is separated (d1=) 50 µm from the structure to allow non-contactmeasurement. Threaded connectors were designed in both the inlet andoutlet ports to ensure leak proof connection with the tubes.The width and height of the microchannel play a vital role in achievinga wideband tuning. The longer length (L) of the channel filled with liquidmetal introduces more loss by allowing the signal to transmit to the groundand resulting in attenuation of the signal. Therefore, the length of themicrochannel was chosen to be 3 cm depending on the arm length of thedesigned resonator structure.The microchannel dimensions were optimized through structural simula-tion. It is realized from the structural simulation that increasing the heightof the microchannel did not impact the resonant profile of the designed res-onator significantly as it does not impact the width or the distance betweenthe two parallel plates (liquid metal filled channel and the resonator). Sim-ulations were performed with different values of w (width) for liquid metal494.1. Design and SimulationFigure 4.5: (a) The cross-section of the designed fluidic channel, (b) channelheight (h=) 1.5 mm, width (w=) 2.0 mm and channel separation (d1=) 50µm and (c) channel length (L=) 3 cm.amount of 20 mm, 26 mm and 32 mm; the results are included here withinFigure 4.6. Increasing the width of the channel slightly varies the frequencyand impacts the quality factor. By varying the width of the microchannel, itaffects the wavelength and enables a shorter path to the signal to transmit.Hence, lower wavelengths increase the frequency and decreases the qualityfactor.Figure 4.6 shows the variation of resonant frequency in different widthswhen the liquid metal has filled the channel by 32 mm, 26 mm, and 20 mm.The widths are chosen to simulate 1.5mm, 2.5mm, 3.5mm, and 4.5mm.504.1. Design and SimulationIt can be depicted from the simulation result that the higher the width,the more space it covers, and it creates an additional capacitance to theground, which allows more signal to transmit to the ground. In summary,the microchannel dimensions (L,w,h) may vary depending on the resonatorstructure and the application. Based on the simulation results, the length,width and height of the fluidic channel are set to 3 cm, 2 µm and 1.5 µm,respectively.Figure 4.6: Resonant frequency shift at different widths when liquid metallength is 32mm, 26mm and 20mm.4.1.3 Structural simulationFor simulation and design purposes, Ansys HFSS (High Frequency Struc-ture Simulation) is used. The resonator patch is made of a layer of copperthat is 35 µm in thickness. The structure is coupled to the input and outputsignal ports using microstrip feedlines. The resonator structure is imple-mented on an RT/duroid 5880 dielectric substrate from Rogers, which has adielectric constant and loss tangent of 2.20 ± 0.02 and 0.0009, respectively.The dielectric substrate has a thickness of 0.79 mm. The substrate’s bottom514.1. Design and Simulationlayer is covered with 35 µm thick copper as the ground layer. The designedfluidic channel is integrated with the resonator structure.To replicate a realistic scenario, a co-polyester with a permittivity of 2.1and loss tangent 0.008 represents the channel. The material that symbolizesliquid metal in the simulation is made of the actual characteristics of liquidmetal in real. Simulations used the HFSS software program with a mess sizeof λ/10; the whole layout is covered with a vacuum airbox with radiatingboundary conditions. The input-output feedlines are connected to portswith a full port impedance of 50 Ω. A parametric analysis was performed toanalyze the frequency response for a different amount of liquid metal insidethe microchannel. Frequency sweep from 2 GHz to 4 GHz with 2001 pointswas performed initially.Figure 4.7 shows the layout of the designed resonator integrated with thedesigned fluidic channel. The resonator patch and the co-polyester channelare separated by 0.1 mm to replicate a realistic scenario. There will al-ways be a negligible airgap between any bonding. The microchannel is alsoseparated by 0.6 mm from the resonator patch to allow for non-contact res-onance. Figure 4.8 shows different levels of separations to reflect a practicalscenario.Figure 4.7: Layout of the designed resonator integrated with fluidic channel.Initially, the simulation was performed for different gap sizes to under-stand the impact of gap size on the designed resonator structure. In orderto do that, the split, “s” was kept 1.5 mm, and the gap, “g” was variedfrom 1.5 to 4.0 mm. The transmission spectra (S21) for different gap sizes524.1. Design and SimulationFigure 4.8: Cross-sectional sideview of the designed resonator integratedwith fluidic channel.is presented in Figure 4.9(a). The result displays one resonant frequencywhen the gap sizes are equal. However, the structure exhibits two resonantpeaks close to each other for unequal sizes of gaps. It is realized from furtherinvestigation that equal gap sizes limit the range of frequency tuning. As aresult, the final structure was designed with a gap size of 2 mm to allow asufficient coupling gap between two resonant structures.Figure 4.9: Transmission spectra (S21) of the designed resonator structurefor (a) different gap sizes, g and (b) different coupling distances, d.534.1. Design and SimulationAnother critical parameter that contributes to the performance of theresonator is the coupling distance (d) between the ring and the transmissionline. The losses would be smaller if a relatively small gap is used, but thefields in the resonant system will also be more greatly affected. A widergap results in less field interference but greater losses [79]. The impactof different coupling distances were analyzed through structural simulationpresented in Figure 4.9(b) and realized that the greater the coupling dis-tance, the greater the effect on the resolution (quality factor) of the ring’sperformance. Considering the effect on the resolution, the effective couplingarea was increased, and the coupling distance was kept at 1 mm for optimumperformance of the designed structure.Figure 4.10: Transmission spectra (S21) of the designed resonator structure.The final design contains two unequal splits of 1.5 mm and 2.0 mmin an SRR, and the coupling gap between the feedlines and the resonatorstructure was chosen to be 1 mm. The transmission spectra (S21) of thedesigned modified resonator structure with an empty fluidic channel is pre-sented in the Figure 4.10. It exhibits two resonant frequencies of 3.52 and3.57 GHz with a resonant amplitude of -24.06 and -23.00 dB, respectively.The proposed resonant structure consists of two rings electromagneticallycoupled with each other. Electromagnetic coupling utilizes both the electric544.1. Design and Simulationand magnetic coupling between the rings. At one resonance, the electriccoupling is dominant, and at another peak, magnetic coupling is dominant.The overall response utilizes a superposition of both the couplings, whichreflects in the transmission spectra having two resonant peaks close to eachother.Figure 4.11: EM field distribution plot of the designed resonator structurefrom HFSS at a resonance of 3.52 GHz.The EM field distribution is a necessary investigation when dealing withmicrowave circuits. It helps to observe the electric and magnetic field in-tensity in different areas of the structure. EM field distribution of the de-signed structure was performed in HFSS at one resonance (3.52 GHz) andpresented in Figure 4.11. The electric field distribution was performed indifferent planes of the structure. It is realized from the field distribution554.2. Fabrication and Experimental Setupplot that there is a high electric field intensity in the gaps of the designedstructure, and the intensity is not only limited to the XY plane, but alsocontributes to XZ and YZ plane.From Figure 4.11, transmission feedlines and the resonator structure aremagnetically coupled with each other. The magnetic coupling between theresonator and the feedlines are observed to be strongest when the couplingdistance is 1 mm. There is no contribution of the magnetic field between thering structures, which means only the electric field contributes to achievingthe resonance of 3.52 GHz. It is evident that any material in the ring gapwould affect the structure’s overall permittivity, and a change in the resonantfrequency will be observed.4.2 Fabrication and Experimental SetupThis section will describe fabrication of the microwave setup as wellas how it was utilized in the experimental setup. The measured resultsare compared with the structure simulation afterward to validate the ideaand concept. The designed resonator structure was fabricated using an in-house fabrication process, and the designed fluidic channel was 3D printedand outsourced. After the fabrication, elements were needed to connectaccordingly to perform the experiment and collect the datasets.4.2.1 Fabrication of highly coupled resonator structureFigure 4.12 shows the fabricated resonator on top of a high-frequencysubstrate whose surface was covered with a thin layer of copper as thegrounding material. In this thesis, Rogers RT/duroid 5880 high-frequencysubstrate was used to fabricate the designed highly coupled resonator struc-ture and the input-output lines. The thickness and dielectric constant of thehigh-frequency substrate were 0.79 mm and 2.20 ± 0.02, respectively, witha loss tangent of 0.0009.Rogers 5880 substrate has the advantages of low electrical signal loss,effective-cost, lower dielectric loss, better thermal management and im-proved impedance control. A layer of copper was covered the surface ofthe substrate with a thickness of 35 µm. The copper was selectively pat-terned to layout the micro-strip lines for the resonator. SMA (SubMiniatureversion A) connectors are soldered with the input-output line to facilitatethe connections with the measurement device.564.2. Fabrication and Experimental SetupFigure 4.12: Fabricated resonator structure on Rogers RT/duroid 5880 high-frequency substrate and SMA connectors are soldered with the feedlines.4.2.2 3D printed fluidic channelFigure 4.13(a) shows the fabricated PDMS microchannel and Figure4.13(b) shows the fabricated PDMS channel integrated with the designedresonator structure. A PDMS microchannel was initially designed based onthe fabricated resonator structure to guide the liquid metal. Due to the highviscosity of liquid metal, the fabricated PDMS channel suffered mechanicalexpansion and shifting, and the resonant response was affected. The highpressure of liquid metal also consistently created leakages in the PDMSchannels, specifically at the inlets and outlets of the fluidic channel. As aresult, an alternative approach of 3D printed microchannel was consideredto facilitate the liquid metal flow.The designed fluidic channel was outsourced, and a co-polyester NGENcolorFabb was 3D printed, which has the permittivity and loss tangent of2.1 and 0.008, respectively. The channel was mechanically fixed with theresonator using plastic nuts and bolts. This allows more electrical stabilityto attain higher precision and improved sensitivity. The channel was sepa-rated from the resonator by 50 µm to allow no contact with the resonator.574.2. Fabrication and Experimental SetupFigure 4.13: (a) Fabricated PDMS microchannel and (b) the designed res-onator structure integrated with the PDMS microchannel.Figure 4.14: Fabricated resonator integrated with 3D printed fluidic channel.Inlet and outlet tubes were attached to the microchannel to allow the liq-uid metal to transfer. Figure 4.14 shows the fabricated resonator integratedwith the 3D printed fluidic channel. The length of the channel is chosen584.2. Fabrication and Experimental Setup3 cm based on the designed resonator structure. Microchannel width andheight are chosen 2 mm and 1.5 mm, respectively. The 3D printed channelwas cross-sectioned, and the surface profile of the embedded channel wastested through a profilometry experiment.Figure 4.15: Surface profile of the 3D printed fluidic channel along the lengthA to C.The surface profile of the 3D printed fluidic channel was inspected througha Dektak XT Bruker Stylus Profiler. Initially, the experiment was performedto measure the surface profile along the length A to C as indicated in Figure4.15, and the measured result is presented in Figure 4.15. The displayedresult indicated three different color-coded regions where “B” represents theembedded channel profile with an average surface roughness of ≈2 microns.Additionally, the measured average surface roughness along the length Aand C were ≈9.58 and ≈0.5 microns, respectively. The surface profile acrossthe embedded fluidic channel was also tested, and the result is shown inFigure 4.16. The result shows an average surface roughness of ≈7 microns.Irregularity on the surface can create surface friction resulting in liquid metalresidues on the surface. This can immensely affect the performance of theresonator. During the experiment, the presence of some liquid metal residueswas found inside the fluidic channel. To eliminate this issue, liquid metal wasinjected and moved back and forth to collect the residues. Hence, the fluidicchannel shows excellent performance and perfectly performs fluid movementwithout adversely affecting the resonator’s performance.594.3. RF CharacterizationFigure 4.16: Surface profile of the embedded channel inside the 3D printedstructure.4.2.3 Experimental setupThe fabricated highly coupled resonator structure and the 3D-printedchannel are integrated using plastic nuts and bolts for mechanical stability.Inlet and outlet tubes are used for the injection and subsequent withdrawalof liquid metal. In order to have control over the injection of liquid metal,11 Pico Plus Elite programmable syringe pump from Harvard instrumentswas used. The standard infuse/withdraw method controlled the flow rateof liquid metal into the microchannel. The input and output ports wereconnected to port 1 and port 2 of the vector network analyzer (VNA) toallow the signal transmission. The experimental setup was surrounded bymicrowave absorbing foam (AN79) to limit mechanical and electromagneticnoise from entering the system. The experimental setup is shown in Figure4.17 to understand the overall setup.4.3 RF CharacterizationIn this section, results from different platforms such as HFSS, ADS, andexperiments are analyzed and compared. Structure simulation results andmeasured results were found in very good agreement. Few other parametersare also analyzed and will be presented in this section.604.3. RF CharacterizationFigure 4.17: Experimental setup used to measure the S21 response of thetunable resonator. The fabricated resonator filled with liquid metal andmicrochannel implementation is shown in the top-left inset.4.3.1 Scattering (s) parameter measurementThe scattering parameters of the structure simulation were performedusing a parametric sweep for the length of liquid metal from 10 mm to 28mm, which covers the whole microchannel. The simulation was performedfor a linear count of frequency from 2 GHz to 4 GHz with 2001 equivalentpoints. The result showed in Figure 4.18 describes the change in resonantfrequency (f ) in GHz and the amplitude (S21) in dB. When the liquid metalstarted to cross the gap (12 mm), it was found that the resonant frequencydecreases with a fairly constant amplitude and quality factor. When theliquid metal length is less than 12 mm, it does not impact the resonantfrequency because it does not alter the resonator’s coupling until it crossesthe gap. The Figure displays the resonant frequency decrease with theincursion of liquid metal.The transmitted signal (S21) was measured with a Fieldfox Vector Net-work Analyzer (VNA). The VNA’s output power and intermediate frequencybandwidth were set to 0 dB and 100 kHz, respectively, and the measure-ments were accrued over 2001 evenly distributed frequency points. Each614.3. RF CharacterizationFigure 4.18: Frequency spectra from structure simulation for different liquidmetal amounts. Few of the instances of liquid metal inside the channel isshown on the left side.dataset has been collected a minimum of five times for consistent data anda better understanding of system errors. The unloaded resonator had centerfrequencies at 3.57 GHz and 3.52 GHz with resonant amplitudes of -23.00and -24.06 dB, respectively. These resonant responses are highly dependenton the length of liquid metal present in the microfluidic channel.To tune the resonator, Galinstan (liquid metal) was pushed using thepump from Harvard instruments. The standard infuse/withdraw methodcontrolled the flow rate of liquid metal into the micro-channel. One issue wasthat Galinstan’s high surface tension made it regularly stick to the walls ofthe tubing and the microchannel. To eliminate this problem, Hydrocal 2400oil initially flowed through the tube and the microchannel as a lubricant,and it also benefited as a potential barrier to liquid metal oxidation. Themicrofluidic channel was filled to varying levels with liquid metal to tunethe circuit’s resonant response subsequently. This technique utilized thereactive loading effect of the liquid metal on the ring resonator structure.As a result of merely increasing the overall circuit’s reactive loading, thefrequency of resonator decreases in all simulations and was also recorded inexperiments.624.3. RF CharacterizationTo investigate the effectiveness of the fabricated microwave circuit as atunable resonator, the S21 response was measured while varying the vol-ume of liquid metal injected in the channel using the pump. The frequencyspectra for different liquid metal amounts were measured during the ex-periment. The length of the liquid metal analyzed during the experimentand Figure 4.19 shows that a full channel length (28 mm) of liquid metalgives the maximum frequency shift. However, it may be possible to achievemore frequency shift by fabricating a longer fluidic channel depending onthe resonator structure.Figure 4.19: Resonant S21 spectra recorded by the VNA as the volume ofmetal inside the channel is varied. Few of the instances of liquid metal insidethe channel are shown on the left side.Without any external reactive loading, the resonator introduces twopeaks very close to each other. By varying the reactive loading using theliquid metal, we can increase the frequency shift while also keeping the am-plitude variation ± 3 dB. A change in resonant frequency from 2.35 to 3.4GHz (relative change: 45%) was measured during the experiment. The aver-age sensitivity of the resonator as a function of liquid metal length was foundto be 375 MHz/cm. Additionally, one gap of the ring resonator was placedelectrically far away from the liquid metal, potentially leaving it available634.3. RF Characterizationto act as a sensor for dielectric materials.4.3.2 Equivalent circuit modelingThe LC circuit modeling is observed to be very useful to analyze andcontrol the behavior of the EM structure for final assessment. The equivalentcircuit model is designed based on the results of the simulation. AgilentAdvanced design simulation (ADS) software was used to design and evaluatethe equivalent circuit model results.Considering the modified SRR to be mutually coupled with the input andoutput (i/o) microstrip line, the i/o lines were modeled by mutual inductancebetween the resonator patch and the i/o line. Each of the resonators arerepresented by a R-L-C combination where R, L and C refers resistance,inductance and capacitance respectively. Both the resonators are electricallycoupled with each other. The liquid metal acts as a single stub and enhancesthe reactive loading by coupling with the resonator patch.Figure 4.20: Equivalent circuit model overlaying the substrate, resonatorand liquid metal.In Figure 4.20, the lumped element equivalent circuit model overlayingthe substrate, resonator, and liquid metal is presented. Lr, Cg and Rrrepresents the inductance of the line, the capacitance between resonatorpatch and ground, and resistance of the line, respectively. To consider theloss in line and substrate, Rsub and Rr are modeled. The electrical couplingbetween the resonators was modeled by capacitance Cr and Cr1. The liquidmetal stub is modeled by a series LC circuit connection, which is representedby C1, L1 and C2. In this case, the values are variable for LC combination644.3. RF Characterizationbecause the liquid metal length defines these reactive elements’ values. Intable 4.2, the calculated values of the corresponding elements are presented.Table 4.2: Defining the values of the parameters of the equivalent circuitmodel based on the simulation results.Symbol Value Unit Symbol Value UnitRr 5.0 Ω C1 5-C2 pFLr 31.485 nH C2 0.28-2.90 pFCr 0.0357 pF L1 9.60-30.21 nHCr1 0.031 pF Cg 0.02 pFT 7.66 Rsub 54.3 ΩWithout the liquid metal stub, the calculated values are fixed for theresonator. Incursion of liquid metal enables external reactive elements C1,L1 and C2 to contribute to the resonator profile. Increasing the liquid metalvolume increases the capacitances and inductance. When liquid metal is veryclose to the gap, the resonant profile begins to be impacted. Since liquidmetal before the gap does not impact the coupling between the resonators,the resonant frequency change is negligible. Liquid metal acts like a bridgethat connects both the resonators and impacts the coupling. When liquidmetal starts crossing the gap, the resonant frequency starts to decrease witha consistent amplitude and quality factor as characterized in the previoussection.4.3.3 Results comparisonThe designed resonator structure had a tuning range of 1.04 GHz, ad-justing between 2.48 to 3.52 GHz. Structure simulation and equivalent cir-cuit modeling of the designed channel with and without liquid metal in thechannel were performed, and results were compared with the experimentin Figure 4.21. Figure 4.21(a) shows the empty channel results and Figure4.21(b) shows the filled channel results. The results are similar but variedslightly, possibly due to the lack of a standard liquid metal in the literatureor oxidation of the Galinstan upon exposure to oxygen in the channel duringthe experiments.654.3. RF CharacterizationFigure 4.21: Comparison between the simulated (HFSS and ADS) and mea-sured S21 frequency spectra for the resonator with the fluidic channel in a(a) empty state and (b) filled state.Figure 4.22: Resonant frequencies recorded for different metal lengths withselected images included.664.4. Chapter summaryFigure 4.22 presents the trend between liquid metal length and resonantfrequency shift found from both experiment and simulations. The blue crossdefines the values from the structure simulation, and the red circle defines themeasured responses from the experiment. Resonant frequency is droppingwith liquid metal length because of additional reactive loading added to theresonator profile. The measured value confirms simulation as both are ingood agreement. Therefore simulation is validated.4.4 Chapter summaryA tunable microwave resonant structure was created with a resonantfrequency that could be adjusted between 2.35 and 3.40 GHz by using thelogical capability of liquid metal as an adjustable capacitive source. Controlof the resonant frequency was conceivable by pushing various volumes ofGalinstan through a 3D printed microchannel situated over the resonator.A resonant frequency shift of 1.05 GHz with a quality factor of 70 (±5)was experimentally measured with insignificant (<3 dB) variety in resonantamplitude. The designed structure has a tunable frequency, empoweringfrequency-specific connections between resonator and sample using a singleresonator. This plan can be effectively executed to many frequency rangesas a microwave filter or sensor and holds potential in the field of microwavefrequency spectroscopy.67Chapter 5Resonant BasedSpectroscopy for multi-phasematerial detectionIn this chapter, resonant based spectroscopy has been utilized to differ-entiate between the samples of similar dielectric properties. The discussionstarts by stating the problem through the Debye model, followed by the de-sign and simulation process of the microwave resonant sensor. The designedsensor utilizes liquid metal to generate spectroscopic data for multi-phasematerial detection. Experiments have been performed to characterize andvalidate the designed sensor. The presented results are the collective presen-tation of more than adequate simulations and measurement efforts that havesometimes been unsuccessful due to proper presentation and experimentalclarification.5.1 Problem statementMany studies have focused on sensing materials using microwave res-onator technology, and by using these resonator sensors, materials with dis-similar properties can easily be identified, and distinguished [26, 161]. Theproblem arises when the resonant properties of different materials are thesame at specific frequencies due to a combination of their size, shape, orcomposition. Introducing a continuously tunable frequency spectrum usinga resonant-based structure could be a possible solution to eliminate this issuebecause of the many factors in a material’s response are affected differentlyby frequency variation.To understand this problem in a microwave resonator sensor, two genericmaterials, A and B, were created, and the dielectric properties of thesematerials were analyzed with the Debye model. Debye model is used for theMUT’s to consider the frequency dependence of the dielectric properties.According to the Debye model, the real (′) and imaginary part (′′) of the685.1. Problem statementcomplex dielectric permittivity are calculated using the following equation:′ = ∞ +s − ∞1 + ω2τ2(5.1)′′ =(s − ∞)ωτ1 + ω2τ2(5.2)Where s and ∞ define the permittivity at low frequency and high fre-quency, respectively. Operating angular frequency and the dielectric relax-ation time are defined by ω and τ , respectively.Figure 5.1: The calculated dielectric complex permittivity of material Aand material B at 23.7°C modelled by Debye method. The parameters formaterial A were set to match water: s=76.58 ∞=4.73 τ=9.3 ps. MaterialB was defined by s=87.37 ∞=68.13 τ=90.9 ps.The designed materials are modeled using the Debye formulation, andthe real and imaginary part of the permittivity was calculated and pre-sented in Figure 5.1. As shown in Figure 5.1, the dielectric permittivity andloss factor for both materials (A and B) were found to intersect at 2.258GHz frequency, which means both materials exhibit identical dielectric be-havior at that frequency. This generates a critical problem for resonantbased methods in terms of sensing and characterizing the materials using695.2. Operating principle of the designed resonator sensorresonant-based structure. Any microwave resonator sensor operating at thisfrequency is unable to distinguish between the materials. However, uponreferencing the Debye model, deviating from the intercept point introducesthe possibility of detecting different values of dielectric permittivity of thematerials. Therefore, a resonator is demanded whose resonant frequencycould be adjusted to avoid the intercept.5.2 Operating principle of the designed resonatorsensorThe previous design of the resonator structure presented in chapter 4was modified by integrating an interdigitated capacitive sensor. Many dif-ferent types of sensors structure are available in microwave technology, andmost of the sensors often work on sensing dielectric properties. The resis-tive and the capacitive techniques are immensely used for the fabricationof the microwave sensors. The capacitive sensor provides low-temperaturedependence, miniaturization potential, small power consumption with possi-bilities of sensing a minimal variation in the materials properties [162–166].Different capacitive structures are possible for the fabrication of the sensor,such as parallel plates, co-axial cylinders, fringing fields [162]. However,among them the co-planar interdigitated capacitive sensor is most widelyused in the microwave technology. Interdigitated capacitive sensors are anon-destructive method of studying material properties such as conductiv-ity, permittivity, and permeability [167].The interdigitated capacitive sensor is implemented in this design tosense the MUT within a large, highly sensitive area [167]. The impact ofthe MUT in the sensitive region can be monitored via the effective dielectricproperty variation in that sensitive region [168]. The large sensitive regionattained by the interdigitated structure enabled sensitive monitoring of per-mittivity and conductivity variation over a larger area [169]. The operationof the interdigitated capacitive sensor is similar to a parallel plate capacitor[162]. The interdigitated capacitive structure is designed to provide highcapacitance when coupled to the resonator, increasing frequency tunability.The design also provides a large region with a high electromagnetic field tohelp identify the MUT. The electric field travels around the resonator andacross the capacitor by penetrating the MUT, which changes the impedanceof the sensor. Therefore, by measuring the changing S21 transmission ofthe sensor, the MUT properties can be evaluated over time. The relationbetween the effective dielectric permittivity and the resonant frequency is705.3. Design methodology and simulationpresented in equation 3.15.Tuning of the resonant frequency follows the similar principle of oper-ation discussed in chapter 4. Liquid metal inside the fluidic channel con-tributes to the resonant frequency tuning by impacting the coupling betweenthe ring structures. The change in the relative permittivity of the resonantprofile due to the presence of the MUT alters the resonant frequency of theresonator, and by calculating the frequency shift, materials can be charac-terized [26].5.3 Design methodology and simulation5.3.1 Structure design and tunable operationFigure 5.2: Structural design of the proposed resonator sensor with dimen-sions.The sensor structure consists of a highly coupled modified split ringresonator integrated with an interdigitated capacitive sensor presented inFigure 5.2. The resonator loop consists of two gaps (gap 1 and gap 2),which made the rings highly electromagnetically coupled. This was furtherimproved by an interdigitated capacitive sensor region for material sensing.The planar structure has resonant frequencies determined by the size andshape of the designed microstrip lines. The modified structure allows theresonator to exhibit two resonant peaks close to each other due to two un-equal gaps in the ring structure. This was optimized in the previous channel.715.3. Design methodology and simulationThe dimensions of the sensor structure are shown in table 5.1.Table 5.1: Defining the values of the parameters of the designed sensorSymbol Value (mm) Symbol Value (mm)l1 8.00 w2 1.50l2 9.75 w3 1.00g1 1.50 w4 0.40g2 2.00 s 1.00w1 2.00 d 1.00Figure 5.3: S21 response of the unloaded resonator from the HFSS simulationsoftware.The resonator sensor structure was implemented in ANSYS High Fre-quency Structure Simulation (HFSS) software to study the liquid metal im-pact on the resonator and material testing. The S21 transmission of theunloaded resonator from the HFSS simulation is presented in Figure 5.3.The unloaded resonator resonates at 2.57 GHz with -14.84 dB and 2.75GHz with -15.85 dB of amplitude based on the simulation result. The fielddistribution of the designed structure is also analyzed in the HFSS platform.725.3. Design methodology and simulationAccording to the field distribution plot, the higher resonance (fH) occursat 2.75 GHz due to the magnetic field coupling between the resonator andthe sensor, and the lower resonance (fL) occurs at 2.57 GHz due to theelectrical field coupling between the resonators. The field distribution plotfrom the HFSS simulation is presented in Figure 5.4. Due to high electricfield intensity in the interdigitated capacitive area shown in Fig. 5.4, thisregion is used in this design to sense the MUT over a large, highly sensitivearea.Figure 5.4: Field distribution plot (a) Electric field distribution at 2.57 GHzand (b) Magnetic field distribution at 2.75 GHz.The designed resonator structure shown in Figure 5.2 was integratedwith liquid metal controlled microfluidic channel placed above gap 2. Thegap location and size relate strongly to the resonant frequency supportedand can be covered with liquid metal to increase the capacitive loadingof the resonator and decrease the resonant frequency [154]. Liquid metalwas injected in different volumes inside the microchannel to perform thefrequency tuning. To manage the controlled flow of liquid metal, the lengthof the channel was measured at regular 3.5 mm intervals from 18-32 mm. Inpractice, tuning could be done with any amount of liquid metal to target anyspecific frequency. According to the simulation results, a frequency tuningof 250 MHz with constant amplitude and quality factor was achieved. The735.3. Design methodology and simulationlayout of the sensor structure and the simulated result from HFSS whenthe channel is filled with liquid metal in the length of 18-32 mm of thefluidic channel are presented in Figure 5.5a and Figure 5.5b, respectively.Additionally, the change in resonant frequency and resonant amplitude withvarying liquid metal length are presented in Figure 5.5c and Figure 5.5d,respectively.Figure 5.5: (a) Layout of the sensor structure from the HFSS software, (b)simulations results from HFSS when the channel is filled with liquid metalin the length of 18-32 mm of microfluidic channel referenced with unloadedresonator response, (c) change in resonant frequency and (d) resonant am-plitude with varying length of liquid metal.745.3. Design methodology and simulation5.3.2 Sensitivity analysis of the designed sensorDifferent permittivity standard solid materialsThe sensitivity of the adjustable sensor was investigated through HFSSsimulation by changing the amount of liquid metal volume inside the micro-channel and placing materials with different permittivity on the sensingarea. The testing material was placed over the microstrip lines due to highelectric field intensity in the interdigitated capacitance region. MUTs ofpermittivity 4.3, 5.7, 6.8, 8.3, 10.0, 12.9, and 16.5 were used to analyze thesensing capability for different amounts of liquid metal inside the microchan-nel. To perform this operation, the microfluidic channel was filled 25mm inlength with liquid metal, and standard materials with different permittivitywere placed individually in the sensing area. Each material demonstratesa unique response based on its permittivity. The higher permittivity ma-terials have higher frequency shifts, and the lower permittivity materialshave lower frequency shifts due to their expected interaction with the pen-etrating electromagnetic field. Materials with higher permittivity interactmore strongly with the electric field compared to the material with lowerpermittivity [170].Figure 5.6: (a) Simulations results from HFSS when liquid metal is filled inthe length of 25 mm of the microfluidic channel, and different solid materialsare tested and (b) scatter plot indicating the achieved resonant frequenciesover materials with different permittivity.According to Figure 5.6(a), when the resonant frequency was adjustedto 2.43 GHz, which corresponds to 25 mm amount of liquid metal, materi-als with different permittivity exhibited different frequency shifts, and that755.3. Design methodology and simulationhappens due to the change in the effective dielectric permittivity in the localsensor environment. As shown in Figure 5.6(b), a material with a permit-tivity of 4.3 shifted to 2.349 GHz from 2.43 GHz and another material witha permittivity of 16.5 moved the resonant frequency to 2.255 GHz.Furthermore, the sensitivity was studied for various amounts of liquidmetal volume inside the microchannel to extract the sensitivity curve fordifferent settings, which has been demonstrated in Figure 5.7. The highestfrequency difference of 224 MHz was attained for material with permittivity16.5 at the 18 mm liquid metal location. The lowest frequency difference,49 MHz, was attained for material with permittivity 4.3 at the 32 mmliquid metal location. According to the simulated results, lower resonantfrequency corresponded to a higher amount of liquid metal, demonstratingless sensitivity to the same MUT, as expected based on equation 4.1. Thus,liquid metal enables characterizing the materials at different frequencies sothat materials properties can be understood over a frequency band, muchlike modern spectroscopic characterization techniques.Figure 5.7: Frequency difference over different dielectric constant materialswhen liquid metal is in different locations inside the microfluidic channel.765.3. Design methodology and simulationDifferent loss tangent standard solid materialsFigure 5.8: Simulation results from HFSS while varying loss tangent of testsamples of permittivity 16.5, 10.0, and 5.7 at different liquid metal settingsinside the microchannel. Varying loss tangent of test samples varies theamplitude of the sensor response while liquid metal is at (a) 18 mm, (b) 21.5mm, (c) 25 mm, (d) 28.5 mm, and (e) 32 mm location. The magnification ofthe amplitude difference is shown in the bottom right inset at every liquidmetal setting.775.3. Design methodology and simulationTo characterize the designed sensor structure, the impact of differentdielectric loss tangent for MUTs with the same real permittivity was stud-ied with different amounts of liquid metal. MUT’s with r = 16.5, 10 and5.7 were placed on top of the interdigitated capacitive region and simulatedwith different values of loss tangents (tan δ = 0.005, 0.01, 0.015 and 0.02)at different liquid metal configurations (18 mm, 21.5 mm, 25 mm, 28.5 mmand 32 mm). With different loss tangents, the transmission amplitude re-sponse corresponds to the loss property of the MUT at resonance, while thefrequency remains unchanged. The S21 amplitudes of the sensor with vary-ing MUT loss tangents are shown in Figure 5.8 under various liquid metalconditions. Based on simulation results, a material with a permittivity of10 and a varying dielectric loss tangent is reported in Figure 5.9 at differentliquid metal levels. The result shows a very stable linear response in termsof amplitude and dielectric loss tangent sensitivity. As shown in Figure 5.9,the sensor had less sensitivity at higher amounts of liquid metal, reflectedin the slope of the lines.Figure 5.9: Amplitude difference compared against the loss tangent for aMUT of permittivity 10 at different locations of liquid metal.785.3. Design methodology and simulation5.3.3 Sensor performance analysisFigure 5.10: Simulated result from HFSS while testing the materials at (a)2.258 GHz and (b) 2.55 GHz of frequency.The dielectric data was extracted from Fig.5.1 and imported into simula-tion software HFSS. The designed resonator with liquid metal was then usedto characterize the materials at a frequency of 2.258 GHz. The simulatedresult is presented in Figure 5.10a. As shown in Fig.5.10a, both materialshave the same frequency response around 2.258 GHz because they have iden-tical dielectric properties at the resonant frequency (Figure 5.1). Looking atthe material’s response, the resonant profile of both the materials is nearly795.4. Experimental setup and measurement discussionidentical, and as a result, the two materials cannot be distinguished. Byadjusting the liquid metal, the resonant frequency of the designed resonatoris moved to 2.55 GHz, and the same materials were simulated with the resultdisplayed in Figure 5.10b. The material properties no longer match identi-cally at this frequency, and the sensor can accurately distinguish betweenthe two materials. Testing the materials at multiple frequencies can fullycharacterize MUTs. Based on the presented results, a tunable resonatorwith an adjustable resonant profile shows promising potential in enhancingmicrowave resonator sensors’ selectivity compared to sensors with a fixedresonant frequency.5.4 Experimental setup and measurementdiscussionSeveral sensors were fabricated with different sizes and structures, andthe presented one was selected based on its performance. At first, the pre-sented resonator sensor was characterized by measuring the sensitivity to-wards solid samples in the presence of the liquid metal. As explained inchapter 4, different volumes of liquid metal were used to control and adjustthe sensor’s resonant profile with a focus on the resonant frequency param-eter. Finding samples/mixtures with similar dielectric properties is one ofthe essential parts of validating sensor performance. This has been done bymeasuring the dielectric properties of liquid samples/mixtures through thecoaxial probe. Experiments were performed in three steps to achieve andprove the benefits of resonant based microwave spectroscopy. The steps areexplained at length in the next sections to understand the overall measure-ment procedure.5.4.1 Characterizing the designed resonator sensorFabrication of the sensor structureTo fabricate the designed resonator sensor and the input-output trans-mission line, a Rogers RT/duroid 5880 high-frequency substrate was usedwith a permittivity of 2.20 ± 0.02. The thickness and the loss tangent ofthe dielectric substrate were 0.79 mm and 0.0009, respectively. The sub-strate surface was covered with a coating of 35 µm thick copper, and copperwas specifically designed to model the microstrip lines of the resonator sen-sor and the input-output transmission lines. Copolyester was used in 3Dprinting technology in a particular pattern to construct the microchannel805.4. Experimental setup and measurement discussionfor liquid metal transfer and solid sensing. The permittivity and the losstangent of the copolyester were 2.1 and 0.008, respectively.Using plastic nuts and bolts, the channel was fixed to the resonatorto avoid instability and attain higher accuracy and sensitivity. The 3D-printed microchannel was detached by 50 µm from the resonator structureto ensure non-contact characterization. Inlet-outlet tubes were attached tothe microchannel to transfer the liquid metal to achieve frequency tuning.To record the transmitted signal strength (S21), an N5222B vector networkanalyzer (VNA) was used by setting the output power and intermediatefrequency bandwidth to 0 dBm and 300 Hz, respectively. The experimentalsetup for this experiment is shown and labeled in Figure 5.11.Figure 5.11: Experimental setup of the designed resonator used for testingstandard samples (solid) at different resonant frequencies. The magnificationof the sensor with MUT is inset in the top right corner.815.4. Experimental setup and measurement discussionMeasured results of standard solid samples detecting in differentresonant frequenciesInitially, the experiment was designed to tune the resonator to achievedifferent resonant frequencies using only liquid metal and to introduce solidmaterials in the sensing region for each resonant frequency to understandthe sensor’s behavior. Solid samples of permittivity 2.20, 6.15, and 10.02and sizes of 0.79 × 10 × 10 mm were chosen for this experiment. Galinstan,a liquid metal alloy of gallium, indium, and tin, were moved through themicrochannel to tune the resonant frequency. Since liquid metal requiressubstantial pressure to move through the channel because of its high surfacetension, Hydrocal 2400 oil was initially passed through the microchannel asa lubricator and anti-oxidation layer.Figure 5.12: Measured response of liquid metal being at different locationsinside the microchannel resulting in tunable resonant frequencies.825.4. Experimental setup and measurement discussionDifferent amounts of liquid metal were pushed through the microchan-nel, indicated as location 1 through location 4. Each of the locations wascharacterized by the length of microchannel filled with liquid metal; withlocation 1, 2, 3, and 4 containing liquid metal lengths of 18 mm, 21 mm,24 mm, and 27 mm of liquid metal as shown in Figure 5.12. The mea-sured resonant frequencies with liquid metal in location 1 to location 4 were2.68, 2.50, 2.20, and 2.10 GHz, respectively. Figure 5.12 shows the differentlocations and their responses regarding resonant frequency and amplitude.The amplitude of the responses shown in Fig.5.12 decreased with the in-creased volume of liquid metal because the increased capacitive couplingintroduced loss mechanisms through the ground layer of the substrate. Atevery recorded resonant frequency, solid samples were placed above the in-terdigitated capacitor, and the measured results are recorded.Figure 5.13: Measured response of standard materials (r=2.20, 6.15 and10.20) influencing resonant frequency of 2.68 GHz.Different materials affect the resonant profile because of their varying di-electric properties. To understand the behavior of different materials on theresonant response, Figure 5.13 presents measured results recorded for differ-ent test samples (r = 2.20, 6.15 and 10.20) while keeping the liquid metalat location 1 corresponding to the resonant frequency of fr = 2.68 GHz.835.4. Experimental setup and measurement discussionAs expected, the higher permittivity materials result in a larger resonantfrequency change, and at location 1, the sensitivity was found to be 35.91MHz / unit . As shown in Fig. 5.13, the sensor structure demonstrateddifferent sensitivity at different resonant frequencies, adjusted by the lengthof the liquid metal over the ring. Resonant frequency-dependent sensitivitycan be attributed to the sensor’s inherent behavior, which originated fromthe sensor’s initial capacitance. At low frequency, the sensor’s initial capac-itance is larger than high-frequency capacitance leading to lower sensitivityat low-frequencies. The measured response of the test sample (r = 6.15)in the presence of liquid metal at different locations is presented in Figure5.14.Figure 5.14: Measured response of standard material (r=6.15) influencingdifferent resonant frequencies.The material response presented illustrates that a lower volume of liq-uid metal has a higher sensitivity to the material than a high volume ofliquid metal inside the microchannel. The results are recorded and ana-lyzed for the other samples at different resonant frequencies. The measuredfrequency shift in different test samples’ presence was compared to the res-onant frequency achieved by adding the liquid metal to the microchannel,as presented in Figure 5.15. Considering frequency difference, ∆F = fH -845.4. Experimental setup and measurement discussionfL, where fH is the resonant frequencies achieved by adding liquid metalwith no material under test and fL is the frequency measured in the pres-ence of test samples in those resonant frequencies. It was realized from themeasured results that when the resonant frequency was 2.68 GHz (liquidmetal at location 1), the test sample of permittivity 10.20 had the high-est frequency difference of 330.4 MHz. The frequency difference graduallydecreases at the lower resonant frequencies as expected, due to their lowerrelative capacitance. The effect of liquid metal on sensitivity was character-ized, and the sensitivities were found to be 35.91, 24.66, 9.09, and 5.93 MHz/ unit  in location 1, location 2, location 3, and location 4, respectively.Figure 5.15: Sensitivity of the designed resonator sensor by testing samplesof permittivity 2.20, 6.15 and 10.20 in different resonant frequency.5.4.2 Standard probe measurementMeasurement setupThe dielectric properties of a material are essential to know before theycan be accurately sensed and understood with a microwave resonator sen-sor such as the one designed. It also adds a greater scientific value to thework presented. Dispersion (′) and absorption (′′) parameters of materi-855.4. Experimental setup and measurement discussionals display material characteristics over a range of frequency and can helpto understand the material behavior. There are several techniques capableof measuring the dielectric properties of a material. The standard coax-ial probe method is best suited for liquids and semi-liquid materials in thiscontext [42]. It is simple, convenient, and non-destructive [42]. In this exper-iment, samples of similar dielectric properties are extracted experimentallyby measuring the dielectric properties of pure and diluted alcohols.Figure 5.16: Experimental setup of the coaxial probe measurement.A high-temperature probe N1501A shown in Figure 5.16 was used forthe experiment, which can output and detect a wideband frequency of 10MHz to 20 GHz. The VNA was connected to the probe for input reflectionco-efficient (S11) measurements. The frequency range, power, and IF band-width of the high-temperature probe was set to 2-4 GHz, 0 dBm, and 300 Hz,respectively. Each of the readings was taken with 201 evenly distributed datapoints. Coaxial probe measurements used a three-load standard calibrationprocedure: open circuit, short circuit, and a broadband load. Deionizedwater was used as the broadband load, and the temperature was recordedat 23°C during the calibration. After the calibration was performed, the865.4. Experimental setup and measurement discussiondielectric properties of liquid samples, ethanol, methanol, and isopropanol(IPA) were measured.Measurement results of the dielectric probe experimentThe dielectric properties of the materials can easily be extracted fromthe literature available in scientific journals. However, in this case, the liquidsamples were tested and compared for every concentration to determine theintersection point of their dielectric property. Referencing the data from theliterature will not help find the same dielectric permittivity of two differ-ent samples. The presented approach here to perform the dielectric probemeasurement helps identify the intersection of two different liquids in a par-ticular frequency. The intersection point shows that two different liquidsamples exhibit the same dielectric permittivity in a particular frequency.Different concentrations of liquid samples were made and tested usingthe dielectric probe. After measuring the pure liquids, mixed solutions weremade with volumetric concentration differences of 5%, and the results wererecorded. This was done to find two different mixtures, or a mixture and apure substance, which have the same dielectric permittivity at a particularfrequency. The collected measured data from this experiment is shown inTable 5.2.Table 5.2: Measured effective permittivity of the sample liquid solutionsConcentrations(%) Frequency Effective permittivityEthanol-water IPA-water (GHz) (eff )15 10 3.13 68.7230 25 3.94 52.5940 35 3.70 45.9765 55 3.22 26.4995 80 3.49 10.48The measured real (′) and imaginary part (′′) of the complex permit-tivity is presented in Figure 5.17. There are samples and mixtures thatshowed equivalent properties, making it impossible to distinguish betweensuch solutions looking at their dispersion property. However, it is possibleto distinguish between samples and mixtures by seeing the absorption prop-erty using the coaxial probe method. Nevertheless, the readout circuitry isexpensive and probably would not be used in everyday sensors.875.4. Experimental setup and measurement discussionFigure 5.17: Measured real (′) and imaginary part (′′) of the complexpermittivity of different solutions where (a) 15% ethanol-water and 10%IPA-water, (b) 30% ethanol-water and 25% IPA-water, (c) 40% ethanol-water and 35% IPA-water and (d) 65% ethanol-water and 55% IPA-water.885.4. Experimental setup and measurement discussionLess complex readout circuitry like oscilloscope or frequency demodula-tor can only operate with a single frequency input or only interpret a reso-nant frequency as its data collection [171]. Based on these limitations, losstangent would not be detected at all. The intersect points highlighted in therelative permittivity over the frequency spectrum in Figure 5.17 create theproblem while distinguishing between the samples using a resonant-basedsensor. The collected measured data from the standard probe measurementshown in table 5.2 is visualized in Figure 5.17.Figure 5.18: Dielectric response of (a) 80% and 85% ethanol-water comparedwith pure methanol and (b) 80% and 85% IPA-water compared with pureethanol.For experimental simplicity, the effective permittivity of pure methanolwas compared with the effective permittivity of an ethanol-water solution.The measured results are presented in Figure 5.18. It is found from Figure5.18(a) that the effective permittivity of pure methanol lies between 80%- 85% ethanol in a water solution. Such a solution highlights the inability895.4. Experimental setup and measurement discussionto distinguish between pure methanol and concentrated ethanol (81%) forthe frequency range using only frequency-based methods. Similarly, Figure5.18(b) describes equivalent behavior in comparing the effective permittivityof pure ethanol and an IPA-water solution. Considering the measured resultsfrom the standard probe experiment shown in Fig.5.18 and the achieved res-onant frequencies, 81% ethanol-water solution and 83% IPA-water solutionwas chosen by data interpolation method to compare with pure methanoland pure ethanol, respectively.The significance of dielectric probe measurements is very crucial to ad-dress the benefits of resonant based spectroscopy. By identifying the con-centrations and their referenced frequency where different samples displaysimilar behavior from the probe measurement, the final step to address thesolution using a resonant-based spectroscopy sensor can be explained. Twounlike liquids showed the same dielectric response at a specific frequency.While specific solutions and frequencies were required to demonstrate theproblem, it could fit many real-world examples as microwave sensors striveto fit more and more applications. Effectively, it is difficult to character-ize and identify such liquids with a simple system. A frequency selectivemicrowave sensor provides the flexibility to measure a frequency selectivematerial’s dielectric properties in an effective manner. It is also beneficialto understand the dielectric behavior of a material in different frequencies.This allows us to identify the like materials using a simple microwave planarresonator sensor.5.4.3 Validation of the designed resonator sensorIn this section of the experiment, the performance of the designed res-onator sensor will be validated experimentally. The designed microwavesensor used for sensing solid samples to characterize the sensor has beenused for this experiment. A new microchannel was 3D-printed to sense liq-uid samples where two different paths have been designed. A straight pathwas considered to flow the liquid metal through, and the U-shaped path wasmade for the liquid samples to flow. Inlet and outlet tubes were connectedto the U-shaped path to allow the liquid samples to pass. The channel wascorrectly aligned with the microwave sensor. A VNA was connected to theresonator ports to record the transmitted power (S21). The experimentalsetup and the 3D printed microchannel for this experiment are presented inFigure 5.19.Initially, four different positions have been marked in the microchannelusing a black marker to limit the liquid metal position. Hydrocal 2400 oil905.4. Experimental setup and measurement discussionFigure 5.19: Experimental setup of the designed sensor used for sensingliquid samples of pure methanol, pure ethanol, 81% ethanol-water and 83%IPA-water solutions at different resonant frequencies.was initially pushed through the microchannel to lubricate the tube andthe channel using an mm syringe. Galinstan was pushed next through themicrochannel till the second mark, which covers half of the gap and indi-cates a resonant frequency of 2.817 GHz. Each of the sample liquids hasbeen pushed through the inlet tube through the channel, and the responsesare recorded through the VNA. The channel and the tube were cleanedproperly by flowing water each time before testing different liquid samples.Each sample reading has been taken three times to validate the response.Following the same procedure, Galinstan was pushed to two other markedlocations, which indicate a resonant frequency of 2.719 GHz and 2.685 GHz,respectively. The sample liquids have been tested about both the resonantfrequencies, and the results were recorded.915.4. Experimental setup and measurement discussionFigure 5.20: Measured frequency shift due to the presence of pure methanoland 81% ethanol-water at resonant frequencies of (a) 2.817 GHz, (b) 2.719GHz, and (c) 2.685 GHz. The corresponding liquid metal position andfrequency shifts are shown in the top right inset and bottom right inset,respectively.925.4. Experimental setup and measurement discussionFigure 5.21: Measured frequency shift due to the presence of pure ethanoland 83% ethanol-water at resonant frequencies of (a) 2.817 GHz, (b) 2.719GHz and (c) 2.685 GHz.935.4. Experimental setup and measurement discussionSample liquids of pure methanol, pure ethanol, 81% ethanol-water, and83% IPA-water solutions were prepared for testing. Each of the sampleliquids were moved into the channel in reference to the new liquid metallocations and resonant frequencies, and the S21 responses are recorded inthe VNA. The channel and the tube were cleaned by forcing air throughthe channel and then allowing 5 minutes of drying before testing the nextliquid samples. Each sample reading was taken three times to validate theresponse.The measured response of pure methanol was compared with the re-sponse of 81% ethanol-water. In contrast, pure ethanol was compared with83% IPA-water. The measured resonant frequencies achieved by positioningdifferent amounts of liquid metal and the impact of sample liquids (puremethanol and 81% ethanol-water) at those resonant frequencies have beenpresented in Figure 5.20. The measured resonant frequencies are 2.817 GHz,2.719 GHz, and 2.685 GHz. The resonant frequency shifts due to the pres-ence of sample liquids (pure methanol and 81% ethanol-water) inside themicrochannel are presented in Figure 5.20(a), 5.20(b) and 5.20(c), respec-tively. The liquid metal position has been shown in the top right inset, andmagnification of the frequency shifts has been presented in the bottom rightinset of Figure 5.20. Similarly, in Figure 5.21, the transmitted power (S21)due to the presence of sample liquids of pure ethanol and 83% IPA-water atdifferent resonant frequencies have been presented. Figure 5.21(a), 5.21(b)and 5.21(c) indicates the response at resonant frequency of 2.817, 2.719 and2.685 GHz, respectively. From Figure 5.20 and 5.21, it is evident that the liq-uid samples have higher sensitivity at the higher resonant frequency (2.817GHz) compared to the lower resonant frequency of 2.685 GHz, just like inthe solid sample tests presented in section 5.4.1.The frequency differences due to the presence of liquid samples in differ-ent resonant frequencies were calculated and presented in TABLE 5.3.Table 5.3: Measured frequency difference due to sample liquids presence inmicrofluidic channel in different resonant frequencies.Test sample liquids Frequency differences, ∆f(MHz), whenfr=2.817 GHz fr=2.719 GHz fr=2.685 GHzPure Methanol 11.0 6.5 3.081% Ethanol-water 11.0 5.5 2.0Pure Ethanol 9.0 5.5 0.583% IPA-water 9.0 4.5 3.0945.5. Chapter summaryThe measured frequency difference of pure methanol and 81% ethanol-water was found 11 MHz when the resonant frequency was set to 2.817 GHz.So, at 2.817 GHz frequency, the sample liquids cannot be distinguished sinceboth samples provide the same frequency difference. The resonant frequencywas tuned to 2.719 GHz by adjusting the volume of liquid metal inside themicrochannel. The tested samples, pure methanol, and 81% ethanol-water,provide frequency differences of 6.5 MHz and 5.5 MHz, respectively whensensing at 2.719 GHz of frequency. The samples can be identified by lookingat their presented frequency difference. Similarly, the resonant frequencywas shifted to 2.685 GHz, and the provided sample response helped to dis-tinguish between the samples.Moreover, in pure ethanol and 83% IPA-water solution, the measuredfrequency differences are similar at the frequency 2.817 GHz, and frequencydifferences are dissimilar in other frequencies like 2.719 GHz and 2.685 GHz.Other resonant frequencies can also be set for interest by adjusting theamount of liquid metal volume inside the microchannel to achieve the re-quired selectivity. Therefore, measuring the frequency difference at multi-ple, easily tuned resonant frequencies can be effectively used to distinguishbetween similar permittivity samples. The presented approach success-fully demonstrated a potential solution to address this issue by combininghigh resolution and sensitive resonant-based solutions with a spectroscopymethod of sweeping the resonant frequency.5.5 Chapter summarySensing materials over a continuous wideband frequency enables study-ing materials in more detail or investigating attractive surface or high-frequency properties. A microwave resonator sensor with a matching mi-crofluidic channel was designed with sensitivities ranging from 36 to 6 MHzper unit  depending on the state of the resonator. Additionally, whilepermittivity is a unique material property, mixtures and compounds maycombine to have a non-unique permittivity at certain frequencies. The de-velopment of a frequency-tunable microwave resonator sensor has overcomethis by utilizing liquid metal to allow the resonator to operate at multiplefrequencies between 2.1 and 2.8 GHz. The presented dielectric probe mea-surements specified the intercept points of the relative permittivity of twodifferent liquid samples. The intercept points from the probe measurementexplained the identical dielectric permittivity of liquid samples at a fre-quency. Sensing these materials using a resonator sensor operating at that955.5. Chapter summaryfrequency will not help distinguish between the samples, and this clearlydemonstrated the challenges in microwave resonator sensors.The sensor operated with a microfluidic channel that could introduce acontrolled amount of liquid metal to increase the overall capacitive loadingof the resonator circuit, adjusting the resonant frequency accordingly. WithHFSS simulations and dielectric probe measurements, the sensor was de-veloped to selectively identify specific standard liquid mixtures of ethanol,methanol, isopropanol, and water. Experimental results confirmed thatsensing at 2.817 GHz was unable to differentiate the two materials, but selec-tivity was achieved by tuning the sensor to 2.719 and 2.685 GHz. Identicaldielectric properties of mixtures could be challenging to predict and untan-gle in industrial or technical sensing. However, with tunable frequencies, itbecomes possible to quickly and easily obtain spectroscopic microwave datato selectively identify materials. The presented results in table 5.3 demon-strate a successful approach towards a potential solution to address thisissue challenge by combining high resolution and sensitive resonant-basedsolutions with a spectroscopy method of sweeping the resonant frequency.96Chapter 6Conclusion and FutureInvestigations6.1 Thesis outcome and conclusionThe main intention of this thesis is to design and develop a planarresonant-based spectroscopy technique that can be employed in sensing ma-terials in microwave sensor technology. The previously reported spectroscopy-based methods perform well towards detecting and characterizing materialsin multiple frequencies. However, due to wideband performance and lessaccuracy of the spectroscopy method, the application scope is limited. Thecombination of resonant-based microwave sensors and the presented wide-band continuous tunability technique based on liquid metal made it possi-ble to achieve highly accurate and wideband frequency tuning in a compactdesign. The operating theory behind this work was investigated, and a suc-cessful potential approach to developing resonant-based spectroscopy waspresented in this thesis.A wideband continuously tunable planar microwave resonator over a fre-quency spectrum was developed by integrating microwave planar resonatorand liquid metal with a 3D printed microfluidic channel. The developed tun-able resonator was utilized as a tool towards an approach to detect materialsof similar dielectric properties in planar sensor technology. The conventionalsplit ring resonator was modified into a highly coupled resonator structureto create higher electromagnetic interaction. Among all the readily availableliquid metals, Galinstan was chosen to use for this investigation because ofits non-toxicity, high electrical conductivity, and environmental compatibil-ity. A 3D printed microchannel lowers the cost and reduces the complexity ofdesign techniques and rapid access to the product. The designed resonatorstructure’s performance for a different amount of liquid metal volume in-side the fluidic channel was validated by comparing the measured resultswith the structure simulation result and presented in subsection 4.3.1. Thestudy shows the great potential of designing SRR sensors with a tunable976.2. Future workfrequency, making a microwave frequency spectroscopy moving forward tosolve the selectivity challenge in microwave planar resonator sensors.In order to enable material sensing, traces of interdigitated capacitivestructure was integrated with the designed structure. The designed sen-sor structure’s performance was characterized by testing the standard solidsamples of different permittivity and loss tangent. A clear concept of theproblem was addressed in subsection 5.1 by using the Debye model, and pureliquids and mixtures that show similar dielectric behavior were determinedby standard probe measurements. The sample liquids of similar dielectricproperties were tested in different frequencies, and the measured results pre-sented in table 5.3 shows multi-phase material identification. Finally, theinvestigation throughout this thesis successfully demonstrated the presentedapproach as a potential solution to distinguish between samples of identicaldielectric properties in microwave sensor technology.The outcomes throughout the investigation of this thesis are as follows:• A wideband continuous frequency tuning of 1.05 GHz with a fairlyconstant amplitude and a quality factor of 70 (±5) was achieved.• While characterizing the designed resonant structure, the resonator’ssensitivity ranging from 36 to 6 MHz per unit  was observed depend-ing on the state of the resonator. The sensor structure demonstratesdifferent sensitivity at different resonant frequencies, adjusted by theliquid metal length inside the microfluidic channel.• The coaxial probe experiment identifies that 81% ethanol-water solu-tion and pure methanol exhibit the same frequency difference of 11MHz at 2.817 GHz of frequency and 83% IPA-water and pure ethanolexhibits the same frequency difference of 9 MHz at 2.817 GHz of fre-quency, indicating that the samples could not be distinguished usinga resonator operating at 2.817 GHz.• A multi-phase material detection was achieved by using the resonantbased spectroscopy technique of sweeping the resonant frequency.6.2 Future workThe presented thesis utilizes the concept of wideband frequency tunabil-ity of the designed resonant structure based on liquid metal as a tool todevelop planar resonant-based spectroscopy. Moreover, several topics have986.2. Future workbeen highlighted as potential future work during the thesis investigation.The suggestions for extending this research are as follows:• In this thesis, frequency tunability using a liquid metal enabled mi-crowave ring resonator is implied in a passive microwave resonator. Apotential future work can be an investigation of the concept in activemicrowave resonators.• The wideband and low-loss tuning performance of the liquid metalintegrated planar microwave resonator sensors encourages the furtherexpansion of liquid metal adaptation to the development of tunableantenna sensors.• Two highly coupled resonators have been designed for achieving tun-ability in this research, and the achieved frequency tuning was 1.05GHz. Different resonant structures can be explored in the future toachieve a broader range of frequency tuning.• The mathematical model that can calculate the amount of liquid metalvolume to achieve a specific resonant frequency is not developed heresince the primary goal of this thesis is to develop the concept of acontinuously tunable microwave resonator. In that case, developing amathematical model to calculate the amount of liquid metal volumeto obtain a particular frequency can be a potential future study.• During an investigation of this thesis, a higher volume of liquid metalover the trace increases the capacitive coupling through the substrate’sground layer, introducing a loss mechanism that results in signal loss.In the future, it can be investigated to eliminate this issue of signalloss and can achieve a consistent signal strength.99Bibliography[1] Elprocus, “Microwaves Technology Basics — Effetcs — Applicationsand Advantages.” [Online]. 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Thesis/Dissertation
eng
Electrical Engineering
Vancouver : University of British Columbia Library
Attribution-NonCommercial-NoDerivatives 4.0 International
Graduate
Tunable microwave resonator using liquid metal and 3D printed fluidic channel for the development of microwave spectroscopy
Text