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An intermediate frequency predistortion linearizer for an earth station traveling-wave tube amplifier Wlodyka, Mark Jan 1986

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AN INTERMEDIATE FREQUENCY PREDISTORTION LINEARIZER FOR AN EARTH STATION TRAVELING-WAVE TUBE AMPLIFIER by MARK JAN WLODYKA B.A.Sc., The Univ e r s i t y of B r i t i s h Columbia, 1976 A THESIS SUBMITTED IN PARTIAL FULFILLMENT OF THE REQUIREMENTS FOR THE DEGREE OF MASTER OF APPLIED SCIENCE i n FACULTY OF GRADUATE STUDIES Department of E l e c t r i c a l Engineering We accept t h i s thesis as conforming to the required standard THE UNIVERSITY OF BRITISH COLUMBIA A p r i l 1986 © Mark Jan Wlodyka In p r e s e n t i n g t h i s t h e s i s i n p a r t i a l f u l f i l m e n t of the requirements f o r an advanced degree a t the U n i v e r s i t y o f B r i t i s h Columbia, I agree t h a t the L i b r a r y s h a l l make i t f r e e l y a v a i l a b l e f o r r e f e r e n c e and study. I f u r t h e r agree t h a t p e r m i s s i o n f o r e x t e n s i v e copying of t h i s t h e s i s f o r s c h o l a r l y purposes may be granted by the head o f my department o r by h i s o r her r e p r e s e n t a t i v e s . I t i s understood t h a t c o p y i n g or p u b l i c a t i o n o f t h i s t h e s i s f o r f i n a n c i a l g a i n s h a l l not be allowed without my w r i t t e n p e r m i s s i o n . Department of t^Lt^C-miClft- /tJ?tJ 6 The U n i v e r s i t y o f B r i t i s h Columbia 2075 Wesbrook P l a c e Vancouver, Canada V6T 1W5 Date QP&IL 1 P _ £ ( 0 /1Q\ ABSTRACT In m u l t i c a r r l e r single-channel-per-carrier (SCPC) communication system design, intermodulation d i s t o r t i o n i s often a l i m i t i n g f a c t o r . The intermodulation d i s t o r t i o n caused by the AM/AM and AM/PM conversion e f f e c t of the microwave transmit high power a m p l i f i e r (HPA) can be reduced by applying l i n e a r i z a t i o n techniques. Various l i n e a r i z a t i o n techniques are reviewed and an appropriate scheme i s selected for use with a SCPC earth s t a t i o n traveling-wave tube a m p l i f i e r (TWTA). A nonlinear model based on the t h i r d -order d i s t o r t i o n phase of a TWTA i s used to develop a new l i n e a r i z e r design model. The amplitude and phase transfer c h a r a c t e r i s t i c s of a 14 GHz TWTA were measured and the r e s u l t s used i n the design of an intermediate frequency (IF) p r e d i s t o r t i o n l i n e a r i z e r . The p r e d i s t o r t i o n l i n e a r i z e r developed uses an a n t i p a r a l l e l diode d i s t o r t i o n generator i n a p a r a l l e l branch network. The l i n e a r i z e r c i r c u i t operates at an IF of 280 MHz preceding an upconverter stage and the 14 GHz TWTA. The l i n e a r i z e r was tested with a 20 watt 14 GHz TWTA under two-tone, m u l t i c a r r l e r , and noise loading conditions. The experimental l i n e a r i z e r c i r c u i t e f f e c t i v e l y reduced the intermodulation d i s t o r t i o n over a consider-able range of normal drive l e v e l s , providing a 3 dB decrease i n output backoff while maintaining a third-order IMD performance of 25 dBc. The l i n e a r i z e r compares favourably with published r e s u l t s of s i m i l a r p r e d i s t o r t i o n l i n e a r i z e r s designed for s a t e l l i t e a p p l i c a t i o n s , and shows po t e n t i a l f or p r a c t i c a l use i n low cost Ku band SCPC earth s t a t i o n s . i i TABLE OF CONTENTS ABSTRACT i i TABLE OF CONTENTS i i i LIST OF FIGURES v LIST OF TABLES v i i i GLOSSARY OF NOTATION i x ACKNOWLEDGEMENTS x i 1. INTRODUCTION 1 1.1 Basic Concepts of L i n e a r i z a t i o n 2 1.2 Objectives of Study 8 1.3 Scope of Study 9 2. REVIEW OF LINEARIZATION TECHNIQUE DEVELOPMENT 12 2.1 P r e d i s t o r t i o n L i n e a r i z a t i o n 12 2.1.1 Analog SSB Applications 13 2.1.2 D i g i t a l T e r r e s t r i a l Radio Applications 20 2.1.3 S a t e l l i t e Communications Applications 27 2.1.3.1 RF L i n e a r i z a t i o n 28 2.1.3.2 Baseband L i n e a r i z a t i o n 38 2.2 Other L i n e a r i z a t i o n Techniques 40 2.2.1 Feed-forward L i n e a r i z a t i o n 40 2.2.2 Negative Feedback L i n e a r i z a t i o n 42 2.3 Comparative Summary 43 2.3.1 C l a s s i f i c a t i o n of L i n e a r i z e r C i r c u i t s 44 2.3.2 P r a c t i c a l Considerations 45 i i i 3. NONLINEAR DEVICE MODELING 49 3.1 Amplitude-Phase Bandpass Nonlinear Device Model 49 3.2 Quadrature Bandpass Nonlinear Device Model 50 3.3 Heiter's Nonlinear Device Model 53 4. DEVELOPMENT OF NEW LINEARIZED TWTA MODEL 55 4.1 TWTA Model 55 4.2 P r e d i s t o r t i o n L i n e a r i z e r Model 58 4.3 Linearized TWTA System Model 63 5. DESIGN OF IF PREDISTORTION LINEARIZER 68 5.1 Measurement of TWTA Amplitude and Phase C h a r a c t e r i s t i c s ....... 68 5.2 Development of L i n e a r i z e r System Structure 71 5.3 Hardware R e a l i z a t i o n of IF P r e d i s t o r t e r 76 5.3.1 L i n e a r i z e r C i r c u i t Operation 76 5.3.2 D i s t o r t i o n Generator Design 81 6. EVALUATION OF LINEARIZED TWTA IMD PERFORMANCE 84 6.1 Development of M u l t i c a r r i e r Intermodulation Test System 84 6.2 Results of Intermodulation Measurements 88 6.2.1 Two-tone IMD Results 88 6.2.1.1 Optimization at Various Output Backoff Levels . 88 6.2.1.2 Frequency Response C h a r a c t e r i s t i c s 90 6.2.1.3 Temperature Response C h a r a c t e r i s t i c s 90 6.2.2 M u l t i c a r r i e r and Noise Loading IMD Results 93 6.3 Overview of L i n e a r i z e r Performance 96 7. CONCLUSIONS • 100 8. REFERENCES 102 APPENDIX A Frequency Plan for M u l t i c a r r i e r IMD Measurements 108 APPENDIX B HP 9826 Programs and Output L i s t i n g s I l l i v LIST OF FIGURES Figure Page 1.1 T y p i c a l TWTA Amplitude and Phase Transfer C h a r a c t e r i s t i c s 3 1.2 Typi c a l Two-Tone Input and TWTA Output Power Spectrum Near Saturation 5 1.3 Block Diagram of L i n e a r i z e r Techniques 7 2.1 Heun and K i e s e l IF P r e d i s t o r t i o n L i n e a r i z e r 14 2.2 Hecken et a l . IF P r e d i s t o r t i o n L i n e a r i z e r 14 2.3 Egger et a l . IF P r e d i s t o r t i o n L i n e a r i z e r 17 2.4 Igarashi et a l . RF P r e d i s t o r t i o n L i n e a r i z e r 17 2.5 Lenz IF P r e d i s t o r t i o n L i n e a r i z e r 19 2.6 Lenz RF P r e d i s t o r t i o n L i n e a r i z e r 19 2.7 Holz IF P r e d i s t o r t i o n L i n e a r i z e r 22 2.8 Nojima et a l . RF P r e d i s t o r t i o n L i n e a r i z e r s 23 2.9 Horn and Egger Hybrid P r e d i s t o r t i o n L i n e a r i z e r 26 2.10 Saleh and Salz Adaptive P r e d i s t o r t i o n L i n e a r i z e r 26 2.11 Bremenson and Jaubert RF P r e d i s t o r t i o n L i n e a r i z e r 29 2.12 Satoh et a l . RF P r e d i s t o r t i o n L i n e a r i z e r 34 2.13 Satoh et a l . Double Loop P r e d i s t o r t i o n L i n e a r i z e r 34 2.14 Kumar et a l . RF P r e d i s t o r t i o n L i n e a r i z e r 37 2.15 Girard and Feher Baseband P r e d i s t o r t i o n L i n e a r i z e r 39 2.16 Seidel Feed-forward L i n e a r i z e r 41 2.17 Gajda and Dou v i l l e Negative Feedback L i n e a r i z e r 41 2.18 C l a s s i f i c a t i o n Diagram of L i n e a r i z a t i o n Techniques 45 v Figure Page 3.1 Quadrature Model of TWTA 52 4.1 Block Diagram of P r e d i s t o r t i o n L i n e a r i z e r Model 59 4.2 Vector Diagram of P r e d i s t o r t i o n L i n e a r i z e r Model 61 4.3 Block Diagram of Linearized TWTA System Model 64 5.1 Block Diagram of TWTA Transfer C h a r a c t e r i s t i c s Measurement System 69 5.2 Measured 14 GHz TWTA Transfer C h a r a c t e r i s t i c s 72 5.3 Proposed Linearized Earth Station TWTA Configuration 75 5.4 280 MHz P r e d i s t o r t i o n L i n e a r i z e r C i r c u i t 77 5.5 A n t i p a r a l l e l Diode Nonlinear Voltage Transfer C h a r a c t e r i s t i c s ... 79 5.6 Photograph of 280 MHz P r e d i s t o r t i o n L i n e a r i z e r 80 6.1 Block Diagram of Linearized TWTA Evaluation System 86 6.2 TWTA Two-Tone IMD with L i n e a r i z e r Optimized at 4 dB Output Backoff 89 6.3 TWTA Two-Tone IMD with L i n e a r i z e r Optimized at 5 dB Output Backoff 89 6.4 TWTA Two-Tone IMD with L i n e a r i z e r Optimized at 6 dB Output Backoff 89 6.5 TWTA Two-Tone IMD with L i n e a r i z e r Optimized at 7 dB Output Backoff 89 6.6 TWTA Two-Tone IMD with IF Carri e r s at 262.745 and 263.654 MHz ... 92 6.7 TWTA Two-Tone IMD with IF C a r r i e r s at 271.145 and 272.054 MHz ... 92 6.8 TWTA Two-Tone IMD with IF Carriers at 287.945 and 288.854 MHz ... 92 6.9 TWTA Two-Tone IMD with IF Ca r r i e r s at 296.345 and 297.254 MHz ... 92 v i Figure Page 6.10 TWTA Two-Tone IMD with L i n e a r i z e r Temperature at 0°C 94 6.11 TWTA Two-Tone IMD with L i n e a r i z e r Temperature at 12°C 94 6.12 TWTA Two-Tone IMD with L i n e a r i z e r Temperature at 38°C 94 6.13 TWTA Two-Tone IMD with L i n e a r i z e r Temperature at 50°C 94 6.14 TWTA Centre Slot Third-Order IMD with 4 Carri e r s 95 6.15 TWTA Centre Slot Third-Order IMD with 6 C a r r i e r s * 95 6.16 TWTA Centre Slot Third-Order IMD with 8 Carri e r s 95 6.17 TWTA Centre Slot NPR 95 6.18 TWTA Saturated Output Power vs. Number of Carrie r s 97 6.19 TWTA Output Backoff at 25 dBc IMD vs. Number of C a r r i e r s 97 6.20 Two-Tone IMD Performance Curves of the Bremenson TWTA Li n e a r i z e r 99 v i i LIST OF TABLES Table Page 2.1 IF/RF P r e d i s t o r t i o n L i n e a r i z e r Feature Summary 48 5.1 Measured and Modeled TWTA Amplitude and Phase Transfer C h a r a c t e r i s t i c s at 14.05 GHz 73 6.1 Two-Tone Test Frequency Assignment 91 A . l M u l t i c a r r l e r Test Frequency Assignment 110 v i i i GLOSSARY OF NOTATION Symbol D e f i n i t i o n A/D Ana l o g - t o - d i g i t a l converter AM Amplitude modulation AM/AM Amplitude-to-amplitude conversion AM/PM Amplitude-to-phase conversion BB Baseband BER B i t - e r r o r - r a t e B/0 Backoff C/N Carri e r - t o - n o i s e r a t i o D/A Digi t a l - t o - a n a l o g converter dB Decibel dBc Decibels below c a r r i e r l e v e l FET F i e l d - e f f e c t Transistor FDM Frequency-division multiplexing FDMA Frequency-division multiple access FM Frequency modulation I In-phase data channel HPA High power amplifier IF Intermediate frequency I-V Current-voltage IMD Intermodulation d i s t o r t i o n KW Kilowatt i x Symbol D e f i n i t i o n LRZ L i n e a r i z e r Mb/s Megabits per second MESFET Metal semiconductor f i e l d - e f f e c t t r a n s i s t o r MIC Microwave integrated c i r c u i t NPR Noise-power-ratio PM Phase modulation PSK Phase-shift-keying Q Quadrature phase data channel QAM Quadrature amplitude modulation QPSK Quadrature phase-shift-keying RAM Random-access memory RF Radio frequency SCPC Single-channel-per-carrier SL Soft l i m i t e r SSB Single sideband SSPA S o l i d - s t a t e power a m p l i f i e r TDM Time-division multiplexing TDMA Time-division multiple access TWTA Traveling-wave tube a m p l i f i e r VHF Very high frequency x ACKNOWLEDGEMENTS The author wishes to acknowledge the consistent support and h e l p f u l guidance of his supervisors, Dr. J.L. F i k a r t and Dr. E.V. J u l l , throughout th i s research project. The tec h n i c a l assistance of Dr. F i k a r t , p a r t i c u l a r l y i n the l i n e a r i z e r model development, was instrumental i n the success of t h i s study. The support of M i c r o t e l P a c i f i c Research for the duration of this project i s appreciated. Helpful suggestions by Mr. W. Street and by Mr. P. Acimovic during the development of the automated test bed as s i s t e d In the e f f i c i e n t c o l l e c t i o n of intermodulation data. F i n a l l y , I would l i k e to thank my wife, K r i s t i n a , for her patience and encouragement during the writing of th i s report. x i 1 1. INTRODUCTION In both t e r r e s t r i a l and s a t e l l i t e communication system, the nonlinear amplitude and phase c h a r a c t e r i s t i c s of microwave transmit high power amplifiers (HPA) are a major l i m i t a t i o n i n the design and performance of various communication networks. The nonlinear behaviour of power amplifiers such as klystrons, traveling-wave tube amplifiers (TWTA), or s o l i d - s t a t e power am p l i f i e r s (SSPA), generates s i g n i f i c a n t d i s t o r t i o n at si g n a l power le v e l s at or near saturation. This s i g n a l d i s t o r t i o n appears as either intermodulation d i s t o r t i o n (IMD) i n m u l t i c a r r i e r or analog sin g l e sideband (SSB) systems, or s p e c t r a l spreading i n s i n g l e - c a r r i e r d i g i t a l systems. Without employing " l i n e a r i z a t i o n " techniques, these power am p l i f i e r s must be operated well below t h e i r saturated power l e v e l s , or "backed-off", i n order to reduce the s i g n a l impairment to t o l e r a b l e l i m i t s . The operation of these HPAs at reduced power l e v e l s r e s u l t s i n the use of higher priced and higher saturated power a m p l i f i e r s . These HPAs, operated at backed-off power l e v e l s , also reduce the o v e r a l l power e f f i c i e n c y . The concept of l i n e a r i z a t i o n has been used i n recent years to a s s i s t i n reducing the d i s t o r t i o n generated by the f i n a l HPA. Several techniques have been s u c c e s s f u l l y developed for e f f e c t i v e l y " l i n e a r i z i n g " the HPA, r e s u l t i n g i n lower IMD, or reduced sideband regeneration and improved BER performance. These l i n e a r i z e d HPAs operate at higher power l e v e l s , providing improved e f f i c i e n c y and lower system cost. 2 In narrowband s a t e l l i t e m u l t i c a r r i e r communication systems, such as single-channel-per-carrier (SCPC), system designers are p a r t i c u l a r l y concerned with the degree of intermodulation d i s t o r t i o n generated by the nonlinear microwave high power ampli f i e r used i n the earth s t a t i o n transmit-ter and s a t e l l i t e transponder. A simple and r e l i a b l e HPA l i n e a r i z e r provides a means for improving system performance and e f f i c i e n c y at a reduced system cost. In l i g h t of these requirements, the following study w i l l investigate various l i n e a r i z a t i o n techniques used i n both t e r r e s t r i a l and s a t e l l i t e communication systems. A study of nonlinear device models w i l l provide i n s i g h t into the d i s t o r t i o n generating mechanism of microwave HPAs and a s s i s t i n the design of a p r a c t i c a l l i n e a r i z e r for use i n a m u l t i c a r r i e r SCPC earth s t a t i o n . 1.1 Basic Concepts of L i n e a r i z a t i o n The complex transfer functions of the three main types of HPAs, namely TWTAs, klystrons, and t r a n s i s t o r SSPAs, exhibit both amplitude compression and phase s h i f t c h a r a c t e r i s t i c s at power l e v e l s close to saturation. The s i n g l e - c a r r i e r amplitude-to-amplitude conversion (AM/AM), and amplitude-to-phase conversion (AM/PM) c h a r a c t e r i s t i c s of a t y p i c a l TWTA are shown i n F i g . 1.1. The input and output backoff (B/0) l e v e l s are defined as the r e l a t i v e power difference between the absolute power and the single-tone saturation power at the input and output r e s p e c t i v e l y . The saturation power i s defined as the maximum output power c a p a b i l i t y of the HPA. As the output power of the HPA approaches saturation, the nonlinear behaviour of the device increases, causing both amplitude and phase d i s t o r t i o n . TWTAs exhibit an F i g . 1 . 1 . T y p i c a l TWTA Amplitude and Phase T r a n s f e r C h a r a c t e r i s t i c s 4 actual decrease i n output power when overdriven past saturation, creating severe d i s t o r t i o n for m u l t i c a r r i e r s i g n a l s -For m u l t i c a r r i e r signals, nonlinear HPAs generate intermodulation d i s t o r t i o n products. In narrowband systems, where the passband i s less than an octave i n bandwidth, only odd-order IMD products are considered s i g n i f i c a n t . For two c a r r i e r s , these IMD products occur i n the output frequency spectrum at equally spaced i n t e r v a l s around the main s i g n a l c a r r i e r s as shown i n F i g . 1.2. The harmonic and even-order IMD products are normally outside the passband of the system and are often removed by bandpass f i l t e r s . For two c a r r i e r s at frequencies A and B, the third-order intermodulation products i n the passband appear at frequency 2A-B and 2B-A. The f i f t h - o r d e r products appear at frequencies 3A-2B and 3B-2A and so on for higher orders. For both two and more c a r r i e r s , the third-order IMD i s dominant [1], and i s usually of primary concern. The two-tone intermodulation measurement i s a standard technique often used i n the comparison of nonlinear devices. The power of the third-order IMD product i s measured r e l a t i v e to the fundamental c a r r i e r power and i s s p e c i f i e d i n dBc. Signal d i s t o r t i o n caused by the HPA n o n l i n e a r i t y also occurs i n s i n g l e - c a r r i e r d i g i t a l l y modulated s i g n a l s . The d i s t o r t i o n appears as s p e c t r a l spreading, with an increase i n the sideband power around the main spectrum of the modulated s i g n a l . The system BER performance i s also a f f e c t e d . To reduce the amplitude and phase n o n l i n e a r i t i e s of HPAs, various l i n e a r i z a t i o n schemes have been developed which attempt to counteract t h i s d i s t o r t i o n causing mechanism. The techniques of l i n e a r i z a t i o n f a l l into 5 H Input Power Spectrum (dB) -50 - 100 Frequency (e,g. 1 MHz/Div) Output Power Spectrum (dB) - 50 - 100 Frequency (e.g. 1 MHz/Div) F i g . 1.2. T y p i c a l Two-Tone Input and TWTA Output Power Spectrum Near S a t u r a t i o n 6 three main categories, namely p r e d i s t o r t i o n , feed-forward, and negative feedback. Block diagrams of the three main techniques are shown i n F i g . 1.3 [2]. The technique of p r e d i s t o r t i o n l i n e a r i z a t i o n p r e d i s t o r t s signals before they are amplified by the HPA. Using t h i s technique, a pr e d i s t o r t e r generates an inverse amplitude and phase transf e r c h a r a c t e r i s t i c that either cancels the odd-order intermodulation produced by the HPA, or reduces the AM/AM and AM/PM d i s t o r t i o n e f f e c t s which degrade s i n g l e - c a r r i e r d i g i t a l systems. The desired net e f f e c t of the cascaded system i s an input-output envelope amplitude trans f e r c h a r a c t e r i s t i c which approaches that of an i d e a l l i m i t e r . The envelope piece-wise l i n e a r l i m i t e r c h a r a c t e r i s t i c provides an excellent c h a r a c t e r i s t i c for most l i n e a r i z e d HPA systems [3-5] . The pr e d i s t o r t i o n technique has been implemented at radio frequencies (RF), intermediate frequencies (IF), and at baseband frequencies. The IF and baseband pr e d i s t o r t e r s are designed to operate preceding the upconverter stage. Feed-forward techniques have also been used to reduce the d i s t o r t i o n generated by the nonlinear HPA. In t h i s approach, an a u x i l i a r y HPA, s i m i l a r i n behaviour to the o r i g i n a l , i s added to the system. By c a r e f u l l y extracting the d i s t o r t i o n components from the o r i g i n a l HPA, and amplifying the d i s t o r t i o n s i g n a l by the a u x i l i a r y HPA, the combined output s i g n a l e f f e c t i v e l y reduces the d i s t o r t i o n products. The process can be applied i t e r a t i v e l y to further reduce d i s t o r t i o n . The t h i r d l i n e a r i z a t i o n approach employs a feedback network around the HPA. C l a s s i c a l feedback analysis i s used to provide the optimum network for 7 1 PREDISTORTER P O W « \ j AMPUflER^> ; OUTPUT INPUT j j PREDISTORTION POWER AMPLIFIER DISTORTION AMPUFIER FEED FORWARD DISTORTION REDUCTION 8 AND « REPRESENT DISTORTION SIGNAL COMPONENTS T AND T' ARE AMPLIFIER DELAYS FEEDBACK NETWORK P O W E R - ^ ^ > > i » INPUT OUTPUT NEGATIVE FEEDBACK FOR DISTORTION REDUCTION F i g . 1.3. B l o c k Diagram o f L i n e a r i z e r Techniques [2] 8 d i s t o r t i o n reduction. S t a b i l i t y problems at high frequencies, however, have reduced the effectiveness of t h i s technique. A l l three techniques have been proven experimentally, but pre d i s t o r t i o n i s the approach used most often i n commercial communication systems. For the main part of th i s l i n e a r i z e r study, the nonlinear transfer function of the HPA and the l i n e a r i z e r w i l l be described as frequency independent, which i s adequate when dealing with most narrowband signals transmitted at microwave frequencies. Nonlinear t r a n s f e r functions are generally frequency dependent when the bandwidth of the input s i g n a l i s comparable to the actual passband of the device, such as may be the case when MESFET power devices, coupled-cavity type TWTAs, or klystrons are used as HPAs. 1.2 Objectives of Study The purpose of t h i s i n v e s t i g a t i o n i s to study HPA l i n e a r i z a t i o n techniques, obtain a s i m p l i f i e d model of a nonlinear TWTA and use t h i s model to develop a p r a c t i c a l l i n e a r i z a t i o n scheme. The d e t a i l e d objectives are: a) to investigate various l i n e a r i z a t i o n techniques and se l e c t a method which can be used i n practice to reduce the intermodulation d i s t o r t i o n generated by a TWTA, b) to investigate HPA models and develop a new s i m p l i f i e d model for a l i n e a r i z e r which can be used conveniently as an e f f e c t i v e design t o o l , c) to design and construct a low cost p r a c t i c a l l i n e a r i z e r using a s i m p l i f i e d design procedure, 9 d) to develop a t e s t procedure which can evaluate the performance of a l i n e a r i z e d TWTA under two-tone, m u l t i c a r r l e r and noise loading conditions, e) to compare the IMD reduction c a p a b i l i t y of the new l i n e a r i z e r with l i n e a r i z e r s reported i n the l i t e r a t u r e . During the i n v e s t i g a t i o n , i t i s hoped that p r a c t i c a l i n s i g h t s into the l i m i t s of l i n e a r i z a t i o n can be obtained. An understanding of the degree of complexity required to su c c e s s f u l l y l i n e a r i z e TWTAs under act u a l p r a c t i c a l operating conditions i s equally important. These t e c h n i c a l factors can be used as guidelines i n the design of future SCPC systems, and to quantify the performance and economic gains achievable i n p r a c t i c a l l i n e a r i z e d TWTA systems. 1.3 Scope of Study Following a comprehensive study of various l i n e a r i z a t i o n techniques and nonlinear HPA models, a s i m p l i f i e d model of a p r a c t i c a l p r e d i s t o r t i o n l i n e a r i z a t i o n scheme i s developed. This model i s used i n the design of a 280 MHz IF p r e d i s t o r t i o n l i n e a r i z e r and the performance of the l i n e a r i z e d TWTA i s examined. The h i g h l i g h t s of t h i s i n v e s t i g a t i o n are summarized below. In Section 2, HPA l i n e a r i z a t i o n schemes for various communication system app l i c a t i o n s are surveyed. Several p r e d i s t o r t i o n techniques are investigated i n depth. For completeness, other l i n e a r i z a t i o n schemes f o r both TWTAs and s o l i d - s t a t e power am p l i f i e r s are summarized. A comparison of design considerations for IF and RF p r e d i s t o r t i o n i s used to s e l e c t a f i n a l design architecture for a p r a c t i c a l TWTA l i n e a r i z e r which can be used i n an SCPC earth s t a t i o n . 10 In Section 3, several a n a l y t i c a l models of nonlinear devices are discussed. These models provide u s e f u l tools which a s s i s t i n developing s u i t a b l e l i n e a r i z e r structures for reducing the d i s t o r t i o n of various HPAs. A s i m p l i f i e d amplitude and phase model based on the third-order d i s t o r t i o n analysis of the HPA was developed i n Section 4. This basic analysis was used to r e l a t e the HPA third-order n o n l i n e a r i t y with the desired nonlinear c h a r a c t e r i s t i c s of the p a r a l l e l branch p r e d i s t o r t e r . A complete system model was then used to analyze the l i n e a r i z e d TWTA. The design of the actual p r e d i s t o r t i o n l i n e a r i z e r i s discussed i n Section 5. The measurement system for obtaining the single-tone amplitude and phase c h a r a c t e r i s t i c s of a t y p i c a l 14 GHz TWTA i s f i r s t described. The development of the l i n e a r i z e d TWTA system and a hardware design procedure i s then shown. Computational formulas were set up to determine i n d i v i d u a l component values and subsystem power l e v e l s . A diode d i s t o r t i o n generator was investigated and integrated into the l i n e a r i z a t i o n system. Computer ass i s t e d numerical simulation studies were performed using the new models, and f i n a l design values for the IF p r e d i s t o r t i o n l i n e a r i z e r were determined. In Section 6, a complex automated test bed developed to measure the performance of both the l i n e a r i z e d and nonlinearized TWTA i s described. Intermodulation measurements were c a r r i e d out over various frequency, temper-ature, and s i g n a l loading conditions. A m u l t i c a r r i e r s i g n a l with s p e c i a l channel spacing was used to accurately measure the intermodulation d i s t o r t i o n of the l i n e a r i z e d system. Third-order and some f i f t h - o r d e r intermodulation measurements were taken to v e r i f y the performance of the l i n e a r i z e r . The o v e r a l l performance of the l i n e a r i z e r was then compared with other p r a c t i c a l l i n e a r i z e r s . F i n a l comments on the l i n e a r i z e r development and p r a c t i c a l implications are summarized i n Section 7. 12 2. REVIEW OF LINEARIZATION TECHNIQUE DEVELOPMENT During the past decade, several l i n e a r i z a t i o n techniques have been developed by researchers for microwave tube a m p l i f i e r s and s o l i d - s t a t e devices. L i n e a r i z a t i o n of nonlinear devices has been accomplished by the three methods of p r e d i s t o r t i o n , feed-forward, and negative feedback. The p r e d i s t o r t i o n technique, which has received the. most attention, w i l l be reviewed i n d e t a i l , with a focus on system a p p l i c a t i o n s . The other two techniques w i l l be b r i e f l y discussed, followed by an o v e r a l l assessment of a l l three l i n e a r i z a t i o n approaches. 2.1 P r e d i s t o r t i o n L i n e a r i z a t i o n P r e d i s t o r t i o n techniques have been used i n the l i n e a r i z a t i o n of TWTAs, klystrons, SSPAs, and s o l i d - s t a t e upconverters i n many types of communication systems. The communication system a p p l i c a t i o n s which have suc c e s s f u l l y used l i n e a r i z e r s include analog SSB t e r r e s t r i a l radio, d i g i t a l m u l t i l e v e l quadrature amplitude modulation (QAM) t e r r e s t r i a l radio, and frequency-d i v i s i o n (FDMA) and time-div i s i o n (TDMA) multiple access earth s t a t i o n s . The l i n e a r i z a t i o n of future s a t e l l i t e transponders has also been investigated, and the benefit of s a t e l l i t e (on-board) l i n e a r i z a t i o n has been demonstrated. P r e d i s t o r t e r s have been developed for operation at either RF at the transmit frequency, or at baseband or IF before the upconverter stage. Tradeoffs i n design complexity and system performance d i c t a t e the appropriate l i n e a r i z e r and operating frequency. 13 2.1.1 Analog SSB Applications A major concern i n SSB t e r r e s t r i a l a p p l i c a t i o n s i s the third-order intermodulation d i s t o r t i o n generated by the f i n a l HPA. Allowable system noise and d i s t o r t i o n l e v e l s impose extreme l i n e a r i t y requirements f o r the f i n a l transmit HPAs. The low le v e l s of IMD to l e r a b l e can only be obtained with r e l a t i v e l y high HPA backoffs. At these high backoffs, the HPAs exhibit only weakly nonlinear c h a r a c t e r i s t i c s , which are predominantly third-order. The suppression of third-order IMD to l e v e l s below 60 dBc requires l i n e a r l z e r s with precise inverse match c h a r a c t e r i s t i c s to be e f f e c t i v e . In l i n e a r i z e d t e r r e s t r i a l SSB radio a p p l i c a t i o n s , major contributions have been made by several research teams during the past decade. Heun and K i e s e l [6] i n i t i a l l y developed a simple IF p r e d i s t o r t i o n scheme for a 12 GHz kl y s t r o n . They constructed the single-stage p r e d i s t o r t e r at 600 MHz using a low frequency t r a n s i s t o r a m p l i f i e r as a d i s t o r t i o n generator. As shown i n F i g . 2.1, the actual c i r c u i t consisted of a p a r a l l e l branch l i n e a r delay l i n e path and the d i s t o r t i o n path. By combining the two paths at a given r a t i o and at 180 degress (antiphase), an e f f e c t i v e amplitude expansion c h a r a c t e r i s -t i c i s obtained. The fundamental requirement i n t h e i r approach was the close match i n the amplitude transfer c h a r a c t e r i s t i c of the t r a n s i s t o r amplifier and the k l y s t r o n over as large a dynamic range as possible. No attempt was made to match the phase c h a r a c t e r i s t i c s , but s i g n i f i c a n t improvements i n third-order IMD, i n the range of 20 dB, were achieved with t h i s simple c i r c u i t . A subsequent report by the authors [7] revealed that a two-tone third-order IMD of 70 dB was obtained at 7 dB below the two-tone saturation power. 14 3 *2 X X Fig. 2.1. Heun and Kiesel IF Predistortion Linearizer [6] MAIN PATH ORIVER AMPLIFIER PHASE RESOLVER DELAY LINE NONLINEAR GENERATOR CXI BE R PATH OUTPUT COUPLER LEVEL SET DISTORTION AOJUST /C>-DISTORTION AMPLIFIER Fig. 2.2. Hecken et a l . IF Predistortion Linearizer [8] 15 Hecken et a l . [8], [9] have developed a p r a c t i c a l 70 MHz IF l i n e a r i z a t i o n scheme used to l i n e a r i z e a 6 GHz TWTA. Their approach i s shown i n F i g . 2.2. Their objective was to reduce the third-order IMD by over 30 dB over a f u l l 30 MHz s i g n a l bandwidth. The operating l e v e l of the TWTA i n the SSB system i s i n the area of weak no n l i n e a r i t y , where the third-order IMD i s r e l a t i v e l y small to begin with. Any reduction i n t h i s small IMD requires a near perfect inverse match of the third-order c h a r a c t e r i s t i c s of the TWTA and the p r e d i s t o r t e r . Hecken et a l . tackled the problem by f i r s t developing a third-order model of a weakly nonlinear TWTA and derived design c r i t e r i a for t h e i r p r e d i s t o r t e r . The actual p r e d i s t o r t e r c i r c u i t u t i l i z e s a p a r a l l e l branch network which combines a l i n e a r s i g n a l and a third-order d i s t o r t i o n component at the desired power r a t i o and phase angle. They generate the third-order d i s t o r t i o n component by means of a patented "cuber" c i r c u i t which consists of four back diodes arranged i n a bridge configuration. Nojima et a l . [10] address a s i m i l a r requirement for reducing the third-order IMD of a 6 GHz TWTA. They described the weak n o n l i n e a r i t i e s of the TWTA by means of the third-order d i s t o r t i o n phase model. They use th i s model to develop a p a r a l l e l branch p r e d i s t o r t e r which operates i n a s i m i l a r fashion to the c i r c u i t design by Hecken et a l . They use a pair of a n t i p a r a l l e l diodes and a four port hybrid c i r c u i t to generate the third-order d i s t o r t i o n component. Nojima et a l . take t h e i r l i n e a r i z a t i o n technique one step further, by adding an adaptive compensating co n t r o l c i r c u i t to t h e i r p a r a l l e l branch p r e d i s t o r t e r . By sampling the re s i d u a l d i s t o r t i o n at the output of the TWTA, t h e i r p r e d i s t o r t e r can automatically adjust i t s d i s t o r t i o n l e v e l to obtain considerable o v e r a l l reduction i n the 16 third-order IMD over longer time i n t e r v a l s without manual readjustment of c i r c u i t elements. At 10 dB output backoff, the third-order IMD reduction was 26 dB. Egger et a l . [11] propose to use an IF p r e d i s t o r t i o n l i n e a r i z a t i o n scheme for a m u l t i c a r r l e r video transmission system using SSB. As ind i c a t e d i n F i g . 2.3, they incorporate an IF p r e d i s t o r t e r (at 650-755 MHz) which provides a separate amplitude and phase compensation network. The amplitude compensator consists of a p a r a l l e l branch network which uses an amplitude expander i n one branch, and an amplitude l i m i t e r i n the other. The amplitude expander i s made up of three diode c i r c u i t s i n s e r i e s , whose nonlinear c h a r a c t e r i s t i c s are c o n t r o l l e d by separate bias currents. The o v e r a l l amplitude response of the p a r a l l e l expander and l i m i t e r approaches the i d e a l inverse amplitude response of the f i n a l 11 GHz TWTA. The desired phase response i s generated by phase modulating the l o c a l o s c i l l a t o r used i n the IF/RF upconversion stage. The phase advance of the l o c a l o s c i l l a t o r i s c o n t r o l l e d by the detected IF s i g n a l envelope power. The seven-carrier third-order IMD using the l i n e a r i z e r i s reduced by over 25 dB from 35 dBc to 60 dBc at 8.5 dB output backoff. Igarashi et a l . [12], have attempted to integrate a RF p r e d i s t o r t i o n l i n e a r i z e r and a 10 watt 6 GHz a m p l i f i e r . They have s u c c e s s f u l l y developed an analog p r e d i s t o r t e r which reduces the third-order IMD by s u f f i c i e n t margin to allow a SSPA to be used i n a SSB t e r r e s t r i a l a p p l i c a t i o n . The pre-d i s t o r t e r consists of a l i n e a r path and a nonlinear path as shown i n F i g . 2.4. The nonlinear path uses a low power FET low power ampli f i e r to generate the third-order d i s t o r t i o n component. This FET i s combined with an a u x i l i a r y 17 F i g . 2.3. Egger e t a l . I F P r e d i s t o r t i o n L i n e a r i z e r [11] RF Input H2 HI 7 Mi^o s i g n a l H3 1 I " H5 114 output Predlst o t e d s i g n a l D i s t o r t i o n component D i s t o r t i o n generator F i g . 2.4. I g a r a s h i e t a l . RF P r e d i s t o r t i o n L i n e a r i z e r [12] 18 branch to produce a s i g n a l which consists of only the third-order d i s t o r t i o n . This s i g n a l i s combined at a preset amplitude and phase with the fundamental s i g n a l i n the main branch. The authors emphasize that a perfect match of p r e d i s t o r t e r and power amplifier stage i s not possible, but i n d i c a t e the range of IMD reduction possible under various amplitude and phase imbalance conditions. The integrated predistorted power ampli f i e r provides a 20 dB reduction i n third-order IMD. Two approaches developed by Lenz [13], [14], are other examples of p r e d i s t o r t i o n l i n e a r i z a t i o n methods for use i n SSB t e r r e s t r i a l a p p l i c a t i o n s . As described i n h i s i n i t i a l work shown i n F i g . 2.5, Lenz has developed a novel IF p r e d i s t o r t i o n technique using two p a r a l l e l connected upconverters• The two mixers upconvert the IF s i g n a l from 70 MHz to 6 GHz. One mixer operates i n i t s l i n e a r region, while the other i s driven at a high l e v e l , generating third-order IMD. The outputs of the two mixers are combined i n a manner which separates the l i n e a r s i g n a l (output of K 3) and the third-order d i s t o r t i o n product (output of K 2 ) . These two output signals are then re-combined at the appropriate power r a t i o and phase angle. A 20 dB reduction i n third-order IMD was obtained using this technique. Lenz' following design shown i n F i g . 2.6 i s a RF p r e d i s t o r t i o n l i n e a r i z e r operating d i r e c t l y at 6 GHz. This p r e d i s t o r t e r also operates by combining a l i n e a r s i g n a l and a third-order d i s t o r t i o n product at a preset power r a t i o and phase angle. Lenz uses mixer diodes to generate the third-order d i s t o r t i o n product, and controls the amount of d i s t o r t i o n by applying an external DC bias current through the diodes. A series of microwave integrated c i r c u i t (MIC) elements, such as couplers, pin diode 19 IF, IF £7 A low intercept-point © Mixer 2 X X IF, LO RF - coupl ing -network X Mixer 1 © © 1 X X -r TWT high intercept-point F i g . 2.5. Lenz I F P r e d i s t o r t i o n L i n e a r i z e r [13] ©L C1 delay line M^ 1 ® a- X OUT © X a, C2 C3 © © X a- 9 © F i g . 2.6. Lenz RF P r e d i s t o r t i o n L i n e a r i z e r [14] 20 attenuators, and a varactor phase s h i f t e r are used to tune the r a t i o of fundamental and d i s t o r t i o n products to the desired operating point. This p r e d i s t o r t i o n l i n e a r i z e r operates over a 100 MHz bandwidth and reduces the third-order IMD by 20 dB at a 10 dB output backoff. The use of these analog p r e d i s t o r t i o n techniques has been proven to be a cost e f f e c t i v e approach i n solving the HPA nonlinear problem i n analog t e r r e s t r i a l system. With only a moderate increase i n c i r c u i t complexity, large performance gains have been achieved. 2.1.2 D i g i t a l T e r r e s t r i a l Radio Applications In d i g i t a l t e r r e s t r i a l radio systems, the nonlinear behaviour of the f i n a l HPA has caused problems with m u l t i l e v e l QAM transmission. The amplitude and phase n o n l i n e a r i t i e s create d i s t o r t i o n i n the d i g i t a l s i g n a l c o n s t e l l a t i o n by compressing and r o t a t i n g the s i g n a l vectors. This e f f e c t increases the p r o b a b i l i t y of error i n the transmission. In addition, the sideband power i s increased due to the n o n l i n e a r i t y , generating adjacent channel in t e r f e r e n c e . Several d i f f e r e n t l i n e a r i z e r design approaches, incl u d i n g both analog and d i g i t a l techniques, have been used to solve these problems. Namiki [15] simulates the performance of an adaptive p r e d i s t o r t i o n l i n e a r i z e r on QAM transmission through a nonlinear TWTA. He develops an adaptive scheme which uses the o r i g i n a l Nojima et a l . p r e d i s t o r t i o n l i n e a r -i z e r [10] i n e i t h e r an IF or RF configuration. Namiki presents an algorithm which minimizes the third-order d i s t o r t i o n e f f e c t s by adaptively c o n t r o l l i n g the gain of the third-order d i s t o r t i o n generator i n the p r e d i s t o r t e r non-l i n e a r branch. At 12 dB input backoff, t h i s c o r r e l a t i o n detection method 21 provides a 10 dB reduction i n the sideband regeneration, and only a 0.5 dB c a r r i e r - t o - n o i s e (C/N) degradation from that obtainable with a perfect l i n e a r transmitter. An e f f e c t i v e analog technique was used by Holz [16] for improving both the sideband regeneration and BER degradation caused by the nonlinear transmit TWTA used i n a QAM system. Holz developed an analog IF pre d i s t o r t e r c i r c u i t which operates over a 60 MHz bandwidth centered at 140 MHz. He selected an IF approach i n order to use a sin g l e l i n e a r i z e r c i r c u i t to improve the performance of several HPAs operating at d i f f e r e n t transmit frequencies. The p r e d i s t o r t e r consists of three p a r a l l e l branches as shown i n F i g . 2.7. The nonlinear gain and phase compensation i s e s s e n t i a l l y provided by two l i m i t i n g amplifiers which saturate at d i f f e r e n t power l e v e l s . The summation of the three IF s i g n a l vectors at appropriate angles e f f e c t i v e l y reduce the d i s t o r t i o n caused by the f i n a l HPA. Ei t h e r upper or lower sidebands of the upconverted s i g n a l could be compensated by switchable s e l e c t i o n of a 180 degree phase s h i f t e r . The p r e d i s t o r t e r provides a two-tone third-order IMD reduction of 14 dB at 5.4 dB output backoff. The performance improvement for the actual QAM s i g n a l i s a 3.5 dB margin improvement at a BER of 10~ 8, with a 15 dB reduction i n sideband l e v e l . Nojima et a l . [17] use an analog RF p r e d i s t o r t i o n l i n e a r i z e r to improve the performance of a QAM 6 GHz t e r r e s t r i a l radio system which uses a 20 watt FET a m p l i f i e r . Their p r e d i s t o r t e r i s based on the scheme which was developed previously for a 6 GHz SSB a p p l i c a t i o n [10]. As shown i n F i g . 2.8b, the new p r e d i s t o r t e r operates d i r e c t l y at RF, and uses two small-signal amplifiers instead of diode pairs ( F i g . 2.8a) as the third-order d i s t o r t i o n causing elements. 22 upper sideband lower sideband Vector diagrams F i g . 2.7. H o l z I F P r e d i s t o r t i o n L i n e a r i z e r [16] 3rd-order distortion generator \ , -j /•/•/y J Variable attenuator 23 Hybrid B. Variable pnase-sfiifter Divider | i n j MYJ~ A 3 x 3 Low noise amollfler Continer r OutDUt Hybrid C ^ Delay line a.] RF predistortion circuit configuration. /Distort ion generator ( 1 Output L^ Circuit configuration of a 6-GHz band PD. F i g . 2.8. Nojima e t a l . RF P r e d i s t o r t i o n L i n e a r i z e r s [17] 24 Nojima et a l . have v e r i f i e d that t h e i r third-order d i s t o r t i o n phase analysis applied equally well to c e r t a i n s o l i d - s t a t e power a m p l i f i e r s . They use th i s a n a l y s i s to develop p r a c t i c a l operating margins for t h e i r p r e d i s t o r t e r by c a l c u l a t i n g the e f f e c t i v e reduction i n third-order IMD possible under various amplitude and phase mismatches between the pr e d i s t o r t e r and FET am p l i f i e r . This p r e d i c t i o n of performance improvement under nonoptimum conditions i s extremely important i n p r a c t i c a l systems, as perfect matching of the pre d i s t o r t e r and HPA n o n l i n e a r i t i e s i s d i f f i c u l t to achieve over large bandwidths. With t h e i r l i n e a r i z e r tuned for optimum wideband operation of 500 MHz, they achieve a substantial performance improvement. For a BER of 10 - 4 , an e f f e c t i v e increase of 2-3 dB i n output power i s possible with the FET HPA. For smaller bandwidth a p p l i c a t i o n s , they obtain higher (3-4 dB) output backoff improvement. The l i n e a r i z e r was adjusted i n both cases for maximum third-order IMD suppression using a two-tone c a r r i e r t e s t . A hybrid a n a l o g - d i g i t a l approach was proposed by Horn and Egger [18]. This 140 MHz IF p r e d i s t o r t i o n l i n e a r i z a t i o n scheme was used for a QAM 12 GHz t e r r e s t r i a l radio with e i t h e r TWTA or FET power a m p l i f i e r s . They i n i t i a l l y investigated the nonlinear c h a r a c t e r i s t i c s of both TWTA and FET power a m p l i f i e r s . They observed that although the amplitude and phase d i s t o r t i o n of the TWTA was much more pronounced than that of the FET, the FET amplitude and phase response was more complex. Their s o l u t i o n to the nonlinear problem was to propose two l i n e a r i z e r versions shown i n F i g . 2.9. The pre d i s t o r t e r for the TWTA would consist of a se r i e s analog network with independent d i g i t a l control used for amplitude and phase processing. The r e a l i z e d TWTA 25 l i n e a r i z e r consisted of a p a r a l l e l branch IF c i r c u i t with a va r i a b l e gain nonlinear branch c o n t r o l l e d by the d i g i t a l network. The important factor of bandwidth l i m i t a t i o n for the d i g i t a l c i r c u i t s i s emphasized. The clocking rate of the d i g i t a l c i r c u i t s i s 105 MHz, which i s s u f f i c i e n t to operate on the 45 MHz wide IF signal without an a l i a s i n g problem. The measured r e s u l t s of the TWTA l i n e a r i z e r indicated that the TWTA could operate at saturation with n e g l i g i b l e degradation i n the BER. Two-tone intermodulation tests confirmed that reduction of third-order by 25 dB to 45 dB with the l i n e a r i z e r . Two other independent research teams have attempted to minimize the ef f e c t s of HPA n o n l i n e a r i t i e s by using d i g i t a l l y c o n t r o l l e d p r e d i s t o r t e r s which operate on the baseband d i g i t a l s i g n a l . Davis et a l . [19], [20] have developed a patented adaptive d i g i t a l p r e d i s t o r t i o n l i n e a r i z e r f o r use i n QAM t e r r e s t r i a l radio. The main innovation i n t h e i r approach i s to provide an adaptive p r e d i s t o r t i o n function on the baseband in-phase (I) and quadrature (Q) data s i g n a l s . An adaptive comparative process continually updates the waveform data words stored i n random-access memory (RAM). These data words are subsequently converted to modulating analog waveforms. These p r e d i s t o r t -ed analog waveforms modulate the I and Q channel reference o s c i l l a t o r s which provide the required IF si g n a l at 70 MHz. This 70 MHz IF i s then upconverted to the f i n a l RF transmit frequency and amplified by the nonlinear HPA. A coupled RF output s i g n a l i s coherently demodulated and i s used to adaptively con t r o l the p r e d i s t o r t e r . Their scheme has provided s a t i s f a c t o r y BER performance at a TWTA output backoff of 1 dB. Post a m p l i f i e r s p e c t r a l I 26 IF 7l -w- AOC IF Predistortion network for TWTA's IF AM -w- ADC i t AOC MEM OAC MEM DAC PM IF Predistortion network for GaAs FETA's Fig. 2.9. Horn and Egger Hybrid Predistortion Linearizers [18] D A T A I N P U T M E M O R Y -L O O K U P E N C O D E R R A N D O M A C C E S S M E M O R Y TV (P.01 \7 L I N E A R I Z I N G P R O C E S S O R D I G I T A L / A N A L O G C O N V E R T E R 7 y P R E D I S T O R T E O I N P U T KI 0) to. .1 (ff.^ l A N A L O G / D I G I T A L C O N V E R T E R Q U A D R A T U R E M O D U L A T O R 0 O U A O R A T U R E D E M O D U L A T O R R A D I O F R E Q U E N C Y A M P L I F I E R Fig. 2.10. Saleh andSalz Adaptive Predistortion Linearizer [21] 27 shaping using waveguide f i l t e r s Is, however, required to reduce adjacent channel Interference. Saleh and Salz [21], [22] have also developed an adaptive p r e d i s t o r t i o n l i n e a r i z e r scheme for QAM transmission which i s shown i n F i g . 2.10. The authors use a d i g i t a l processor to p r e d i s t o r t the data waveform at baseband and, following d i g i t a l - t o - a n a l o g (D/A) conversion, modulate the transmitted c a r r i e r with the processed analog waveform. Saleh and Salz developed an algorithm for the adaptive process and present an a n a l y t i c a l proof of i t s convergence. The l i n e a r i z e d system expected performance i s obtained using simulation studies. Their approach i s i n p r i n c i p l e s i m i l a r to the scheme by Davis et a l . They conclude that adaptive p r e d i s t o r t i o n l i n e a r i z e r can allow operation of the TWTA at saturation, providing maximum power e f f i c i e n c y . In summary, both analog and d i g i t a l p r e d i s t o r t i o n techniques have shown promise i n various d i g i t a l t e r r e s t r i a l radio system. The use of adaptive techniques has overcome some of the p r a c t i c a l problems of maintaining a good match between the p r e d i s t o r t e r and HPA over a longer time period. 2.1.3 S a t e l l i t e Communications Applications As i n t e r r e s t r i a l systems, the n o n l i n e a r i t i e s of HPAs i n both earth stations and s a t e l l i t e transponders cause various forms of system impairments i n s a t e l l i t e communication a p p l i c a t i o n s . Although recent improvements i n SSPA technology are allowing s u b s t i t u t i o n of more l i n e a r FET power amplifiers i n place of lower powered TWTAs i n c e r t a i n 6 GHz (C band) and 14 GHz (Ku band) a p p l i c a t i o n s , the f i n a l HPA Is s t i l l most often e i t h e r a TWTA or 28 kl y s t r o n a m p l i f i e r . The predominant e f f e c t s caused by the TWTA and kl y s t r o n near saturation, and to a lesser extent by the SSPA, include intermodulation d i s t o r t i o n i n both analog and d i g i t a l FDMA systems, such as SCPC, and intersymbol interference and sp e c t r a l spreading i n d i g i t a l s i n g l e - c a r r i e r TDMA ap p l i c a t i o n s . Attempts to reduce these e f f e c t s have generated considerable i n t e r e s t i n p r e d i s t o r t i o n l i n e a r i z e r technique development. Several design approaches, consisting of either RF or baseband p r e d i s t o r t i o n , have been investigated, and s i g n i f i c a n t improvements i n system performance and HPA e f f i c i e n c y have been achieved. 2.1.3.1 RF L i n e a r i z a t i o n In both present earth s t a t i o n and future s a t e l l i t e transponder ap p l i c a t i o n s , p r e d i s t o r t i o n l i n e a r i z a t i o n development has focused on analog RF signal processing i n a stage immediately p r i o r to the f i n a l HPA. Most work focused on techniques for the l i n e a r i z a t i o n of large INTELSAT earth stations and transponder HPAs where considerable power savings are of prime concern. A major e f f o r t i n earth s t a t i o n l i n e a r i z a t i o n was f i r s t reported by Bremenson and Jaubert [23] . They developed a p a r a l l e l branch RF predistor-t i o n l i n e a r i z e r f o r use at an INTELSAT IV m u l t i c a r r i e r FDM earth s t a t i o n which operated i n the 6 GHz uplink band. The pr e d i s t o r t e r consisted of a li n e a r path and nonlinear path and i s shown i n F i g . 2.11. The l i n e a r path included a va r i a b l e attenuator and phase s h i f t e r . The nonlinear path consis-ted of an a n t i p a r a l l e l diode pair as the nonlinear d i s t o r t i o n generator. The combination of the two si g n a l vectors provided both an amplitude expansion and phase advance c h a r a c t e r i s t i c s which e f f e c t i v e l y reduced the third-order IMD of the 6 GHz TWTA. The l i n e a r i z e r provided over 3 dB improvement i n 29 F i g . 2 . 1 1 . B remenson and J a u b e r t RF P r e d i s t o r t i o n L i n e a r i z e r [23] 30 output backoff for a third-order IMD operating l e v e l of 25 dBc with several TWTAs. A further a p p l i c a t i o n of th i s l i n e a r i z e r was reported by Bremenson and Lombard [24]. In t h i s study, they examined the performance of the l i n e a r i z e r In both an earth s t a t i o n and s a t e l l i t e transponder a p p l i c a t i o n f o r phase-shift-keying (PSK) TDMA s i g n a l s . With t h e i r diode p r e d i s t o r t e r , they observed both an improved BER performance and reduction i n sideband regeneration at various operating points of an earth s t a t i o n TWTA and the Symphonie s a t e l l i t e transponder TWTA. Bremenson et a l . [25] reported a thorough i n v e s t i g a t i o n of the advantages of applying p r e d i s t o r t i o n l i n e a r i z e r s to a future INTELSAT transponder for both FDMA and TDMA systems. They used a diode p r e d i s t o r t e r modified to l i n e a r i z e a 4 GHz TWTA. For m u l t i c a r r i e r operation, they determined that the l i n e a r i z e r provided best o v e r a l l reduction of third-order IMD when optimized at 5 dB TWTA output backoff. At a two-tone third-order IMD l e v e l of 25 dBc, they observed a 3.8 dB improvement i n output power over the f u l l 500 MHz bandwidth. The f i f t h - o r d e r IMD, although increased with the l i n e a r i z e r , was maintained at acceptable system l e v e l s . The o v e r a l l AM/PM d i s t o r t i o n was reduced from a maximum of 40 degress phase s h i f t at saturation, to below 10 degrees with the l i n e a r i z e r . This design approach i s a t t r a c t i v e since i t uses simple c i r c u i t structure with r e l a t i v e l y low component cost, and achieves good performance. The long term s t a b i l i t y aspects of s a t e l l i t e transponder (on-board) l i n e a r i z e r s , which i s of prime importance since readjustment due to TWTA aging i s not possible, was investigated by Cahana [26]. This study discussed 31 the performance of the Bremenson et a l . diode p r e d i s t o r t i o n l i n e a r i z e r under simulated and actual operating conditions. These r e s u l t s i n d i c a t e that some degradation of performance over time i s i n e v i t a b l e , but t h i s degradation can be minimized i f the l i n e a r i z e r i s c a r e f u l l y tuned and optimized for a s p e c i f i c TWTA. The f e a s i b i l i t y of using on-board l i n e a r i z a t i o n was also investigated by Charas and Rogers [27]. They experimentally attemped to determine the IMD improvement possible with a d i f f e r e n t l i n e a r i z a t i o n scheme. The l i n e a r i z e r tested operated at a lower frequency of 925 MHz, where component technology was a v a i l a b l e to test the l i n e a r i z a t i o n concept for Ku band TWTAs. They used l i n e a r up and down converters to translate the predistorted s i g n a l to the f i n a l TWTA operating frequency. The l i n e a r i z e r consisted of two independent stages, the f i r s t providing amplitude expansion, the second phase advance. Both stages used bias c o n t r o l l e d diode elements to generate the desired response. For m u l t i c a r r i e r FDM/FM systems, a third-order IMD was improved from 10 dBc to 16 dBc at two-tone saturation. For PSK/TDMA systems, an improvement i n BER was also achieved. This corresponded to 0.5 dB improvement i n c a r r i e r - t o - n o i s e r a t i o at saturation. This i n v e s t i g a t i o n also proposed the use of l i m i t e r s following the pr e d i s t o r t e r to prevent overdrive of the TWTA. This l i m i t e r concept was l a t e r used to considerable advantage i n other l i n e a r i z a t i o n schemes. A major independent p r e d i s t o r t i o n l i n e a r i z a t i o n development was repor-ted by Satoh [28]. Satoh f i r s t reported h i s i n v e s t i g a t i o n of l i n e a r i z i n g a 6 GHz 12 KW coupled-cavity type TWTA used i n an INTELSAT earth s t a t i o n . This l i n e a r i z e r consisted of a simple p a r a l l e l branch c i r c u i t a r c h i t e c t u r e , 32 with a l i n e a r delay l i n e path and a FET t r a n s i s t o r a m p l i f i e r i n the nonlinear path. The two s i g n a l paths are combined i n antiphase at a preset combining r a t i o . This c i r c u i t i s s i m i l a r i n p r i n c i p l e to Heun and K i e s e l ' s ( F i g . 2.1) IF p r e d i s t o r t i o n l i n e a r i z e r [6]. As the drive into the l i n e a r i z e r i s increased, the FET gain decreases and the i n s e r t i o n phase decreases. The sum of the two s i g n a l vectors at the combiner output imposes the desired ampli-tude expansion and phase advance c h a r a c t e r i s t i c . The l i n e a r i z e r reduced the f o u r - c a r r i e r IMD by 10 dB at 6 dB output backoff over a 500 MHz bandwidth. Kurokawa et a l . [29] continued work on Satoh's p r e d i s t o r t i o n l i n e a r -i z e r . They provided a s i m p l i f i e d power serie s analysis of the nonlinear transfer function of the p r e d i s t o r t e r and HPA. They confirmed a n a l y t i c a l l y that the p a r a l l e l branch structure provided the desired amplitude expansion and phase advance c h a r a c t e r i s t i c required for third-order IMD reduction. The pr e d i s t o r t e r was then used to l i n e a r i z e h e l i x and coupled-cavity TWTAs, and a k l y s t r o n . S i g n i f i c a n t improvements i n third-order m u l t i c a r r i e r IMD were observed using the wideband h e l i x tube operating at 7 dB output backoff. Over 10 dB of IMD reduction with the coupled-cavity tube was also measured, but amplitude and phase equalizers were required between the l i n e a r i z e r and TWTA. The k l y s t r o n exhibited a very narrowband response, as IMD reduction was possible only over a 40 MHz band using an amplitude and phase equalizer. A further report by Satoh et a l . [30] presented a d d i t i o n a l long term IMD performance r e s u l t s using both a FET l i n e a r i z e r and a diode l i n e a r i z e r with various HPAs. The p a r a l l e l branch diode l i n e a r i z e r operates i n a s l i g h t l y d i f f e r e n t manner, combining the l i n e a r and nonlinear signals at 90 33 degress as done by Bremenson [23] . The third-order m u l t i c a r r i e r IMD reduction c a p a b i l i t y of both types of l i n e a r i z e r s were measured to be equivalent. A refinement to the FET l i n e a r i z e r was also reported by Satoh et a l . [31]. This modified l i n e a r i z e r included a double loop c i r c u i t which contain-ed the basic FET l i n e a r i z e r , v a r i a b l e attenuator, and phase s h i f t e r i n one p a r a l l e l path, and a delay l i n e i n the other l i n e a r path. The provision of the double loop provided a f i n e r adjustment c a p a b i l i t y , as the s i g n a l vectors could be combined over a larger dynamic range, providing a closer inverse match c h a r a c t e r i s t i c to the HPA. This improved performance, however, was obtained by an increase i n c i r c u i t complexity. The next major innovation i n the FET l i n e a r i z e r , was the addition of a separate l i m i t e r c i r c u i t which was inserted immediately a f t e r the pr e d i s t o r t e r stage. This improvement, reported by Satoh et a l . i n l a t e r work [32-33], provided a s i g n i f i c a n t performance improvement for l i n e a r i z e d HPAs operating at saturation i n both FDMA and TDMA systems. The new single loop l i n e a r i z e r was also modified by replacing the delay l i n e In the main l i n e a r s i g n a l path with a FET. This FET i s s i m i l a r to the FET used i n the nonlinear path. To maintain a l i n e a r main path, the FET i s biased for operation i n i t s l i n e a r region. By adjustment of DC bias to both FETs, and by employing a varactor phase s h i f t e r , the l i n e a r i z e r was tuned e l e c t r o n i c a l l y . The two si g n a l vectors could be combined i n various power r a t i o s and phase angles, r e s u l t i n g i n f l e x i b i l i t y i n tuning for best performance with several TWTAs. A double loop version of the l i n e a r i z e r with the l i m i t e r was also suggested. These improved l i n e a r i z e r s are shown i n F i g . 2.12 and F i g . 2.13. . 34 L R Z V ' — ATT Limiter amplifier S L - L R Z TWTA (a) Block diagram of S L - L R Z 71 Circuit lb) Linearizer circuit F i g . 2.12. Satoh e t a l . RF P r e d i s t o r t i o n L i n e a r i z e r [33] ATT Phase shifter L R Z Input Divider •^nrp-Oeloy-circuit Combiner Output F i g . 2.13. Satoh e t a l . Double Loop P r e d i s t o r t i o n L i n e a r i z e r [33] 35 The l i n e a r i z e r with l i m i t e r was developed for operation at both 6 GHz and 4 GHz, and was designed for operation with both earth s t a t i o n and future INTELSAT transponder TWTAs. The l i n e a r i z e r was tested with both m u l t i c a r r i e r noise-loading and s i n g l e - c a r r i e r quadrature-phase-shift-keying (QPSK) sig n a l s . The noise-power-ratio (NPR) r e s u l t s showed a 5 dB improvement i n output power of the s a t e l l i t e TWTA while maintaining a NPR of 25 dB. For the QPSK s i g n a l , the sideband power density was maintained at 26 dB below the sig n a l density at an output backoff of 1 dB. A 3 dB output backoff was required to maintain the same power density r a t i o without l i n e a r i z a t i o n . S i g n i f i c a n t improvement i n BER was measured with the l i n e a r i z e r over a large dynamic range. The l i n e a r i z e r was also r e l a t i v e l y i n s e n s i t i v e to gain s t a b i l i t y . A v a r i a t i o n i n gain of the TWTA of ± 1 dB was not detrimental to l i n e a r i z e r performance. The combined l i n e a r i z e r and l i m i t e r scheme shows promise as a v e r s a t i l e technique, as improvements i n both FDMA and TDMA systems are possible with only moderate c i r c u i t complexity. A wideband 6 GHz coupled cavity l i n e a r i z e d TWTA has been reported by Chakraborty and Kappes [34], [35]. Their wideband l i n e a r i z a t i o n test i n c l u d -ed a double loop l i n e a r i z e r developed by Satoh et a l . [31]. By c a r e f u l l y adjusting the slope and equalizing the phase c h a r a c t e r i s t i c s between the l i n e a r i z e r and TWTA, the o v e r a l l l i n e a r i z e d system operated with a modest IMD reduction of from 2-6 dB over the 500 MHz bandwidth. Further t e s t i n g with the l i n e a r i z e r and a 3 KW 6 GHz klystron for a QPSK TDMA system was reported by Chakraborty and McCune [36]. Improvements i n the BER were observed at 2-4 dB input backoff compared to the klystron without the l i n e a r i z e r . 36 A novel RF p r e d i s t o r t i o n scheme has recently been reported by Kumar et a l . [37], [38]. As shown i n F i g . 2.14a, the approach taken i n t h i s scheme i s the a p p l i c a t i o n of a pa i r of e l e c t r o n i c a l l y c o n t r o l l e d dual-gate FETs i n a p a r a l l e l branch network. The dual-gate FETs are biased near pinchoff, i n order to generate gain expansion c h a r a c t e r i s t i c s In each path. The bias voltages are s l i g h t l y o f f s e t i n order to cause a differe n c e i n the nonlinear gain between the two devices. The phase and amplitude diffe r e n c e between the two s i g n a l vectors at the combiner provide the desired resultant vector expansion and phase r o t a t i o n as shown i n F i g . 2.14b-e. The i n d i v i d u a l FETs have an i n s e r t i o n loss of approximately 20 dB, with the o v e r a l l l i n e a r i z e r e x h i b i t i n g a loss of 30-35 dB. The l i n e a r i z e r loss i s compensated by a li n e a r post a m p l i f i e r . Two separate l i n e a r i z e r s were developed and tested with a 12 GHz TWTA, and a 4 GHz SSPA. The TWTA l i n e a r i z e r improved the over-a l l AM/PM phase s h i f t from 45 degrees to less than 14 degrees over a 500 MHz bandwidth. The two-tone third-order IMD was reduced 7 dB over a 10 dB dyna-mic range. The l i n e a r i z e r also performed well over a 15 to 55 degrees C temperature range, but required s l i g h t changes i n gate bias voltages to main-t a i n performance over the required temperature range. Kumar et a l . suggest that sensistors be used i n the bias c i r c u i t s to provide optimum bias voltage at d i f f e r e n t temperatures. The 4 GHz l i n e a r i z e r was tested with the SSPA and provided s i m i l a r improvements i n IMD performance over 400 MHz bandwidth. The AM/PM of the SSPA was reduced from 8 degrees to 4 degrees at saturation. The l i n e a r i z e r could thus be adjusted for best performance depending on s i g n a l t r a f f i c type. Some reduction of higher-order IMD d i s t o r t i o n was also observed using e i t h e r l i n e a r i z e r . This l i n e a r i z e r i s a t t r a c t i v e from both a IN» 37 TWTA POST AMPLIFIER (LINEAR) DUAL GATE FET (a) (a) Schematic diagram of the dual-gate FET linearizer. a (c) (b)-(c) Linearizer vector diagrams. F i g . 2.14. Kumar e t a l . RF P r e d i s t o r t i o n L i n e a r i z e r [38] 38 performance and t u n a b i l i t y perspective. I t s main disadvantage appears as a r e l a t i v e l y high i n s e r t i o n l o s s , which requires use of c o s t l y post a m p l i f i e r s . This cost could be reduced by i n v e s t i g a t i o n of th i s technique at IF. In summary, the analog RF l i n e a r i z a t i o n scheme reported for s a t e l l i t e earth s t a t i o n and transponder a p p l i c a t i o n can be c l a s s i f i e d into three categories. These are the p a r a l l e l path diode l i n e a r i z e r , the p a r a l l e l path FET a m p l i f i e r l i n e a r i z e r with and without a l i m i t e r , and the p a r a l l e l path dual-gate FET var i a b l e attenuator l i n e a r i z e r . A l l techniques show low to moderate c i r c u i t complexity, with moderate r e a l i z a t i o n costs. Further s a t e l l i t e system studies using these p r e d i s t o r t i o n type l i n e a r i z e r s have also been reported by Gray et a l . [39], and Welti et a l . [40]. S i g n i f i c a n t t h e o r e t i c a l work rel a t e d to optimum p r e d i s t o r t i o n l i n e a r i z a t i o n , with both modeling and simulation r e s u l t s , have been reported by other researchers [41-44]. 2.1.3.2 Baseband L i n e a r i z a t i o n A new baseband l i n e a r i z e r scheme was developed by Girard and Feher [45], [46]. This technique has indicated p o t e n t i a l for improving HPA e f f i c i e n c y for a QPDK TDMA system while meeting the INTELSAT V performance s p e c i f i c a t i o n with the earth s t a t i o n HPA operating near saturation. Girard and Feher have simulated the performance of a l i n e a r i z e r which p r e d l s t o r t s the baseband I and Q data channels before the QPSK modulator. A block diagram of t h e i r system i s shown i n F i g . 2.15. The proposed system consists of a low pass f i l t e r which bandlimits the input data streams. The envelope power of the f i l t e r e d s i g n a l i n each channel i s detected continuously, and an analog form of look up tables are used to co n t r o l a pair of var i a b l e 39 Premodulation lowpass f i l t e r s Envelope p r e d i s t o r t i o n Phase P r e d i s t o r t i o n pass f i l t e r Low pass f i l t e r ( ) Ins lanlon»ou« Input Envelope'* Power Detection V a r i a b l e Attenuator V a r i a b l e Attenuator COS 9 S l t l Q smd cos 8 Nonlinear Look.up Table f o r Envelope P r e d i s t o r t i o r r 180' 0" Non L i n e a r Look up Table for Phase P r e d i s t o r t i o n I • sm 0 • cos 0 Conventional QPSK Modulator 90' A / Fig. 2.15. Girard and Feher Baseband Predistortion Linearizer [46] 40 attenuators and phase s h i f t e r s . The attenuator look up table provides the inverse amplitude response of the HPA, and the phase s h i f t table provides the opposite phase r o t a t i o n of the s i g n a l vector to correct for the AM/PM d i s t o r t i o n caused by the HPA. Their simulated computer r e s u l t s for a s i n g l e -c a r r i e r QPSK s i g n a l i n d i c a t e an improvement i n sideband regeneration, and better performance under adjacent cochannel interference conditions. An improvement of over 3 dB i n output backoff to just below saturation i s predicted with the baseband l i n e a r i z e r . This l i n e a r i z a t i o n scheme requires moderate c i r c u i t complexity, and could be applied i n p r a c t i c a l systems. Hybrid systems, which use HPAs to amplify m u l t i c a r r i e r FDM and TDM s i g n a l s would not appear to be s u i t a b l e for such a baseband technique, as a composite s i g n a l p r e d i s t o r t e r would be required. 2.2 Other L i n e a r i z a t i o n Techniques Although p r e d i s t o r t i o n has been proven to be the most p r a c t i c a l method for l i n e a r i z a t i o n of microwave HPAs, other techniques have been implemented i n s p e c i a l a p p l i c a t i o n s with some success. The two main approaches which have been reported use feed-forward or negative feedback schemes to l i n e a r i z e power a m p l i f i e r s or converter stages. 2.2.1 Feed-Forward L i n e a r i z a t i o n This technique was i n i t i a l l y reported by Seidel [47], and was followed by further work by Bakken [48], [49], and Meyer et a l . [50]. System studies using feed-forward were also reported by Chakraborty [51] and Javed et a l . [52]. The basic p r i n c i p l e behind t h i s technique involves the use of two i d e n t i c a l HPAs for d i s t o r t i o n reduction. Seidel*s approach i s shown i n F i g . 2.16. The input s i g n a l i s i n i t i a l l y s p l i t i n t o two paths using an input M A I N AMPLIFIER Co(l+*) 41 G 0 ( i - « « ) AUXILIARY AMPLIFIER (a) single stage - * G 0 ( i + c) G0['+o(<3)] F i g . 2.16. S e i d e l Feed-Forward L i n e a r i z e r [47] COUPLER 1 DELAY c EQUALIZER IT MAIN AMPLIFIER COUPLER 2 A; PHASE (ft |^ A D J U S T V L / I ^ LOOP AMPLIFIER W AMPLITUDE A ADJUST CANCELLATION COMBINER F i g . 2.17. Gajda and D o u v i l l e Negative Feedback L i n e a r i z e r [57] 42 coupler, where one path Includes the main am p l i f i e r (HPA//1), and the other consists of a l i n e a r path with delay equal to the nominal i n s e r t i o n delay of HPA#1. The output of HPA/i1! i s then s p l i t , where one path i s attenuated by the nominal gain of HPA#1, and t h i s s i g n a l i s then summed with the output from the o r i g i n a l delay l i n e . The other path following HPA#1 consists of a separate delay l i n e . The summed output of HPA//1 and the o r i g i n a l delay l i n e provides input to the a u x i l i a r y a m p l i f i e r (HPA#2), which has output power c h a r a c t e r i s t i c s which are s i m i l a r to HPA#1. The output of HPA//2 i s then summed with the s i g n a l from the second delay l i n e producing the f i n a l output. It has been demonstrated that considerable d i s t o r t i o n reduction i s possible using t h i s technique. I t s main advantage i s the f a c t that no feedback loops are required and s t a b i l i t y i s assured. This technique has, however, several disadvantages as two large HPAs are required, and f a i r l y t i g h t performance s p e c i f i c a t i o n s are required for the delay l i n e s and couplers. Presently t h i s i economic disadvantage deters use of t h i s technique i n commercial communica-tions systems. 2.2.2 Negative Feedback L i n e a r i z a t i o n This technique has found a p p l i c a t i o n s i n low frequency s o l i d state am p l i f i e r l i n e a r i z a t i o n and upconverter stages. The basic concept behind negative feedback l i n e a r i z a t i o n employs c l a s s i c a l c o n t r o l theory to reduce the nonlinear behaviour of a m p l i f i e r s . At microwave frequencies, however, the time delay encountered i n the feedback loops i s almost i n v a r i a b l y equal to several cycles of RF s i g n a l . This leads to s t a b i l i t y problems i n the con t r o l loop which has deterred i t s use with slow-wave devices such as TWTAs where t r a n s i t times through the device are s i g n i f i c a n t . Special cases of 43 feedback techniques, however, have been investigated using s o l i d - s t a t e devices. Blanke et a l . [53] have attempted to l i n e a r i z e a s o l i d state amplfier using negative feedback which operates on the envelope of the s i g n a l , as opposed to the feedback of the instantaneous RF output of the a m p l i f i e r . They use two independent loops for t h e i r intermodulation reduction scheme. One feedback loop i s used to co n t r o l an amplitude modulator and the other loop i s used to c o n t r o l a phase modulator. Loser and Schunemann [54], [55] attempt to replace the use of HPAs i n c e r t a i n a p p l i c a t i o n s by i n v e s t i g a t i n g high power upconverters using varactor diodes. They use negative feedback techniques to obtain s a t i s f a c t o r y intermodulation performance. Weaver and Taylor [56] use a feedback and demodulation scheme to e f f e c t i v e l y feedback the baseband modulated s i g n a l for error comparison. They propose to use t h e i r approach i n a m u l t i l e v e l d i g i t a l radio system using s o l i d - s t a t e FET power a m p l i f i e r s . Gajda and Douvi l l e [57] use a hybrid negative feedback and feed-forward intermodulation reduction scheme, but l i m i t t h e i r study to a VHF s o l i d - s t a t e power ampli f i e r where RF feedback time delays are small enough so as not to cause i n s t a b i l i t y . Their approach i s shown i n F i g . 2.17. Perez et a l . [58] have proposed another feedback scheme, where they develop active feedback at 1 GHz using a t r a n s i s t o r a m p l i f i e r i n the feedback loop. 2.3 Comparative Summary Several reported l i n e a r i z a t i o n schemes have been reviewed with an emphasis on commercial system a p p l i c a t i o n s . The p r e d i s t o r t i o n technique i s the favoured approach at present, showing considerable performance gains i n both analog and d i g i t a l systems. Feed-forward and negative feedback schemes, 44 however, have c e r t a i n desirable c h a r a c t e r i s t i c s i n s p e c i a l i z e d a p p l i c a t i o n s . A l l three concepts w i l l now be c l a s s i f i e d i n terms of common hardware components, and the most appropriate design approach for an SCPC TWTA ap p l i c a t i o n w i l l be selected. 2.3.1 C l a s s i f i c a t i o n of L i n e a r i z e r C i r c u i t s As shown i n the block diagram i n F i g . 2.18, each main l i n e a r i z a t i o n technique can be c l a s s i f i e d i n terms of hardware and c i r c u i t implementation. The feed-forward approach can be generally c l a s s i f i e d as an analog s i g n a l processor, with implementation d i r e c t l y at RF using an a u x i l i a r y HPA. The negative feedback approach has been implemented i n a few d i f f e r e n t c i r c u i t structures. The feedback control s i g n a l can operate on the instantaneous RF s i g n a l , the modulated envelope, or the demodulated baseband s i g n a l . The RF feedback c i r c u i t has been used for both upconverters and t r a n s i s t o r ampli-f i e r s using active feedback networks. The envelope detection technique has been developed for feedback control of eit h e r an amplitude or phase modulator preceding a SSPA. The d i g i t a l demodulation scheme has been applied to process the d i g i t a l baseband signal before modulation. The p r e d i s t o r t i o n technique has been implemented using a v a r i e t y of s i g n a l processing c i r c u i t s . Analog p r e d i s t o r t i o n s i g n a l processing has been accomplished at baseband, IF, or RF. The IF and RF processors have used eit h e r diodes or FETs as the nonlinear c i r c u i t elements. A hybrid approach used high speed d i g i t a l c i r c u i t s to provide an analog c o n t r o l for amplitude or phase modulators. The d i g i t a l schemes used d i g i t a l processors to modify the baseband data sequences before modulation and a m p l i f i c a t i o n by the HPA. An adaptive co n t r o l algorithm has been included i n both an analog and d i g i t a l c i r c u i t implementation. L inear izat ion Basic Feed-forward Concept: l Control: Predistortion Frequency: Circuit: Negative Feedback Analog Analog Hybrid Digital i i l Hybrid (Demod) Analog (Direct) Analog (Envelope Detection © ( ^ 6 ©0 © © © © © Aux HPA Diode FET 2 FETs Logic Ram Analog Look-up Tables Logic Ram Comparator Upconverter Aux. Amp Comparator F i g . 2 . 1 8 . C lass i f i ca t ion D iagram o f L i n e a r i z a t i o n Techniques 46 2.3.2 P r a c t i c a l Considerations As discussed i n the previous sections on l i n e a r i z a t i o n technique development, l i n e a r i z a t i o n i s a powerful technique which can be used to great advantage i n a v a r i e t y of analog and d i g i t a l communication systems. With t h i s myriad of approaches available for HPA l i n e a r i z a t i o n , however, the design problem of f i n d i n g a su i t a b l e l i n e a r i z a t i o n scheme for a small single-channel-per-carrier (SCPC) earth s t a t i o n s t i l l remains. A simple design r u l e , low component and manufacturing cost, temperature s t a b i l i t y , and r e p e a t a b i l i t y i n performance are p r a c t i c a l constraints i n the f i n a l s e l e c t i o n of a design approach. A system which may work well i n a laboratory environment, may cause problems when integrated into an operating system. P r e d i s t o r t i o n , with i t s f l e x i b i l i t y and proven performance i s the preferred choice for the SCPC a p p l i c a t i o n . The high cost of feed-forward and time delay problems with negative feedback r u l e out these approaches. The SCPC earth s t a t i o n TWTA w i l l be used to amplify a FDM m u l t i c a r r i e r s i g n a l , where third-order IMD i s the system performance c r i t e r i o n . The technical problems with c i r c u i t design and manufacture at Ku band, i n d i c a t e that an IF pre d i s t o r t e r may provide a lower cost approach. Adequate gain s t a b i l i t y between the pr e d i s t o r t e r and the TWTA, however, would be required. Appro-p r i a t e f i l t e r i n g for an IF pr e d i s t o r t e r i s also required. For m u l t i c a r r i e r s i g n a l s , a passband of three s i g n a l bandwidths i s necessary to completely pass the predistorted s i g n a l spectrum. P o t e n t i a l use with other transmit bands i s also d e s i r a b l e . The f i n a l s e l e c t i o n of the Bremenson et a l . p a r a l -l e l branch diode l i n e a r i z e r [23-25] approach with a new IF implementation was based on these and other f a c t o r s . To the author's knowledge, the use of an 47 IF p r e d i s t o r t i o n l i n e a r i z e r for s a t e l l i t e earth stations i s new. A comparative summary of the advantages and disadvantages of the IF vs the RF approach i s shown i n Table 2.1. TABLE 2.1 IF/RF P r e d i s t o r t i o n L i n e a r i z e r Feature Summary IF RF Percentage Bandwidth of C i r c u i t high (> 10%) low (< 10%) Interstage gain s t a b i l i t y requirements high low Interstage f i l t e r i n g considerations yes no Implementation d i f f i c u l t y low high MIC s i z e average small Reuse c a p a b i l i t y f o r other transmit bands yes no L i n e a r i z e r component cost low high Post l i n e a r i z e r a m p l i f i e r cost low high 49 3. NONLINEAR DEVICE MODELING Although the focus of t h i s study i s on TWTA l i n e a r i z a t o n f o r single-channel-per-carrier (SCPC) earth s t a t i o n a p p l i c a t i o n s , i t i s useful to summarize the development of several nonlinear models for HPAs and examine t h e i r use i n developing various l i n e a r i z a t i o n schemes. Before p r a c t i c a l l i n e a r i z e r s could be r e a l i z e d , researchers required t r a c t a b l e a n a l y t i c a l models which described the nonlinear behaviour of HPAs. Their i n i t i a l e f f o r t concentrated on the development of models for TWTAs, as these amplifiers were most often used i n microwave communication systems. Several models have been developed for the simulation of HPA performance i n many communication system ap p l i c a t i o n s . Three of the main HPA modeling approaches are b r i e f l y described. 3.1 Amplitude-Phase Bandpass Nonlinear Model The nonlinear c h a r a c t e r i s t i c s of TWTAs are most often measured by observing the amplitude compression and phase s h i f t imposed on a single unmodulated c a r r i e r amplified by the TWTA. This technique provides a measurement of the envelope response of the TWTA, referred to as the AM/AM amplitude and AM/PM phase response of the device. This bandpass response y(t) can be generally modeled as a function of the envelope of the input s i n u s o i d a l s i g n a l x ( t ) . Using Saleh's [59] notation we can write x(t) = r ( t ) cos (u) 0t + <|>(t)) (3.1) where u 0 i s the c a r r i e r frequency, and r ( t ) and t\>(t) are the modulated envelope and phase r e s p e c t i v e l y . The bandpass output y(t) i s 50 y(t) = A ( r ( t ) ) cos(o) 0t + <p(t) + «(r(t))) (3.2) where A ( r ( t ) ) i s an odd function of r ( t ) , with a l i n e a r leading term, and $ ( r ( t ) ) i s an even function of r ( t ) , with a quadratic leading term. In t h i s model A represents the AM/AM conversion function and $ represents the AM/PM conversion function. Several a n a l y t i c a l formulas have been used to represent A and For narrowband systems, where the s i g n a l bandwidth i s s i g n i f i c a n t l y less than the bandwidth of the TWTA, A and $ are considered to be frequency independent, and the device i s ref e r r e d to as " e f f e c t i v e l y memoryless" [60], The f i r s t TWTA models considered the TWTA as a s t r i c t l y "memoryless" device with no amplitude dependent phase s h i f t . This approximate model implies that the AM/PM e f f e c t i s nonexistent, as the function y(t) becomes y(t) = A ( r ( t ) ) cos ( u 0 t + <|,(t)). (3.3) The piece-wise instantaneous l i m i t e r used by Sunde [61] i s an example of such a s i m p l i f i e d approximate model for the TWTA. This amplitude-only nonlinear model was used for a n a l y t i c convenience and computational e f f i c i e n c y . Although such a s i m p l i f i e d model has Its l i m i t a t i o n s , i t d i d provide a useful t o o l i n estimating the degree of intermodulation d i s t o r t i o n which could be expected when operating TWTAs i n t h e i r nonlinear regions. The requirement for more accurate predictions of intermodulation d i s t o r t i o n generated by TWTAs necessitated that both amplitude and phase d i s t o r t i o n s be taken into account. The work by Shimbo [62], and Fuenzalida et a l . [63] showed a rigorous time-domain analysis of intermodulation d i s t o r t i o n using the amplitude-phase model. 3.2 Bandpass Quadrature Nonlinear Model The model which has proved most useful i n the analysis of TWTAs i s the 51 narrowband quadrature model developed by Kaye et a l . [41]. This model separates the TWTA amplitude (AM/AM) and phase (AM/PM) c h a r a c t e r i s t i c s i n t o two amplitude transfer functions, which are frequency independent i n the narrowband case. The actual TWTA nonlinear amplitude and phase c h a r a c t e r i s t i c s can also be obtained by using a single-tone to measure the gain compression and phase from low power l e v e l s up to and past the satuation point of the device. Both the previous amplitude-phase and the quadrature model use the measured amplitude and phase c h a r a c t e r i s t i c s of the HPA to determine functions of the envelope or rms power of the input single-tone s i g n a l , rather than the instantaneous s i g n a l voltage. In the quadrature model, these measured envelope and phase transfer c h a r a c t e r i s t i c s can be modeled by two quadrature functions which are dependent only on the amplitude of the s i g n a l . The corresponding instantaneous amplitude function, i f desired, can be obtained for each channel of the model by applying the inverse Chebyshev transformation [64]. As shown i n F i g . 3.1, with input s i g n a l x(t) given by (3.1), the output s i g n a l of the model y(t) can be expressed a n a l y t i c a l l y as a sum of the in-phase and quadrature components. Using Saleh's notation [59], the two si g n a l components are where P and Q are odd functions of the envelope r ( t ) , with l i n e a r and cubic leading terms r e s p e c t i v e l y . The summed output i s p(t) = P [ r ( t ) ] cos [ u 0 t + <|»(t)] (3.4) q(t) = -Q[r(t)] s i n [ u Q t + (|»(t)] (3.5) y(t) = p(t) + q ( t ) . (3.6) This r e s u l t i s rela t e d to the amplitude-phase model by 52 P(t) - P [ r ( t ) ] cos [u) 0t + (J,(t)] p.() I N P U T S I G N A L x(t) = r ( t ) cos (a)0t + <KO) q(t) = -<3[r(t)] s i n [a) 0t + <j,(t)] 0 O U T P U T S I G N A L 90» Q < ) y ( t ) - P ( t ) + F i g . 3.1. Quadrature Model of TWTA 53 P(r) = A(r) cos $ (3.7) Q(r) = A(r) s i n «. (3.8) Using the envelope quadrature model, Kaye et a l . have a n a l y t i c a l l y described the required c h a r a c t e r i s t i c s of a p r e d i s t o r t i o n l i n e a r i z e r . Their p r e d i s t o r t i o n l i n e a r i z e r i s also modeled as a nonlinear quadrature device, and the model i s used to describe a p o t e n t i a l r e a l i z a t i o n of the l i n e a r i z a t i o n scheme. Hetrakul and Taylor [65,42,3] continued work using the quadrature model and described optimum p r e d i s t o r t e r c h a r a c t e r i s t i c s which are dependent on the input s i g n a l c a r r i e r - t o - n o i s e r a t i o . In re l a t e d work, E r i c [66] has developed a technique for c a l c u l a t i n g the intermodulation d i s t o r t i o n of TWTAs under m u l t i c a r r i e r conditions using the envelope transfer functions of the quadrature model. E r i c uses a multiterm odd-order power se r i e s to represent the in-phase and quadrature amplitude functions. Related studies were also done by Saleh [59] and Abuelma'atti [67] where they expanded the use of the quadrature model by i n v e s t i g a t i n g the e f f e c t s of frequency dependence on the model. Kaye et a l . [41] and Saleh [44], use the quadrature model to predict the performance of a third-order l i n e a r i z e r and a t y p i c a l TWTA, which provide a good upper bound for achievable system performance. 3.3 Heiter's Nonlinear Model Heiter [68] took a s l i g h t l y d i f f e r e n t approach to the problem of modeling TWTAs, concentrating on the development of a simple r e l a t i o n s h i p between the measured single-tone amplitude and phase c h a r a c t e r i s t i c , and the instantaneous voltage transfer function. He extended the conventional 54 amplitude power s e r i e s expansion of nonlinear amplitude functions to include order dependent time delays that are i n d i r e c t l y r e l a t e d to the AM/PM e f f e c t s observed i n TWTAs. The power series representation i s separated into in-phase and quadrature phase Fourier s e r i e s expansions of the instantaneous input s i n u s o i d a l s i g n a l x ( t ) , rather than the envelope of the input s i g n a l . For a bandpass s i g n a l , Heiter uses an odd-order ser i e s expansion. For the s i n u s o i d a l input tone of peak voltage r, x(t) = r cos a) Qt (3.9) the fundamental bandpass s i g n a l output y(t) i s rela t e d to the input x(t) by y(t) = Ap cos d) 0t + A^ s i n co0t (3.10) where 3 3 A = a, r + — a, r + ... (3.11) P IP 4 3P A = a. r + - a, r 3 + ... (3.12) q i q 4 3q and the a 's and a 's are determined by the time delay d i f f e r e n c e s . P q This model i s useful when the nonlinear device i s predominantly third-order, as i s the case when a TWTA i s operated several dB below saturation. Heiter used h i s model to r e l a t e the two-tone third-order intermodulation with the single-tone amplitude and phase d i s t o r t i o n measurements of a TWTA. It i s t h i s approach which, i n a modified form, was used for the l i n e a r i z e d TWTA modeling i n th i s study. 55 4. DEVELOPMENT OF NEW LINEARIZED TWTA MODEL In order to determine the required c h a r a c t e r i s t i c s of a p r e d i s t o r t i o n l i n e a r i z e r , a s u i t a b l e mathematical r e l a t i o n s h i p between the nonlinear transfer functions of the TWTA and the l i n e a r i z e r i s required. A s i m p l i f i e d narrowband model of the TWTA based on the work by Heiter [68], i s f i r s t described, followed by the development of an appropriate l i n e a r i z e r model. The f i n a l l i n e a r i z e d TWTA model i s then developed. 4.1 TWTA Model H e i t e r 1 s TWTA model was selected since i t combined r e l a t i v e a n a l y t i c s i m p l i c i t y and experimental convenience i n determining relevant amplitude and phase parameters which adequately characterized the nonlinear behaviour of the TWTA. The vector network analyzer provides a suitable measurement of the amplitude and phase functions of the TWTA, and i s used to provide the data points for the model. This model i s constructed by extending the conventional amplitude power serie s expansion which describes the AM/AM d i s t o r t i o n , to include order dependent time delays which describe the AM/PM d i s t o r t i o n e f f e c t s of the TWTA. The power serie s considers only odd-order terms, as the d i s t o r t e d signals generated by even terms are outside the passband of i n t e r e s t . Heiter's model r e s u l t s In an odd-order trigonometric Fourier s e r i e s representation of the nonlinear output s i g n a l . As shown by Heiter, with the input s i g n a l tone of peak voltage E T and angular frequency U Q expressed as V ^ t ) = E cos u 0 t , (4.1) 56 the output of the TWTA can be written as an n-term odd-order expansion of the form V 2 n ( t ) = a 1E f c cos ^ ( t - ^ ) + a 3 E t 3 cos 3 u J 0 ( t - t 3 ) + ... (4.2) where a., i s the series expansion c o e f f i c i e n t of odd-order j and i s the time delay of the j ^  order term. Heiter uses trigonometric i d e n t i t i e s to reduce the output s i g n a l to a convenient fundamental term which includes the d i s t o r t i o n e f f e c t s of higher odd-order terms. The fundamental output s i g n a l at u j Q can be written as _,, _ s i n (ji„t p u q where V 2 ( t ) = A cos ojgt + A s i n u>Q (4.3) A = a. E + 1 a, E „ 3 + L a. E t 5 + . . . (4.4) p l p t 4 3p t g 5p t A = a. E + 1 a. E „ 3 + - a, E t 5 + . . . (4.5) q l q t 4 3q t g 5q t and A and A are the resultant in-phase and quadrature-phase components P «1 expressed i n terms of the new gain and phase c o e f f i c i e n t s a ^ and a j q * Referencing the measured output s i g n a l to the l i n e a r time delay t j _ , the delay t, i s set to zero. The c o e f f i c i e n t s a. and a. are now related to the c o e f f i c i e n t s of (4.2) by a. = a. cos ( o j n t.) = a. cos d>. (4.6) JP j 0 j 3 3 = a j s i n ( u > Q t^) = a^ s i n $ y (4.7) Including fundamental and third-order terms, a l p = a L (4.8) a, = 0 since t = 0 (4.9) l q 1 a 3 p - a 3 cos 4,3 T^IO) 57 = a^ s i n <j>3. (4.11) The fundamental output of the TWTA can now be expressed as V 2 ( t ) = A cos ( u 0 t + <3?) (4.12) where 2 2 1/2 A = (A + A y'C (4.13) p q A <S> = arctan (_!). (4.14) Ap Expanding A, the r e s u l t i s A = [ ( a 1 E t + 1 a 3 E ^ c o s <J>3 ) 2 + (2 a 3 E ^ s i n <t>3 ) 2 ] 1 / 2 4 4 = Ej.aJU + 1 — E t 2 c o s (J,3 ) 2 + (2 — E t 2 s i n <|>3 ) 2 ] 1 / 2 4 a^ 4 a^ 3 a 3 „ 2 A ^ 9 r a3N2 „ 4 n l / 2 = E ^ ^ l + E t cos <|)3 + _ (_) E f c J 2 a^ 16 a^ 3 a 3 2 1/2 = E a 1 (1 + — - — E cos <(>, ) (4.15) 2 a x In equation (4 .15) , E^a^ i s the l i n e a r part of A and the bracketed term represents the gain compression R^: R = [ l + 2 — E 2cos <K ] 1 / 2 (4.16) 2 3 ] l so that A = E. a. R . (4.17) t i c Expanding the AM/PM conversion term 3 a3 2 U — E t S i n ^ a l $ = arctan ,[ - ] (4.18) (1 — E t 2 c o s <t>3) a l 58 This i s the same r e s u l t as obtained by Nojima et a l . [10]. A s i m i l a r i t y between the AM/AM and AM/PM terms can simp l i f y the r e l a t i o n s h i p further. The unique interdependence of these factors i s shown as a function of the independent third-order d i s t o r t i o n phase ^ where, R 2 - l $ = arctan [ — tan <(>,] (4.19) 2 R + 1 c This s i m p l i f i e d expression has been v e r i f i e d by Nojima et a l . [10], [17] for several devices. The r e s u l t , however, i s only v a l i d up to power l e v e l s a few dB below saturation where the third-order e f f e c t s dominate. 4.2 P r e d i s t o r t i o n L i n e a r i z e r Model To e s t a b l i s h a piece-wise l i n e a r l i m i t e r c h a r a c t e r i s t i c for the li n e a r i z e d TWTA, the pre d i s t o r t e r must compensate for both amplitude and phase d i s t o r t i o n . A simple architecture which accomplishes these functions was i n i t i a l l y developed by Bremenson [23-25]. Although the c i r c u i t used by Bremenson operates at the f i n a l RF transmit frequency, the basic concept can be applied at lower frequencies before upconversion i f the s i g n a l i s narrowband. The p r e d i s t o r t e r ( F i g . 4.1) i s a p a r a l l e l branch c i r c u i t which consists of two si g n a l paths. The main s i g n a l path provides a l i n e a r amplitude transfer function, and the secondary path provides a nonlinear amplitude transfer function. Both paths are i d e a l l y phase d i s t o r t i o n f r e e . The s i g n a l present at the input of the pr e d i s t o r t e r i s s p l i t equally, with 59 Variable Attenuator Variable Phase Shifter 90 Deg. Sp l i t t e r 0 Deg. Combiner Nonlinear Distortion Generator F i g . 4.1. Block Diagram of P r e d i s t o r t i o n L i n e a r i z e r Model 60 the main path lagging the secondary path by 90 degrees and containing gain and phase processing blocks. The secondary path consists of a nonlinear d i s t o r t i o n causing element whose c h a r a c t e r i s t i c can be modeled as an odd-order amplitude power s e r i e s . The two signals are then added. The o v e r a l l e f f e c t of the nonlinear c i r c u i t can be seen i n the vector diagram shown i n F i g . 4.2. With a large increase i n the input s i g n a l , A increases l i n e a r l y from A to A', and B increases nonlinearly from B to B'. The resultant sum of the two sig n a l vectors C, provides both an amplitude expansion and a phase r o t a t i o n . These e f f e c t s are i n opposition to the AM/AM and AM/PM conversion of the TWTA. By c a r e f u l adjustment of the s i g n a l vector combining r a t i o and angle, a s u i t a b l e inverse AM/AM and AM/PM c h a r a c t e r i s t i c i s generated by the nonlinear p r e d i s t o r t e r as predicted by the third-order l i n e a r i z e r model. An exact amplitude and phase match over a large dynamic range i s not possible with a p r a c t i c a l c i r c u i t , but reasonable reduction of third-order IMD i s expected. The a n a l y t i c a l expression for the pr e d i s t o r t e r can be developed i n terms of the third-order power serie s expansion of the instantaneous voltage s i g n a l . The d i s t o r t i o n generator i s modelled as a two term odd-order amplitude expander. The voltage at the input of the branch c i r c u i t can be represented as a sin u s o i d a l tone V ^ ' ( t ) , V,'(t) = E cos u-t (4.20) 61 F i g . 4.2. V e c t o r Diagram of P r e d i s t o r t i o n L i n e a r i z e r Model 62 where E^ i s the peak input voltage and IXI^ i s the c a r r i e r frequency. Assuming an i d e a l 3 dB/90 degree s p l i t t e r at the input and a 3 dB/0 degree combiner at the output, the following expressions are obtained: The output voltage component from the nonlinear branch at the fundamental frequency u^, i s given by b. - E 2 b„ V = _ E (1 + - — ) cos u.t (4.21) D 2 P 4 2 b L 1 where l i n e a r and third-order terms of the power series expansion are i n c l u d -ed. The output voltage component from the l i n e a r branch i s written as V = —— K E cos (u.t - - ) . (4.22) L 2 P 1 2 The variables are defined as b^ = l i n e a r voltage transfer c h a r a c t e r i s t i c c o e f f i c i e n t of d i s t o r t i o n generator c i r c u i t ; b^ = third-order voltage transfer c h a r a c t e r i s t i c c o e f f i c i e n t of d i s t o r t i o n generator c i r c u i t ; K = low voltage l e v e l combining voltage r a t i o of both branches. The resultant vector i s b R_ P V '(t) = E — (K 2 + 1 ) 1 / 2 cos (u.t + a + B) (4.23) I p E 2 J-where Rg = Gain expansion factor of p r e d i s t o r t e r b- E 2 1/2 = (1 + 1 — I ) (4.24) 4 b l (K 2 + 1) 63 a = arctan (i) - constant (4.25) K 8 = AM/PM conversion term 1 — ( - ) E 2 8 b l K 2 + 1 P = arctan [ ] . (4.26) - b~ E 2  8 b l K 2 + l E f f e c t i v e l y , the voltage gain of the pr e d i s t o r t e r i s _ JV<t)| _ b, /2 + 3 2 _^ /2 p 2 4 p 2 |Vx'<t>| b ^ K ^ + l ) = — (K 2 + l ) 1 / 2 R_ . (4.27) 2 4.3 L i n e a r i z e d TWTA System Model Simple expressions have been derived for both the transfer functions of the TWTA and the i d e a l third-order p r e d i s t o r t e r ( F i g . 4.3). A suitable equation r e l a t i n g the two transfer functions w i l l express the d i s t o r t i o n generator third-order c o e f f i c i e n t i n terms of the TWTA third-order d i s t o r t i o n phase. The voltage gain of the TWTA from equation (4.15) G T = — = a. (1 + 1 E 2 — cos < U 1 / 2 (4.28) t 2 a l The peak input of the TWTA E £ , i s rela t e d to the output of the pr e d i s t o r t e r ; by neglecting the small d i s t o r t i o n term i n Rg, E = C G E * C — (K 2 + 1 ) 1 / 2 E (4.29) t P P , P 64 r INPUT I VOLTAGE j o EpCOS W,t. n 90 WHVBRID ATTEMUATOR r 1 VT 1 i 1 1 vr 1 INSTANTANEOUS VOLTAGE TRANSFER FUNCTION J IDEAL 0° COMBINER LINEAR GAIN AND FREQUENCY CONVERSION BLOCK V , » E T C O S W n"t VOLTAGE GAJN = C TWTA LINEAR VOLTAGE G A l W » a , V a= a ,E T R c cos ( w 0 i +^ ) OUTPUT VOLTAGE F i g . 4.3. Block Diagram of L i n e a r i z e d TWTA System Model 65 where C i s the conversion block voltage gain. The TWTA gain can now be written as b 2 G_ = a. ( 1 + 2 E 2 C 2 — (K 2 + 1) -2 cos <tu) 1 / 2 . (4.30) T 1 2 p 4 & 1 3 The o v e r a l l voltage gain of the p r e d i s t o r t e r , gain block, and TWTA i s G Q = G p C G T . (4.31) The p r e d i s t o r t e r gain can also be expressed as G p - — (K 2 + 1 ) 1 / 2 (1 + Z x ) l / 2 (4.32) with 3 2 b3 Z. = — E - (4.33) 1 4 p 2 b x ( K Z + 1) which i s the d i s t o r t i o n term of Rg. The TWTA gain can be expressed i n a s i m i l a r form, G T = 3 l ( l + Z 2 ) 1 / 2 (4.34) with 9 v> 2 3 9 C 1 ? a3 Z = — E _ _ (K^ + 1) — cos <u (4.35) 1 2 P 4 a L which i s the d i s t o r t i o n term of R . c The form of equations (4.31), (4.32), and (4.34) indicates that the o v e r a l l amplitude or gain function v a r i a t i o n (AM/AM) of the system at high power l e v e l s can be minimized i f Z L = - Z 2. (4.36) This r e s u l t s i n the expression 66 b 2 G Q = - i - (K 2 + 1 ) (1 + Z x) C 2 a x 2 (1 - Z L) h 2 1 2 2 2 2 = — — (K + 1) f / a / (1 - Z / ) 2 b 2 - — (K 2 + 1) C 2 a 2 since Z 2 i s small. (4.37) 2 From equation (4.36), we obtain 3 3 3 2 1 2 a3 ± f = - — C — — (K + 1) — - cos <|)3 (4.38) 4 b x ( K 2 + 1 ) 2 4 a l and 2 3 ~ C b l 2 2 a 3 b- - _ (K + 1)^ _ cos . (4.39) 2 a l To minimize the o v e r a l l phase s h i f t (AM/PM) through the system, set P = - $. (4.40) Matching the numerators of the phase expression for both the p r e d i s t o r t e r and the TWTA gain functions, equations (4.18) and (4.26), and neglecting the d i s t o r t i o n term i n the denominators, the r e s u l t i s b b 2 a -12 [-1 ) E 2 = - 2 C 2 J _ (K 2 + 1) E 2 ll s i n <j>3 (4.41) 8 b l (K 2 + 1) P 4 4 P a l and , I^V <1±J^ >2 !i s l n . (4.42) 2 K a t 67 Equations (4.39) and (4.42) reduce to the f i n a l r e l a t i o n s between the a 3 p r e d i s t o r t e r parameters K and b.., and the TWTA parameters — and <|>-. The r e s u l t i s K = tan <t>3 (4.43) - b l 3c 2 ! L a i b. = . (4.44) 3 2 cos <|>3 For the p r e d i s t o r t i o n l i n e a r i z e r system, the value of b^ i s determined a f t e r s e l e c t i n g suitable values of the intermediate gain block C and d i s t o r t i o n generator l i n e a r gain term b^. The value of K i s determined from the experimental value of <J>3 obtained for a p a r t i c u l a r TWTA. The response of the l i n e a r i z e d TWTA was checked using program "PRE" l i s t e d i n Appendix B. 68 5. DESIGN OF IF PREDISTORTION LINEARIZER Having selected a c i r c u i t a r c h i t e c t u r e based on the Bremenson diode l i n e a r i z e r structure [23], the p r a c t i c a l problem of r e a l i z a t i o n required a t t e n t i o n . Before the actual TWTA p r e d i s t o r t i o n l i n e a r i z e r could be designed, the transfer c h a r a c t e r i s t i c s of the TWTA were required. The single-tone envelope amplitude (AM/AM) and phase (AM/PM) transfer functions of a Thomson 14 GHz 20 watt h e l i x TWTA were measured using an HP 8510 network analyzer t e s t system. A system l e v e l design of a p r a c t i c a l SCPC earth s t a t i o n transmitter using the l i n e a r i z e d TWTA was then determined. The measured c h a r a c t e r i s t i c s of the TWTA were f i t t e d to the narrowband TWTA model and design requirements for the actu a l p r e d i s t o r t e r stage were established. The l i n e a r i z e r c i r c u i t was then developed, and c i r c u i t elements were determined using a simple third-order model for the diode wave-shaping network. 5.1 Measurement of TWTA Transfer C h a r a c t e r i s t i c s A complete set of amplitude compression (AM/AM) and phase delay (AM/PM) c h a r a c t e r i s t i c curves can be obtained by e x c i t i n g the TWTA with a single-tone at various drive l e v e l s using a vector network analyzer. The measurement set up i s shown i n F i g . 5.1. This set up consisted of a HP 8510 Network Analyzer System with a HP 8340A synthesized s i g n a l generator and HP 8515A S-parameter test set, a 14 GHz HP 8349A wide band s o l i d - s t a t e pre-amplifier, two HP 436A power meters, c a l i b r a t e d d i r e c t i o n a l couplers, 23 watt RF <5.VSMJ load, and miscellaneous low l o s s c o a x i a l cables and connectors. A HP 9826 computer was used as HP 8510 Network Analyzer System HP 7475A P l o t t e r HP 8340 Signal Source HP 8515A S-Parameter Test Set Port 1 -20 dB HP 8349A Power Amplifier Port 2 20 dB D i r e c t i o n a l Coupler HP 436A Power Meter 20 dB D i r e c t i o n a l Coupler HP 9826 Computer -20 dB HP 436A Power Meter Computer Interface Bus (HP-IB) Fi g . 5.1. Block Diagram of TWTA Transfer C h a r a c t e r i s t i c s Measurement System 70 the instrument c o n t r o l l e r and data processor. The s o l i d - s t a t e power ampli f i e r was inserted before the TWTA i n order to provide s u f f i c i e n t drive l e v e l to saturate the TWTA. The HP 8515A S-parameter test set was used to provide the S21 magnitude and phase measurements. Calibrated d i r e c t i o n a l couplers were used to monitor the absolute power l e v e l s at both the input and output of the TWTA. The HP 8510 system was i n t e r n a l l y c a l i b r a t e d for f u l l two-port S-parameter measurements. The system response covered 14 - 14.5 GHz i n a 51 point frequency span. The con t r o l program for the transfer c h a r a c t e r i s t i c measurement i s l i s t e d i n Appendix B ("HPA"). Before actual measurements on the TWTA can be made, a c a l i b r a t i o n procedure must be introduced to accurately separate the TWTA response from the associated test equipment used i n the test set up. This was accomplished by using the normalization function as described i n the HP 8510 operating manual [69]. This procedure e f f e c t i v e l y removed the transfer response of the a d d i t i o n a l devices. Although the r e s o l u t i o n of the network analyzer system i s better than 0.01 dB and 0.01 degrees, the c a l i b r a t e d accuracy of the r e l a t i v e magnitude and phase measurements i s approximately ± 0.2 dB and ± 0 . 5 degrees r e s p e c t i v e l y . The power meters are also accurate to within ± 0.2 dB. At a source frequency of 14.05 GHz, the 1 dB compression point occurred at 3 dB output backoff, and the phase delay at t h i s point was approximately 13 degrees. A plo t of the envelope amplitude and phase 71 response i s shown i n F i g . 5.2. The r e s u l t s i n d i c a t e that the TWTA i s r e l a -t i v e l y l i n e a r up to 8 dB below saturation, but exhibits f a i r l y s i g n i f i c a n t amplitude compression and phase delay past t h i s point. At drive l e v e l s beyond saturation, the TWTA output power begins to decrease as expected. The value of the third-order d i s t o r t i o n phase <t>3 can be determined using equation (4.16). Since the expected operating point of the l i n e a r i z e d TWTA i s near the 1 dB compression point, t h i s value was selected for the c a l c u l a t i o n . The 1 dB compression point i s also the highest l e v e l at which the d i s t o r t i o n phase model i s usable. The equivalent input peak voltage corresponding to the 1 dB power gain compression point of the TWTA at 14.05 GHz was measured at 0.23 v o l t s . The value of obtained at t h i s frequency was -116.5 degrees and the value of a^/a^ was 5.8, which are s i m i l a r to the res u l t s obtained by Nojima [10]. The expected amplitude and phase c h a r a c t e r i s t i c s of the TWTA at 14.05 GHz calculated using the third-order d i s t o r t i o n phase model at the 1 dB compression point are compared with the experimentally determined response i n Table 5.1. A comparison of the r e s u l t s i n d i c a t e a good f i t between the experimental data and the modeled response up to the 1 dB compression point. As expected, the TWTA model simulated response deviates from the measured response above the 1 dB compression point, as the simple third-order model does not account for higher order e f f e c t s which are s i g n i f i c a n t above the 1 dB compression point. 5.2 Development of L i n e a r i z e r System Structure The actual c i r c u i t structure selected for developing an IF p r e d i s t o r t i o n l i n e a r i z e r i s based on the scheme used by Bremenson et a l . 50 50 -15 -10. -5 0 5 10 INPUT PONER (DBM) F i g . 5 . 2 . M e a s u r e d 14 GHz TWTA T r a n s f e r C h a r a c t e r i s t i c s TABLE 5.1 Measured and Modeled TWTA Amplitude and Phase Transfer C h a r a c t e r i s t i c s at 14.05 GHz TWTA Input Power (dBm) Amplitude Response (Gain Compression) (dB) Measured Modeled Phase Response (Phase Delay) (Degrees) Measured Modeled - 14 - 12 - 10 - 8 - 6 - 4 - 2 0 0 0.1 0.1 0.2 0.5 0.8 1.4 2.2 0.05 0.1 0.15 0.25 0.45 0.75 1.2 2.1 0 0.5 1.7 3.4 5.8 9.9 15.4 21.8 0.9 1.4 2.3 3.6 5.9 9.6 15.9 25.9 Measured 1 dB power gain compression point = - 3 dBm at TWTA input 74 Although the structure has been used only i n d i r e c t RF l i n e a r i z a t i o n , the pre d i s t o r t e r can also be used at a suitable IF Frequency i n an SCPC a p p l i c a t i o n . The present Spacetel SCPC system used by M i c r o t e l has IF frequencies of 70 MHz and 280 MHz. The 280 MHz IF provides a good compromise between c i r c u i t r e a l i z a b l l i t y and percentage bandwidth constra i n t s . The actual operating bandwidth requirement at 280 MHz spans the range from 262 - 298 MHz, or 13% of the centre frequency. If frequency response v a r i a t i o n s are minimized between the IF p r e d i s t o r t i o n stage and TWTA, l i t t l e i f any degradation i n performance i s expected. S u f f i c i e n t bandwidth between the pre d i s t o r t e r and TWTA was also required i n order to pass the f u l l third-order predistorted s i g n a l spectrum to the TWTA. Drive l e v e l s into the 280 MHz/14 GHz upconverter stage must be kept r e l a t i v e l y low, ensuring that the upconversion process does not cause a d d i t i o n a l s i g n a l d i s t o r t i o n , which might e f f e c t the l i n e a r i z e r performance. A block diagram of a possible system structure for a l i n e a r i z e d earth s t a t i o n i s shown i n F i g . 5.3. System l e v e l s were determined from p r a c t i c a l constraints imposed by the l i n e a r i t y s p e c i f i c a t i o n s of a v a i l a b l e a u x i l i a r y components such as the 70/280 MHz and 280 MHz/14 GHz upconverter stages, and the inherent i n s e r t i o n loss of the p r e d i s t o r t e r stage [70]. The maximum allowable drive l e v e l at the input to the f i r s t converter without generating s i g n i f i c a n t IMD was s p e c i f i e d at -20 dBm. The maximum drive l e v e l at the input of the second upconverter was -35 dBm and -21 dBm for the f i r s t and second upconverters r e s p e c t i v e l y . The TWTA input l e v e l required for saturation was 5 dBm. These power l e v e l s necessitated the addition of I F UPC - I n t e r m e d i a t e F r e q u e n c y U p c o n v e r t e r RF UPC - R a d i o F r e q u e n c y U p c o n v e r t e r 7 0 / 2 8 0 M H x L I N E A R I Z E R I F U P C R F U P C 14 GHTL TRANSMIT I N T E R F A C E F I L T E R T W F i g . 5 . 3 . P r o p o s e d L i n e a r i z e d E a r t h S t a t i o n TWTA C o n f i g u r a t i o n 76 s u i t a b l e gain block stages before the l i n e a r i z e r at 280 MHz, and a f t e r the l i n e a r i z e r at 14 GHz. This basic system structure was used to integrate the IF l i n e a r i z e r and the TWTA i n a p r a c t i c a l earth s t a t i o n configuration. 5.3 Hardware R e a l i z a t i o n of IF P r e d i s t o r t e r The actual 280 MHz l i n e a r i z e r stage consisted of a p a r a l l e l branch s i g n a l processing c i r c u i t ( F i g . 5.4). Several v a r i a b l e controls were added to provide tuning c a p a b i l i t y . A d e s c r i p t i o n of the c i r c u i t operation and methodology for designing the diode d i s t o r t i o n generator follows. 5.3.1 L i n e a r i z e r C i r c u i t Operation The input c i r c u i t consisted of a pre-amplifier for providing the necessary drive l e v e l for the nonlinear d i s t o r t i o n generator i n the secondary branch. The amplified s i g n a l was s p l i t by a 90 degree 3 dB s p l i t t e r into the primary and secondary branches. The primary path consisted of a v a r i a b l e attenuator and phase s h i f t e r . The secondary (nonlinear) path consisted of a d i s t o r t i o n generator and phase s h i f t e r . The d i s t o r t i o n generator was made up of a p a r a l l e l connected network of components, inc l u d i n g a pair of a n t i p a r a l l e l Schottky b a r r i e r diodes, a va r i a b l e r e s i s t o r R , and a n e u t r a l i z i n g inductor L . The diode network was P 5 P followed by a f i x e d attenuator. The signals from the two p a r a l l e l branches were v e c t o r i a l l y summed i n a 3 dB in-phase combiner. The var i a b l e attenua-tor i n the primary path provided the necessary control for f i n e adjustment of the combining power r a t i o K. The var i a b l e phase s h i f t e r s provided independent co n t r o l of the i n s e r t i o n phase of both branches, and allowed INPUT G = i q d B DI ,D2 * T O S H I B A VARIABLE ATTEkl. 30 VARIABLE SH IFTER VAR IABLE ATTEKJ. ( O - I S d B ) OVERALL UKIEAR GAIN* 0 d&. F i g . 5.4. 280 MHz P r e d i s t o r t i o n L i n e a r i z e r C i r c u i t 78 adjustment of the combining angle of the two s i g n a l vectors over the operating bandwidth. The basic operation of the p r e d i s t o r t e r c i r c u i t can be described as follows. For low input power l e v e l s , the c i r c u i t i s l i n e a r , imposing no amplitude and phase co r r e c t i o n . At low l e v e l s the diodes are seen as a r e l a t i v e l y high impedance, and the current flows i n the p a r a l l e l r e s i s t o r Rp. This corresponds to the l i n e a r power l e v e l s at the TWTA well below saturation. At higher input power l e v e l s , the nonlinear action of the diodes provides a mechanism for generating a decrease i n attenuation with an increase i n drive l e v e l . As the input s i g n a l l e v e l i s increased, the diode pair begins to conduct, e f f e c t i v e l y decreasing the dynamic impedance of the diodes. A t y p i c a l voltage transfer c h a r a c t e r i s t i c of the a n t i p a r a l l e l diodes i s shown i n F i g . 5.5. The s i g n a l voltage developed across the terminating r e s i s t o r R^ e s s e n t i a l l y generates a nonlinear voltage expansion transf e r function, which i s shaped by the I-V current voltage character-i s t i c s of the diode p a i r . The n e u t r a l i z i n g inductor L p i s used to resonate the p a r a s i t i c capacitance of the diode p a i r , minimizing the frequency dependent reactance e f f e c t s . By c a r e f u l adjustment of the v a r i a b l e attenuator and phase s h i f t e r s , a suitable inverse c h a r a c t e r i s t i c of the network i s obtained as predicted by the l i n e a r i z e r model. A perfect amplitude and phase match over a large dynamic range i s not possible with t h i s nonideal c i r c u i t , but a reasonable reduction of third-order IMD i s expected. A photograph of the experimental 280 MHz l i n e a r i z e r i s shown i n F i g . 5.6. The d i s t o r t i o n generator i s integrated with commercially a v a i l a b l e couplers, combiners, attenuators, and phase s h i f t e r s . F i g . 5 . 5 . A n t i p a r a l l e l D i o d e N o n l i n e a r V o l t a g e T r a n s f e r C h a r a c t e r i s t i c s F i g . 5.6. 280 MHz P r e d i s t o r t i o n L i n e a r i z e r 81 5.3.2 D i s t o r t i o n Generator Design The diode wave-shaping c i r c u i t elements i n the nonlinear branch were designed using a simple third-order power serie s representation of the a n t i p a r a l l e l diode network. The i n i t i a l design step required DC ch a r a c t e r i z a t i o n of the diodes. The s t a t i c diode measurements were then used to f i t the diode response to the well known diode equation. I = I [exp (Hi) - 1] (5.1) J s nkT where v = voltage across diode junction i j = current through diode junction I = reverse saturation current s and , . N = q/nkT q = e l e c t r o n i c charge = 1.6 x 10~ 1 9 C n = i d e a l i t y f a c t o r k = Boltzmann's Constant = 1.38 x 10" 2 3 J/°K T = Absolute temperature °K The o v e r a l l instantaneous voltage transfer function of the a n t i p a r a l l e l diode network with p a r a l l e l and terminating r e s i s t o r s can be written as v — v R R v ^ = {2 l s sinh [ N ( v R i - v )] + ( 1 r )} R 2 ( 5 ' 2 ) where v D = input voltage across R. R l 1 82 v = output voltage across R. . This odd function can also be modeled as a two term odd order power series expansion of the instantaneous input RF voltage under the following assumptions: 1. The diode junction capacitance i s constant over the dynamic RF voltages expected during operation. 2. The capacitive reactance i s e f f e c t i v e l y n eutralized by the p a r a l l e l inductance over the operating frequency range. 3. Other energy storage e f f e c t s i n the diode are n e g l i g i b l e at the operating frequency of 280 MHz. 4. The bulk series resistance i s much less than the dynamic resistance of the diode. As stated by Weiner and Spina [71], these are reasonable assumptions for obtaining a usable dynamic diode model from s t a t i c DC measurements. The two term odd-order power serie s expansion can be expressed as For the Schottky b a r r i e r diodes which were used i n the l i n e a r i z e r , the measured DC I-V c h a r a c t e r i s t i c s are accurately modeled when the following constants are used i n the diode equation. Toshiba 1SS239 Diode I = 1.0 x 10 _ 8A. N = 1.17 R = 4 ohms ( n e g l i g i b l e for third-order nonlinear operation). s The third-order power serie s c o e f f i c i e n t s b, and b- were then 3 (5.3) X 83 obtained by using a least-squares polynomial curve f i t t i n g program to best f i t the response of equation (5.2) over the expected input power range determined from the o v e r a l l system design. Several i t e r a t i o n s of a numerical analysis program were used to obtain the desired value of the p a r a l l e l r e s i s t o r R^, which provided the best match for the calculated value of the f i r s t and third-order expansion c o e f f i c i e n t s b^, bj obtained using equation (4.44). The i n s e r t i o n loss of the diode network and.the intermediate stage gain were estimated for the f i r s t i t e r a t i o n . S l i g h t s h i f t s of these values were required to provide a good o v e r a l l match. A difference of 10% between the desired and c a l c u l a t e d values of the power se r i e s c o e f f i c i e n t s was considered s u f f i c i e n t for the experimental l i n e a r i z e r . The f i n a l c a l c u l a t i o n value of the p a r a l l e l r e s i s t o r was 420 ohms. The HP 9826 programs used for the diode p r e d i s t o r t e r design are l i s t e d i n Appendix B ("ROOT" and "POLY"). For tuning c a p a b i l i t y , R^ was made adjustable i n the actual l i n e a r i z e r c i r c u i t . The nominal values of interstage gain and p r e d i s t o r t e r gain were 21 dB and -31 dB r e s p e c t i v e l y . 84 6. EVALUATION OP LINEARIZED TWTA IMD PERFORMANCE In order to v e r i f y the l i n e a r i z e r design methodology and assess the l i n e a r i z e r performance, a su i t a b l e test system for measurement of IMD reduction was required. For evaluation of the l i n e a r i z e d TWTA under normal operating m u l t i c a r r i e r s i g n a l conditions, an automated test system for two-tone, m u l t i c a r r i e r , and noise loading signals was designed. The l i n e a r i z e r was evaluated over a large dynamic power range, various s i g n a l frequencies, and temperature. These r e s u l t s were then compared with the performance of other published l i n e a r i z e r r e s u l t s . 6.1 Development of M u l t i c a r r i e r Intermodulation Test System The automated l i n e a r i z e d TWTA evaluation system consisted of a mock-up SCPC earth s t a t i o n transmitter using modules from the M i c r o t e l SPACETEL terminal. A d d i t i o n a l RF attenuators, couplers, and pre-amplifiers were integrated into the system to a s s i s t i n the adjustment of the l i n e a r i z e r . A test rack of eight SPACETEL modulators and a white noise generator were used to provide various s i g n a l loadings for d r i v i n g the l i n e a r i z e d TWTA. For two-tone s i g n a l , the measurement system was configured to evaluate both t h i r d and f i f t h - o r d e r IMD. For four or more s i g n a l c a r r i e r loading, only third-order IMD measurements were taken. For noise-loading conditions, the NPR was determined. The s i g n a l power l e v e l s were i n i t i a l l y set to conform with the power l e v e l s used i n the numerical analysis of the l i n e a r i z e r c i r c u i t . Power meters were located at convenient monitor points to ensure c a r e f u l alignment of the l i n e a r i z e r . An HP 8566B Spectrum analyzer was used for d i r e c t 85 measurement of IMD l e v e l s . A complete block and l e v e l diagram of the test system i s shown i n F i g . 6.1. Careful attention was paid to ensure that a l l subsystems were operating i n a l i n e a r mode. The r e s i d u a l third-order IMD was measured at below 40 dBc over the e n t i r e dynamic power range. A l l data c o l l e c t i o n was obtained using a s p e c i a l l y programmed instrument c o n t r o l system. An HP 9826 Computer was used to automatically c o n t r o l the input power l e v e l for the system and process the actual power and IMD l e v e l s measured by the spectrum analyzer. The program used i s l i s t e d i n Appendix B (INTERMOD" and PLTNPRAUTO). An averaging routine was used i n the measurement system to obtain improved accuracy. The nonlinearized TWTA reference curves were obtained by disconnecting the d i s t o r t i o n generator c i r c u i t i n the l i n e a r i z e r . The l i n e a r i z e r performance evaluation was determined under the following system configurations. I. Two-tone measurements a) E f f e c t of IMD reduction with l i n e a r i z e r optimized at d i f f e r e n t output backoff operating points. b) E f f e c t of IMD reduction over a 36 MHz bandwidth with l i n e a r i z e r optimized at 5 dB output backoff. c) E f f e c t of IMD reduction over an operating temperature v a r i a t i o n of 50 degrees with l i n e a r i z e r optimized at 5 dB output backoff. I I . M u l t i c a r r i e r and NPR signal measurements a) E f f e c t of IMD reduction with 4-8 c a r r i e r loading, with l i n e a r i z e r optimized at 5 dB output backoff. b) E f f e c t of NPR reduction with l i n e a r i z e r optimized at 5 dB output backoff. (-46 dBm) MULT I TONE &EKI. Id6 S T E P •TTEUUATOR / ATTEN- 2 o d B m / t o n e ( o - 3 o d B ) L= 2d& IF UPC 20 dB O U T P U T E>/O P O W E R L E V E L S P E R . T O N E A R E S H O W N F O R R E F E R E N C E TRW TRW c . CA2800 CA2640 ,0dB lOdB PAD CT~. MONITOR MERRlMAC ARM-I G«i7dB G=2Zd6 DIRECTIOKJAL COUPLER MCL PDC IO-I L=ld& ^ ISPLIT-TER l /ARlABLE ATTeN.(0-l9<»6) RRIMAC £'LP5M ME  2 8 Q MHz. L I M E A R I - Z E R RG 188 C O A X c 3«aa£m M C L ZFDC 10-1 1 j ^ v W - ' L.idB wajC/MAC POM-io-250 L«4d& 10 dB -19 840SA VECTOK. VOLTMETER BdftPAD | Raiaa ^ iod6 -40 (d«»J[ o" COM-MERRlMAC A«M-> -33 CdBro) 4-F I X E D P A D L»ldB VARIA&LE A T T E M . - 4 6 L»2odB T F O 14 GHz TRANSMIT 'NTERFACE MONITOR POINT *2 VARIABLE APOLLO Wlt>L> MERRlMAC (-S4dBmJ r t u i A lOdB J Co-2D L-o.sdB • L=ldB f-bo C» 20 d» RF UPC 2£0MHr/l4 GHx L»5dB DIRCCTIONAL C O U P L E R M 6 R P I M A C CjM-K3- I2G ARRA ',5 dB M O U I T O R POIMT »3 MERRlMAC C3A1-20-I2S \dBrO APOLLO ISOLATOR. THOMSON MERRlMAC TWTA CiM-20-l2<S HP436A <S-d.4d& HP 436A POWER METER*I / W I T H 2odBPAD) M P SPEC-(HIGH P0W€R 2.5 WATT HEAD} F i g . 6 . 1 . B l o c k D i a g r a m o f L i n e a r i z e d TWTA E v a l u a t i o n S y s t e m 00 87 For the m u l t i c a r r i e r IMD<evaluation, the unmodulated c a r r i e r s were spaced at a set frequency pattern i n order to measure the third-order IMD with a spectrum analyzer. The development of the frequency plan and IMD measurement s l o t i s described i n Appendix A [72]. The input c a r r i e r s were adjusted at a uniform power l e v e l . The i n d i v i d u a l c a r r i e r powers were a l l measured to be within 1 dB of each other, i n d i c a t i n g an acceptable frequency response i n the test system. The use of equal power c a r r i e r s ensured that the intermodulation power measured i n the IMD frequency s l o t was representative of a l l third-order A+B-C IMD products generated across the passband. For noise loading measurements, a S c i e n t i f i c Atlanta 4647 IF Noise Generator and IF bandpass f i l t e r were used to provide a wide band noise loaded s i g n a l 20 MHz wide centered at a 75 MHz input frequency. A band r e j e c t f i l t e r at 80 MHz was used to s e l e c t i v e l y notch out a s p e c i f i c narrow s l o t of noise. This s l o t was observed on the spectrum analyzer and a noise power r a t i o (NPR) was obtained. The NPR i s known to vary from a maximum at the centre of the noise band to a value approximately 2 dB lower at the band edge when the noise s i g n a l i s passed through a nonlinear device such as a TWTA [1]. The NPR at the centre of the band i s approximately 0.8 dB worse than the NPR measured 5 MHz from centre. The actual output backoff of the TWTA, r e l a t i v e to single-tone saturation, was determined by subtracting the IMD noise power from the output power meter reading. The l i n e a r i z e r was optimized at various operating points by minimizing the third-order IMD observed on the spectrum analyzer by f i n e adjustment of the v a r i a b l e attenuators and phase s h i f t e r s . A l l test 88 instruments used i n the l i n e a r i z e r evaluation system were c a r e f u l l y c a l i b r a t e d to ensure adequate system measurement accuracy. The errors i n absolute power measurements were determined by the accuracy of the HP 436A power meters, and the errors i n the r e l a t i v e IMD measurements were determined by the accuracy of the HP 8566B spectrum analyzer. The power backoff measurements have a possible error of ± 0.2 dB, and the IMD l e v e l s are within ± 1 dB. These error l i m i t s are s p e c i f i e d by the instrument manufacturer [73]. 6.2 Results of Intermodulation Mesurements The TWTA intermodulation performance, both with and without the l i n e a r i z e r , was evaluated under various operating conditions. The output backoff for a l l measurements i s referenced to the single-tone saturation output power l e v e l of the TWTA without the l i n e a r i z e r . Only minor adjustments to the va r i a b l e attenuators and phase s h i f t e r s were required to tune the l i n e a r i z e r for optimum performance at various output backoffs. 6.2.1 Two-Tone IMD Results The f i r s t set of performance curves were obtained using the industry standard two-tone measurement technique. Both third-order and f i f t h - o r d e r IMD l e v e l s were recorded over a large dynamic range. The two-tone performance curves are shown i n F i g . 6.2-6.13. 6.2.1.1 Optimization at Various Output Backoff Levels A series of IMD measurements was done with the third-order IMD l e v e l optimized at output backoffs ranging from 4 to 7 dB. Reduction of the third-order IMD products i s evident i n a l l cases, but the e f f e c t of varying the optimization point i s quite v i s i b l e . The shape of the IMD reduction 89 4 E OUTPUT B'O (DB) 4 6 OUTPUT B'O (DB) Fig. 6.2. TWTA Two-Tone IMD with Linearizer Optimized at 4 dB Output Backoff E8 30 IM (DBC) 20 A 7 \ > // '/ 7 \ 1 1 / . / , 1 if / y i' if / ^ 4 E OUTPUT B'O (DB) Fig. 6.3. TWTA Two-Tone IMD with Linearizer Optimized at 5 dB Output Backoff • in a » . i Itl CL1N.1 40 30 IM (DBC) 20 h / i \ i / 7 y A. ' / / \ \ 1 I i / / / i I if / / i ' (/ '/ / / / 4 E OUTPUT B'O (DB) Fig. 6.4. TWTA Two-Tone IMD with Linearizer Optimized at 6 dB Output Backoff Fig. 6.5. TWTA Two-Tone IMD with Linearizer Optimized at 7 dB Output Backoff 90 curve changes from a rather peaked IMD reduction at a 4 dB output backoff to a les s dramatic IMD reduction over a larger dynamic range at a 7 dB backoff. The peaked response of F i g . 6.2 provides the highest reduction of third-order IMD at powers close to saturation, but i t s response curve at 8 dB backoff a c t u a l l y crosses the IMD curve of the TWTA without the l i n e a r i z e r . The broad response of F i g . 6.3, optimized at 5 dB backoff, provides the best o v e r a l l response. The s e l e c t i o n of the 5 dB backoff point for a l l subsequent measurements i s i n agreement with the optimal tuning procedure observed by Bremenson et a l . [25]. 6.2.1.2 Frequency Response C h a r a c t e r i s t i c s In order to evaluate the l i n e a r i z e r performance over a 36 MHz bandwidth, four a d d i t i o n a l measurement sets were taken with the two input tones s h i f t e d i n frequency. The c a r r i e r frequencies used for these a d d i t i o n a l four measurements are l i s t e d i n Table 6.1. The l i n e a r i z e r was adjusted to provide optimum third-order IMD suppression at 5 dB backoff, and was set with the two c a r r i e r s i n the middle of the passband. Without further adjustment, the l i n e a r i z e r IMD performance curves were obtained over the frequency range. The r e s u l t s of the four measurement sets i n d i c a t e that the l i n e a r i z e r e x h i b i t s reasonable broadband performance. Although there i s some degradation of IMD reduction at 8 dB backoff at the upper band edge ( F i g . 6.9), the improvement i n output backoff at 25 dBc third-order IMD l e v e l i s better than 3 dB for a l l cases. 6.2.1.3 Temperature Response C h a r a c t e r i s t i c s With the l i n e a r i z e r optimized at 5 dB output backoff, and with the tones set to the o r i g i n a l frequencies i n the middle of the passband, the 91 TABLE 6.1 Two-Tone Test Frequency Assignment F i g . Input (MHz) 2nd IF (MHz) RF (GHz) 6.2-6.5 F l = 69.545 279.545 14.0035 F2 = 70.454 280.454 14.0045 6.6 F l = 52.745 262.745 13.9867 F2 = 53.654 263.654 13.9877 6.7 F l = 61.145 271.145 14.9951 F2 = 62.054 272.054 14.9961 6.8 F l = 77.945 287.945 14.0119 F2 = 78.854 288.854 14.0129 6.9 F l = 86.345 296.345 14.0203 F2 = 87.254 297.254 14.0213 92 0 2 « S e IB 0 2 4 E 8 IB OUTPUT B'O (DB) OUTPUT B/O (DB) Fig. 6.6. TWTA Two-Tone IMD with IF Carriers Fig. 6.7. TWTA Two-Tone IMD with IF Carriers at 262.745 and 263.654 MHz at 271.145 and 272.054 MHz wo e*oat in _ m m m IN M D OROOt in (L.XN.1 STH ORDER i n ( L I N . ) 0 2 4 G B 10 0 2 4 6 8 10 OUTPUT BXO (DB) OUTPUT B/0 (DB) Fig. 6.8. TWTA Two-Tone IMD with IF Carriers at 287.945 and 288.854 MHz Fig. 6.9. TWTA Two-Tone IMD with IF Carriers at 296.345 and 297.254 MHz 93 l i n e a r i z e r was subjected to a temperature cy c l e . This cycle consisted of a temperature increase from 0 to 50 degree Celsius over a period of four hours, maintenance at 50 degrees over a 16 hour period, and a temperature decrease of 12.5 degrees per hour, returning the unit to the s t a r t i n g temperature of 0 degrees. A complete IMD measurement set was obtained one hour a f t e r each temperature change. A representative set of IMD curves showing the performance of the l i n e a r i z e r of a temperature v a r i a t i o n from 50 to 0 degrees i s shown i n F i g . 6.10-6.13. Although there appears to be some change i n the shape of the IMD reduction curves as the temperature i s varied, the performance shows reasonable s t a b i l i t y . The backoff improvement was better than 3 dB with a 25 dBc third-order IMD l e v e l over the f u l l temperature range. 6.2.2 M u l t i c a r r i e r and Noise-Loading IMD Results Third-order IMD vs output backoff performance curves were obtained for 4, 6, and 8 c a r r i e r loading, and NPR vs output backoff curves were obtained f o r a noise-loaded s i g n a l . The m u l t i c a r r i e r IMD power shown on the curves i n F i g . 6.14 through 6.16 represents the t o t a l third-order IMD power (both A+B-C and 2A-B types) present i n the centre s l o t of an equivalent equal power and equal frequency spaced c a r r i e r grouping. The IF predi s t o r -t i o n l i n e a r i z e r was used to obtain curves which show the reduction of i n t e r -modulation d i s t o r t i o n possible r e l a t i v e to the IMD performance of the TWTA without the l i n e a r i z e r under m u l t i c a r r i e r s i g n a l conditions. The l i n e a r i z e r was optimized to provide best third-order IMD suppression using two c a r r i e r s at the 5 dB output backoff operating l e v e l . The l i n e a r i z e r was not readjusted for the duration of the m u l t i c a r r i e r t e s t i n g . A set of curves 94 e 2 4 6 B IB 0 2 4 6 8 10 OUTPUT B'O (DB) OUTPUT BxO (DB) Fig. 6.10. TWTA Two-Tone IMD with Linearizer Fig- 6.11. TWTA Two-Tone IMD with Linearizer Temperature at 0 Deg. C Temperature at 12 Deg. C MO UKUUf XM rra ORDER m M D ORDER 11 (LIM. ) HTM ORDER XM (LIM. I 4 6 OUTPUT B^ O (DB) 4 6 OUTPUT B/O (DB) Fig. 6.12. TWTA Two-Tone IMD with Linearizer Temperature at 38 Deg. C Fig. 6.13. TWTA Two-Tone IMD with Linearizer Temperature at SO Deg. C 95 30 IM (DB) 20 30 IM (DB) 20 -1 1 2 4 6 OUTPUT B/O (DB) B 18 2 4 E OUTPUT B'O (DB) B IB Fig. 6.14. TWTA Centre Slot Third-Order IMD with 4 Carriers Fig. 6.15. TWTA Centre Slot Third-Order IMD with 6 Carriers WD MDEH IH MD 0R0O1 I H ( L I M . ) 30 IM (DB) 28 10 EB SB 3B IM (DB) 2B 10 / 2 4 6 OUTPUT B'O (DB) B IB 2 4 6 OUTPUT B^ O (DB) B 10 Fig. 6.16. TWTA Centre Slot Third-Order IMD with 8 Carriers Fig. 6.17. TWTA Centre Slot NPR 96 i n d i c a t i n g l i n e a r i z e r IMD suppression i s presented i s shown i n F i g . 6.14-6.17. As expected [1], the measured r e s u l t s i n d i c a t e that the intermodulation power increases with an increase i n the number of c a r r i e r s for a given t o t a l c a r r i e r output power. The l i n e a r i z e r , however, does provide a r e l a t i v e suppression of IMD for a l l m u l t i c a r r i e r signals at power lev e l s s l i g h t l y below saturation. At close to the saturation l e v e l , the l i n e a r i z e r does not appear to reduce the IMD. This f a c t was also observed by Satoh [32] during the development of a p r e d i s t o r t i o n l i n e a r i z e r with an optional l i m i t e r . At saturation, the l i n e a r i z e r provides a stronger instantaneous s i g n a l , whose peak s i g n a l power tends to severely overdrive the TWTA. The larger peak-to-rms power r a t i o of the m u l t i c a r r i e r s i g n a l cannot be compensated by the l i n e a r i z e r at saturated power l e v e l s . As shown by Satoh, the addition of a l i m i t e r between the l i n e a r i z e r and the TWTA can su b s t a n t i a l l y improve the l i n e a r i z e r performance at saturation. In small SCPC earth sta t i o n s , where the number of c a r r i e r s i s low, a simple l i n e a r i z e r without l i m i t e r would be s u f f i c i e n t when operating at a t y p i c a l IMD l e v e l of 25 dBc. 6.3 Overview of L i n e a r i z e r Performance The experimental IF p r e d i s t o r t i o n l i n e a r i z e r has s i g n i f i c a n t l y re-duced the third-order IMD of the 20 watt Ku band TWTA. The re s u l t s obtained are comparable with those of Bremenson's et a l . [25] RF p r e d i s t o r t i o n l i n e a r i z e r (shown i n F i g . 6.20). The IF l i n e a r i z e r r e s u l t s also compare well with those of other p r e d i s t o r t i o n l i n e a r i z e r s [30], [38]. The separation of the IF diode l i n e a r i z e r by a stage of upconversion and gain blocks does not degrade the IF l i n e a r i z e r performance. The IF pr e d i s t o r t i o n OUTPUT POWER 1 BELOW SAT. 0 .2 .4 .e .8 97 (DB) 1.2 1.4 1.6 1.8 2 2.2 2.4 : V \ — - — _ X t ^ _ _ — , WITH LINERRIZER WITHOUT LINEARIZER 8 10 12 NUMBER OF CARRIERS F i g . 6.18. TWTA S a t u r a t e d Output Power vs Number o f C a r r i e r s 0 l 2 12 I | | I I I | | 1 | 1 1 0 2 4 6 8 10 12 : NUMBER OF CARRIERS F i g . 6.19. TWTA Output B a c k o f f a t 25 dBc IMD vs Number o f C a r r i e r s 98 l i n e a r i z e r has shown third-order IMD reduction c a p a b i l i t y which approaches the t h e o r e t i c a l l i m i t s established by Saleh [44]. As shown i n F i g . 6.19, at an IMD l e v e l of 25 dBc, the l i n e a r i z e r has demonstrated that the output backoff performance of a t y p i c a l TWTA can be improved by at le a s t 3 dB over a s i g n i f i c a n t bandwidth and temperature range, and with various c a r r i e r s e t s . This could enhance t y p i c a l SCPC performance by almost doubling the c a r r i e r capacity of a given TWTA. The f i f t h - o r d e r IMD was s l i g h t l y degraded, but was maintained at l e v e l s below the third-order l i m i t of 25 dBc. At saturated power l e v e l s , the l i n e a r i z e d TWTA performance was i n f e r i o r to the TWTA without l i n e a r i z e r both i n terms of IMD and t o t a l saturated output power ( F i g . 6.19). As shown by Satoh [32], t h i s could be improved with the addition of a l i m i t e r . F i n a l l y , the l i n e a r i z e r designed required only minor tuning from o r i g i n a l design l e v e l s In order to provide su b s t a n t i a l IMD reduction. The o r i g i n a l p r e d i s t o r t i o n modeling approach appears sound, since the o v e r a l l design procedure has established a working l i n e a r i z e r prototype which can be manufactured for commercial use i n SCPC earth s t a t i o n s . Lower cost IF c i r c u i t components used i n the l i n e a r i z e r provide considerable cost saving p o t e n t i a l . E l e c t r o n i c a l l y tunable components, such as the va r i a b l e attenuator and phase s h i f t e r s could be configured to an adaptive c o n t r o l scheme. A l i m i t e r could be included i n future designs to further improve the TWTA IMD performance i n large m u l t i c a r r i e r earth s t a t i o n s . 99 Two tone intermodulation products Two tone intermodulation products F-| = 3720 MHz ?2 = 3730 MHz F, = 3970 MHz F 2 = 398O MHz Two tone intermodulation products F, = 1*170 MHz F 2 = M 8 0 MHz F i g . 6.20. Two-Tone IMD Performance Curves of the Bremenson TWTA Linearizer [25] 100 7. CONCLUSIONS A comprehensive review of l i n e a r i z a t i o n techniques has indicated that s i g n i f i c a n t improvements i n communication system performance are possible by using various l i n e a r i z a t i o n schemes. These techniques reduce the AM/AM and AM/PM d i s t o r t i o n of nonlinear HPAs. The method of l i n e a r i z a t i o n selected depends on both system a p p l i c a t i o n and the type of s i g n a l t r a f f i c amplified by nonlinear HPAs. The p r e d i s t o r t i o n technique i s found to be the simplest and most economic technique i n both t e r r e s t r i a l and s a t e l l i t e communication system a p p l i c a t i o n s . For SCPC s a t e l l i t e earth s t a t i o n TWTAs, an IF p r e d i s t o r t i o n scheme using a simple p a r a l l e l branch c i r c u i t a r c h i t e c t u r e was investigated. A nonlinear model of the TWTA and l i n e a r i z e r was developed to a s s i s t In the f i n a l hardware design of the pr e d i s t o r t e r c i r c u i t . The r e s u l t s of amplitude and phase measurements of a t y p i c a l TWTA were used to provide s u i t a b l e design parameters for the l i n e a r i z e r c i r c u i t . The c i r c u i t was integrated into a small SCPC Ku band earth s t a t i o n and i t s performance was evaluated under various c a r r i e r loading conditions. The performance of the l i n e a r i z e d TWTA was characterized over a wide frequency range, temperature, and si g n a l type. The experimental l i n e a r i z e r s u b s t a n t i a l l y reduced the third-order IMD generated by the TWTA. An o v e r a l l improvement of better than 3 dB i n output backoff was achieved at a third-order IMD l e v e l of 25 dBc. The l i n e a r i z e r provided an adjustment c a p a b i l i t y for optimizing performance at various output l e v e l s . The IF p r e d i s t o r t i o n l i n e a r i z e r performance compared well with other l i n e a r i z e r s . 101 A new l i n e a r i z e r design procedure has been v e r i f i e d f o r a commercial 14 GHz TWTA SCPC a p p l i c a t i o n . The improvements i n the e f f i c i e n c y of the TWTA c l e a r l y show the benefits of a low cost IF p r e d i s t o r t i o n l i n e a r i z e r for use i n small SCPC earth s t a t i o n a p p l i c a t i o n s . Further areas of study should include long term s t a b i l i t y performance of the TWTA l i n e a r i z e r . An adaptive l i n e a r i z e r could provide increased system performance and should be investigated. The measurement of performance gains using a l i m i t e r stage between the l i n e a r i z e r and the TWTA are suggested. The study of l i n e a r i z a t i o n of SSPAs with a s i m i l a r l i n e a r i z e r Is recommended. The measurement of spe c t r a l regrowth and BER c h a r a c t e r i s t i c s with s i n g l e - c a r r i e r d i g i t a l signals amplified by both TWTAs and SSPAs would also be u s e f u l . 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Abuelma'atti, "Frequency-Dependent Nonlinear Quadrature Model f o r TWT Amplifi e r s , " IEEE Trans, on Communications, Vol. COM-32, Aug. 1984, pp. 982-986. [68] G.L. Heiter, "Characterization of N o n l i n e a r i t i e s i n Microwave Devices and Systems," IEEE Trans, on Microwave Theory and Techniques, Vol. MTT-21, Dec. 1973, pp. 797-805. [69] Hewlett-Packard, "HP 8510 Network Analyzer System Operating and Programming Manual," Jan. 1985. [70] M. Wlodyka, "HPA L i n e a r i z e r Study: Hardware Development of a TWTA P r e d i s t o r t i o n L i n e a r i z e r , " M i c r o t e l P a c i f i c Research Technical Report TR 62-84-001, July 1984. [71] D.D. Weiner and J.F. Spina, "Sinusoidal Analysis and Modeling of Weakly Nonlinear C i r c u i t s with A p p l i c a t i o n to Nonlinear Interference E f f e c t s , " Van Nostrand Reinhold Co., 1980, Sec. 7.2. [72] M. Wlodyka, " M u l t i c a r r i e r Intermodulation Performance of a TWTA P r e d i s t o r t i o n L i n e a r i z e r , " M i c r o t e l P a c i f i c Research Technical Report TR 883-86-001, Jan. 1986. [73] Hewlett-Packard, "HP 8566B Spectrum Analyzer Operating and Programming Manual," July 1985. [74] W.C. Babcock, "Intermodulation Interference i n Radio Systems," B e l l System Tech. Journal, Jan. 1953, pp. 63-73. [75] R.J.F. Fang and W.A. Sandrin, "Carrier Frequency Assignment f o r _ _ Nonlinear Repeaters," COMSAT Technical Rev. Vol. 7, No. 1, Spring 1977, pp. 227-245. 108 APPENDIX A Frequency Plan for M u l t i c a r r i e r IMD Measurements The third-order IMD i s known to dominate a l l higher odd-order IMD i n TWTA systems [1] and provides a good i n d i c a t o r for assessing the nonlinear behaviour of both l i n e a r i z e d and nonlinearized TWTAs. For two c a r r i e r s , the IMD product 2A-B i s dominant, generating two IMD products equally spaced around the c a r r i e r p a i r . For three or more c a r r i e r s , the IMD product A+B-C dominates i n power content over the 2A-B and other f i f t h and higher-order IMD products. The r e s u l t s of two-tone IMD t e s t i n g i n d i c a t e that the f i f t h - o r d e r 3A-2B IMD product was at le a s t 30 dB below the c a r r i e r l e v e l for for both a l i n e a r i z e d and nonlinearized TWTA operating below 3 dB output backoff. From 3 dB output backoff up to saturation, the f i f t h - o r d e r IMD was lower that the third-order IMD. Since the f i f t h and higher-order IMD contributions are secondary, and due to the d i f f i c u l t y i n separating these products, these contributions are not included i n the 4, 6, and 8 c a r r i e r study. In order to carry out quan t i t a t i v e third-order IMD measurements with a m u l t i c a r r i e r s i g n a l spectrum, several constraints are imposed on the measurement techniques which can be used. Among the most serious r e s t r i c t i o n s i s the d i f f i c u l t y In separating various IMD products which can be generated by d i f f e r e n t mixing combinations, such as 2A-B, A+B+C, 2A+B-C-D, e s p e c i a l l y when several IMD products f a l l on the same frequency. Therefore, a s p e c i a l frequency plan was developed which enabled the measurement of third-order 2A-B and A+B-C IMD products. 109 J The i n d i v i d u a l c a r r i e r assignment was based on an optimum frequency plan [74,75], which ensured that no third-order IMD product f a l l s on any m u l t i c a r r i e r s i g n a l c o n s i s t i n g of up to 9 c a r r i e r s . This was necessary to accurately measure each i n d i v i d u a l c a r r i e r power at the output of the TWTA. The c a r r i e r frequency assignments were modified s l i g h t l y from the optimum plan to ensure that an IMD frequency s l o t contained only a sing l e A+B-C IMD product. This was required since the unmodulated c a r r i e r s were a l l phase co r r e l a t e d . A sum of correlated IMD products f a l l i n g i n the same s l o t would not add on a pure power basis and would cause inconsistent power measurements. A sing l e product IMD s l o t i s s u f f i c i e n t , however, as the peak-to-rms r a t i o of the m u l t i c a r r i e r s i g n a l and i t s e f f e c t on IMD and output power would translate back onto the si n g l e A+B-C product measured. The RF test bed consisted of 9 SPACETEL IF modulators which have the c a p a b i l i t y to generate unmodulated c a r r i e r s i n 600 channel s l o t s between 52.02-87.96 MHz. The 60 kHz re s o l u t i o n i n c a r r i e r spacing provided the necessary c a r r i e r separation for the modified frequency plan. This plan separated the c a r r i e r s i n multiples of 180 kHz, with the exception of one c a r r i e r and the IMD s l o t which were s h i f t e d by 60 kHz. The m u l t i c a r r i e r frequency plan i s shown i n Table A . l . 110 TABLE A . l M u l t i c a r r i e r Test Frequency Assignment Optimum Frequency Plan (for 9 c a r r i e r s ) C a r r i e r # Channel # 1 1 2 2 3 6 4 13 5 26 6 28 7 36 8 9 42 45 Modified Frequency Plan C a r r i e r # Channel t 1 2 3 4 5 6 7 8 9 36 42 44.667 1 2 6 13 26 28 IF Frequency (MHz) 77.1 78.18 78.66 70.8 70.98 71.7 72.96 75.3 75.66 * 9th c a r r i e r not used Intermodulation Measurement S l o t 2 c a r r i e r s (GHz) 3-8 c a r r i e r s (GHz) RF Frequency (GHz) 14.0111 14.01218 14.01266 14.0048 14.00498 14.0057 14.00696 14.0093 14.00966 Noise Loading (GHz) 14.01002 14.01062 14.014 I l l APPENDIX B HP 9826 Programs and Output L i s t i n g s DESCRIPTION PAGE PROGRAM HPA:' HP 8510 NETWORK ANALYZER AUTOMATED AMPLITUDE AND PHASE MEASUREMENT OF POWER AMPLIFIERS 2. PROGRAM PRE: 3. PROGRAM ROOT: 4. PROGRAM POLY: a) LINEARIZED TWTA MODEL SIMULATION b) TYPICAL OUTPUT LISTING a) PROGRAM FOR FINDING THE ROOT OF THE NONLINEAR ANTIPARALLEL DIODE EQUATION b) TYPICAL OUTPUT LISTING a) PROGRAM FOR CURVE FITTING WITH ODD ORDER POLYNOMIAL OF ORDER 3 b) TYPICAL OUTPUT LISTING 5. PROGRAM INTERMOD: TWTA LINEARIZER INTERMOD MEASUREMENT PROGRAM 6. PROGRAM PLTNPRAUTO: PROGRAM TO MEASURE NPR OF LINEARIZER RACK USING THE SPECTRUM ANALYZER AT 14 GHz 112 10 20 30 40 50 60 PROGRAM "HPA" 8510 NETWORK ANALYZER AUTOMATED AMPLITUDE AND PHASE MEASURMENT OF POWER AMPLIFIERS BY MARK WLODYKA 70 ! 30 COM Spl,Sp2.Fq!,Fq2.Fqc.Mg<50),Pow1<50>.Pow2<50>,P1c,P2c 90 DIM Am P(50),Phs(50),Pou1a(50>,Pow2a(50),Pow1p<50).Pou2p<50) 100 REAL Freq,Rl,Im,Mag,Phase 110 ASSIGN @Nua TO 716 120 ASSIGN @Nuia_systbus TO 717 130 ASSIGN @Nua_data1 TO 716;FORMAT ON 140 ASSIGN $Pm1 TO 712 150 ASSIGN @Pm2 TO 713 160 ASSIGN @Mm1 TO 714 170 ASSIGN @M(i)2 TO 715 180 BEEP 190 OUTPUT ©Pml:"9D-V" 200 OUTPUT @Pm2;"9D-V" 210 OUTPUT @Mm1;"F1R0S0T0Y1" 220 OUTPUT ©Mc^'TIROSOTOYI" 230 L=1 240 PRINTER IS 1 250 BEEP 260 PRINT "INPUT DATE" 270 INPUT Rem$ 280 PRINT "INPUT TITLE" 290 INPUT Rem1$ 300 PRINT "INPUT STARTING SOURCE POWER AND POWER STEP SIZE AND CENTRE FREQ" 310 INPUT S p l . S t z . F c 320 PRINT "INPUT PM1 AND PM2 COUPLING FACTORS" 330 INPUT P1c,P2c 340 PRINT "CALIBRATE?" 350 INPUT Y£ 360 IF Y$="N" THEN GOTO S t a r t 370 GOSUB C a l 380 PRINT "USER DATA REQUIRED?" 390 INPUT YS 400 IF Y$="N" THEN GOTO S t a r t 410 PRINT "MEASURE USER PAR?" 420 INPUT YS 430 IF Y$="N" THEN GOTO Norm 440 ! MEASUREMENT OF USER 1 (A1) AND USER 2(B2) DATA 450 User: ! 460 OUTPUT @Nua:"CHAN1" 470 OUTPUT @Nua;"MENUDISP" 480 OUTPUT @Nwa;"OVER" 490 OUTPUT @Nua:"MENUPARA" 500 OUTPUT @Nua:"USER1" 510 OUTPUT @Nwa:"CHAN2" 520 OUTPUT «Nua:"USER2" 530 OUTPUT @Nua:"MENUSTIM" 540 OUTPUT QNua;"POWE";SP1 550 OUTPUT @Nwa:"REST" 560 PRINTER IS 701 570 PRINT Rem$ 580 PRINT Rem1$ 590 PRINT "MEASUREMENT OF USER 1 AND USER 2" 600 PRINT "SOURCE (DBM) PM1 (DBM) PM2 USER (DB)" 113 610 PRINTER IS 1 620 GOSUB P l o t r 630 PRINTER IS 1 640 PRINT "USER DATA COMPLETE" 650 PRINT "REPEAT?" 660 INPUT Y$ 670 IF Y$="Y" THEN GOTO User 680 S t a r t : PRINT "CONNECT HPA TO BE TESTED" 690 OUTPUT @Nua;"CHAN1" 700 OUTPUT @Nwa;"S21" 710 PRINT "READY?" 720 INPUT Y$ 730 PRINT "LOAD CAL SETS?" 740 INPUT YS 750 IF Y$="N" THEN GOTO Meas 760 OUTPUT @Nwa;"MENUCAL" 770 OUTPUT @Nu.a;"CORROFF" 780 OUTPUT @Nu>a;"MENUTAPE" 790 OUTPUT @Nua:"LOAD" 800 OUTPUT @Nwa;"FILE1" 810 Meas: OUTPUT @Nua:"MENUCAL" 820 OUTPUT @Nua;"CORRON" 830 LOCAL 716 840 PRINT "SELECT CAL SET ON ANALYZER" 850 PRINT "READY?" 860 INPUT Y$ 870 ENTER @Mm1;Volt1:V1S 880 ENTER €>Mm2;Volt2;V2$ 890 OUTPUT (s>Nu.a:"CHAN1" 900 PRINTER IS 701 910 PRINT Rem$ 920 PRINT Rem1$ 930 PRINT "MEASURMENT OF AMPLITUDE RESPONSE TEST#";L 940 PRINT "MM1= ";Vol11;VI $;" MM2 = ";Volt2:V2S;" CENTRE FREQ = ";Fc GHZ" 950 PRINT "PM1 COUPLING= ":P1c:"DB PM2 COUPLING- ";P2c;"DB" 960 PRINT "SOURCE (DBM) PM1 (DBM) PM2 GAIN (DB) S21" 970 PRINT "POWER INPUT OUTPUT MAG" 980 PRINT " 990 PRINTER IS 1 1000 GOSUB P l o t r 1010 PRINTER IS 1 1020 PRINT "AMPLITUDE RESPONSE COMPLETE" 1030 FOR 0=1 TO 50 1040 Amp(Q)=Mg(Q) 1050 Pow1a<Q)=Pou1(Q) 1060 Pou)2a(Q)=Pou2<Q> 1070 NEXT 0 1080 OUTPUT @Nua;"MENUSTIM" 1090 OUTPUT <s>Nua;"POWE";Sp1 1100 L=L+1 1110 PRINT "REPEAT?" 1120 INPUT Y$ 1130 IF Y$="Y" THEN GOTO S t a r t 1140 ! 1150 L=1 1160 Stptv: PRINTER IS 701 1170 PRINT RemS 1 180 PRINT RemlS -1190 PRINT "MEASURMENT OF PHASE RESPONSE TEST#":L 1200 PRINT "MM1= ";VoIt1:V1S:" MM2= ":Voit2:V2S:" CENTRE FREQ 114 ;Fc;"GHZ" 1210 PRINT "PM1 COUPLING- ":P1c:"DB PM2 C0UPLING= ";P2c;"DB 1220 PRINT "SOURCE (DBM) PM1 <DBM) PM2 GAIN S21" 1230 PRINT "POWER INPUT OUTPUT PHASE<DEG>" 1240 PRINT " " 1250 PRINTER IS 1 1260 OUTPUT €>Nua;"CHAN2" 1270 OUTPUT @No«a:"PHAS" 1280 GOSUB P l o t r 1290 PRINTER IS 1 1300 PRINT "PHASE RESPONSE COMPLETE" 1310 FOR 0=1 TO 50 1320 Phs<Q)=Mg<Q) 1330 Pou1p<Q)=Pou1<Q> 1340 Poui2p<0>=Pou2(Q) 1350 NEXT 0 1360 OUTPUT @Nua:"MENUSTIM" 1370 OUTPUT «Nua;"POWE":SP1 1380 PRINT "REPEAT?" 1390 L=L+1 1400 INPUT Y$ 1410 IF Y$="Y" THEN GOTO Stph 1420 PRINT "HPA TEST COMPLETE" 1430 PRINT "GRAPH TRANSFER FUNCTION?" 1440 INPUT Y$ 1450 IF Y$="N" THEN GOTO F i n 1460 GINIT 1470 DUMP DEVICE IS 701.EXPANDED 1480 GOSUB Graph 1490 PRINTER IS 1 1500 PRINT "REPEAT GRAPH?" 1510 INPUT Y$ 1520 IF Y$="Y" THEN GOTO 1480 1530 PRINT "IF GRAPH ON PLOTTER DESIRED. CHANGE HPIB CONNECTION" 1540 INPUT Y$ 1550 PLOTTER IS 705,"HPGL" 1560 GOSUB Graph 1570 PENUP 1580 PRINTER IS 1 1590 PRINT "CHANGE HPIB CABLE ON PLOTTER " 1600 F i n : STOP 1610 ! PLOTTING SUBROUTINE PLOTR 1620 P l o t r : ! 1630 L P=1 1640 PRINT "ENTER MAX SOURCE POWER IN DBM" 1650 INPUT Sp2 1660 Spl=S P1 1670 PRINT "CHECK PLOTTER FOR PAPER" 1680 LOCAL 716 1690 PRINT "SET SCALE AND REF LEVEL" 1700 PRINT "READY?" 1710 INPUT YS 1720 OUTPUT @Nua:"MENUSTIM" 1730 OUTPUT «Nua;"POWE":SPl 1740 OUTPUT @Nwa;"REST" 1750 OUTPUT <s>Nua;"MENUCOPY" 1760 OUTPUT @Nua;"PLOTALL" 1770 WAIT 1 1780 OUTPUT C«)Nwa:"ADDRPASS 31" 1790 GOSUB Power 1800 Mg(Lp)=Mag 1810 1820 1830 1840 1850 1860 1870 1880 1890 1900 1910 1920 1930 1940 1950 1960 1970 1980 1990 2000 2010 2020 2030 2040 2050 2060 2070 2030 2090 2100 2110 2120 2130 2140 2150 2160 2170 2180 2190 2200 2210 2220 2230 2240 2250 2260 2270 2280 2290 2300 2310 2320 2330 2340 2350 2360 2370 2380 2390 2400 2410 Rep: Pou1<L P)=P1 P O U J2<L P)=P2 Spl=Spl+Stz L P=Lp+1 OUTPUT @Nua;"MENUSTIM" @Nua;"POWE":SPl @Nwa;"REST" 115 OUTPUT OUTPUT WAIT 1 OUTPUT OUTPUT OUTPUT €>Nua GOSUB Power Mg<Lp)=Mag Poul(Lp)=P1 Pou>2<LP)=P2 IF S P2>Spl THEN PRINTER IS 701 PRINT CHR$<12) RETURN t @Nua;"MENUCOPYM @Nwa:"PLOTTRAC" ADDRPASS 31 GOTO Rep ! POWER METER MEASUREMENT SUBROUTINE POWER Power: OUTPUT @Nua;"MENUSTIM" @Nwa:"SINP" «Pn>1 ;"9D-V" @Pm2:"9D-V" Graph OUTPUT OUTPUT OUTPUT WAIT 2 OUTPUT <s>Nuia;"OUTPMARK" ENTER @Nua_data1:Mag,Phase ENTER @Pm1:P1 ENTER GPm2:P2 P1=Pt+P1c P2=P2+P2c LOCAL 713 LOCAL 729 OUTPUT @Nuja:"RAMP" PRINTER IS 701 PRINT USING "6D.2D":Spl:PI PRINTER IS 1 RETURN i PRINT "ENTER X AND Y SCALE INPUT X o f f . Y o f f FOR J=1 TO 50 NEXT J WINDOW -5.30.-5.30 DRAW X AND Y AXIS ;P2:P2-P1 :Mag OFFSETS IN DB" PEN 1 MOVE -3.0 DRAW 20.0 MOVE 0,0 DRAW 0.20 LINE TYPE 1 DRAW DIVISIONS ON GRAPH LINE TYPE 1 FOR 1=0 TO 20 STEP 2 MOVE 1,0 DRAW 1.20 2420 NEXT I 2430 LINE TYPE 1 2440 FOR 1=0 TO 20 STEP 2 2450 MOVE 0,1 2460 DRAW 20 ,1 2470 NEXT I 2480 LINE TYPE 1 2490 MOVE -2,0 2500 CSIZE 3..6 2510 ! 2520 ! PRINT LABELS 2530 ! 2540 MOVE -5.-5 2550 LABEL Rem$ 2560 FOR 1=0 TO 20 STEP 4 2570 MOVE -2,1 2580 LABEL(I)+Yof f 2590 NEXT I 2600 FOR 1=0 TO 20 STEP 4 2610 MOVE 1-1,-2 2620 LABEL<I)+Xof f 2630 NEXT I 2640 ! 2650 ! DRAW AMP TRANSFER CURVE 2660 ! 2670 PEN 2 2680 X=Pou1a(1>-Xoff 2690 Y=Pou2a(1>-Yoff 2700 MOVE X,Y 2710 FOR J=2 TO Lp 2720 IF Pou2a(J)=P2c THEN GOTO St g r 2730 X=Pou1a(J)-Xoff 2740 Y=Pou2a(J)-Yoff 2750 PRINT X,Y 2760 DRAW X,Y 2770 NEXT J 2780 ! 2790 PRINTER IS 1 2800 PRINT "PLOT PHASE CHARACTERISTICS?" 2810 INPUT Y$ 2820 IF Y$="N" THEN GOTO St g r 2830 PRINT "INPUT PHASE SCALE OFFSET IN DEGREES" 2840 INPUT Phoff 2850 PEN 1 2860 !PRINT PHASE LABELS 2870 ! 2880 FOR 1=0 TO 20 STEP 4 2890 MOVE 20 .1 2900 LABEL<I*2)+Phoff 2910 NEXT I 2920 . • ! 2930 ! DRAW PHASE CURVE 2940 ! 2950 PEN 1 2960 MOVE 22,10 2970 LABEL "PHASE" 2980 MOVE 22.9 2990 LABEL "(DEG)" 3000 PEN 3 3010 X=Pou1p<1)-Xoff 3020 Y=(Phs(1>-Phoff)/2 3030 LINE TYPE 7 1 1 7 3040 MOVE X,Y 3050 FOR J=2 TO L P 3060 IF Pou2p(J)=P2c THEN GOTO St g r 3070 X = Pou»1p(J)-Xoff 3080 Y = ( P h s ( J ) - P h o f f ) / 2 3090 PRINT X,Y 3100 DRAW X.Y 3110 NEXT J 3120 MOVE 21,7 3130 DRAW 23.7 3140 LINE TYPE 1 3150 ! PRINT TITLE AND COMMENTS" 3160 ! 3170 S t g r : MOVE 0,22 3180 PEN 1 3190 PRINTER IS 701 3200 LABEL Rem1$ 3210 MOVE 0,21 3220 LABEL "MEASURED TRANSFER CHARACTERISTICS AT";Fc;"GHZ" 3230 MOVE 5,-3 3240 LABEL "INPUT POWER (DBM)" 3250 MOVE -5.10 3260 LABEL "OUTPUT" 3270 MOVE -5,9 3280 LABEL "POWER" 3290 MOVE -5.8 3300 LABEL "(DBM)" 3310 PEN 2 3320 MOVE -5,7 3330 DRAW -3,7 3340 RETURN 3350 C a l : PRINT "CALIBRATE FREQUENCY RESPONSE AND STORE ON TAPE 3360 PRINT "READY?" 3370 INPUT YS 3380 RETURN 3390 END 118 10 20 30 40 50 GO 70 80 90 100 110 120 130 140 150 160 170 180 190 200 210 220 230 240 250 260 270 280 290 300 310 320 330 340 350 360 370 380 390 400 410 420 430 440 450 460 470 480 490 500 510 520 530 540 550 560 570 580 590 600 610 i PROGRAM PRF LINEARIZED TWTA MODEL "SIMULATION PROGRAM WRITTEN BY M. WLODYKA LAST REV JUL 85 DEG PRINTER IS 701 PRINT "LINEARIZED TWTA SIMULATION STUDY" PRINT PRINT "THIRD ORDER DISTORTION PHASE MODEL" PRINTER IS 1 Inp: PRINT "ENTER P3 IN DEGREES-INPUT P3 PRINT "ENTER R3N" INPUT R3n Out: PRINTER IS 701 PRINT "P3=",P3."DEG"."R3N=".R3n PRINTER IS 1 PRINT "INPUT LOW.HIGH.STEP SIZE IN DBM" INPUT K1.J.O Qs=INT((J-K1)/Q)+1 PRINTER IS 701 PRINT PRINT "CALCULATION OF PREDISTORTER CHARACTERISTICS" PRINTER IS 1 PRINT "ENTER VALUE OF K (USE TAN(P3>>" INPUT K PRINTER IS 701 PRINT "K=".K PRINTER IS 1 PRINT "ENTER VALUE OF PREDISTORTER GAIN (DB)" INPUT Pg PRINTER IS 701 PRINT "PREDISTORTER GAIN=",Pg."DB" PRINTER IS 1 PRINT "ENTER NOMINAL TWT GAIN (DB)" INPUT TQ PRINTER IS 701 PRINT "TWT GAIN=",Tg,"DB" Pgv=10~(Pg/20> B1=2*Pgv/SQR(K~2+1) B1 P=20*LGT(B1) PRINT "B1=".B1,B1p."DB" PRINTER IS 1 PRINT "ENTER INTERMEDIATE STAGE GAIN (DB)" INPUT C PRINTER IS 701 PRINT "C=",C."DB" Cv=1(T<C/20> B3 = - .5*Cv'*2*B1 *3*< 1 + K"2> '"2*R3n*C0S<P3> PRINT "B3=",B3 PRINT PRINT "LINEARIZED TWTA GAIN/PHASE RESPONSE" PRINT PRINT "I/P = INPUT RMS POWER" PRINT "O/P = OUTPUT RMS POWER" PRINT "G.E. = PREDISTORTER GAIN EXPANSION" PRINT "P.A. = PREDISTORTER PHASE ADVANCE" PRINT "G.C. = TWTA GAIN COMPRESSION" 620 PRINT "P.D. = TWTA PHASE DELAY" 630 PRINT "NGC = LINEARIZED TWTA GAIN COMPRESSION" 640 PRINT "NPS = LINEARIZED TWTA PHASE SHIFT" 650 PRINT 660 PRINT " PREDISTORTER TWTA ADE" 670 PRINT. NPS" 680 PRINT (DEG)" 690 PRINT ' 119 I/P G.E. P.A. I/P O/P G.C. (DBM) (DB) (DEG) (DBM) (DB) P.D. (DEG) CASC NGC (DB) 700 710 720 730 740 750 760 770 780 790 800 810 820 830 840 850 860 870 880 f i 890 900 910 920 930 B" 940 950 960 970 980 990 1000 1010 1020 1030 1040 1050 1060 1070 1080 1090 1100 1110 1120 FOR 1=1 TO Qs V=K1+(I-1)*0 V1=SQR(50*(10~((V-30)/10)))*SQR(2) Gev=(3*V1~2*B3/(4*B1*(K'2+1))) Ge=10*LGT(Gev+1) Numer=Gev*K*.5 Beta=ATN(Numer/((Gev*.5>+1 )) PRINTER IS 1 PRINT "PRE. I/P=".DR0UND(V.4),"DBM","GAIN EXP. PRINT "BETA=",DR0UND(Beta.4)."DEG" PRINT A1=C+v"+Pg+Ge A=SQR(50*(10~((A1-30)/10)))*SQR(2) R=SQR(1+(1.5*A*A*R3n*C0S(P3))) D=1+(.75*A*A»R3n*C0S(P3)) Y1=((.75*A*A*R3n*SIN(P3))/D) R1=20*LGT(R) Y=ATN(Y1> PRINT "TWT I/P=",DROUND(A1.4)."DBM ,DROUND(Ge,4)."DB II t f GAIN COM.=".DROUND(R1,4),"DB PRINT "FY = ",DR0UND(Y.4VDEG" PRINT Pout=V+pg+C+Tg+Rl Gn=Pout-V PRINT "TWT 0/P=".DROUND(Pout.4),"DBM"."BLK GAIN=".DR0UND(Gn,4)."D NET PHASE SHIFT=";N Ps;"DEG" Ngc=DROUND(R1.4)+DROUND(Ge.4) Nps=DROUND(Be ta.4)+DROUND(Y,4) PRINT "NET GAIN COMP=":Ngc;"DB PRINT PRINT PRINTER IS 701 V=DR0UND(V.3) Ge=DROUND(Ge.3) Beta=DR0UND(Beta.3) A1=DROUND(A1,3) Pout = DROUND(Pout:.3) R1=DROUND(R1.3) Y=DR0UND(Y.3) Ngc=DROUND(Ngc,3> Nps=DR0UND(N Ps.3) PRINT USING "4D.2D";V;Ge:Beta:A1;Pout;R1;Y;Ngc;N Ps NEXT I STOP END LINEARIZED TWTA SIMULATION STUDY 120 THIRD ORDER DISTORTION PHASE MODEL P3= -116.5 DEG R3N= 5.8 CALCULATION OF PREDISTORTER CHARACTERISTICS K= 2 PREDISTORTER GAIN= -31 DB TWT GAIN= 43.4 DB B1= .025208 -31.969 DB C= 21 DB B3= .065238 LINEARIZED TWTA GAIN/PHASE RESPONSE I/P O/P G.E. P.A. G.C. P.D. NGC NPS = INPUT RMS POWER = OUTPUT RMS POWER = PREDISTORTER GAIN EXPANSION = PREDISTORTER PHASE ADVANCE = TWTA GAIN COMPRESSION = TWTA PHASE DELAY • LINEARIZED TWTA GAIN COMPRESSION * LINEARIZED TWTA PHASE SHIFT PREDISTORTER TWTA CASCADE I/P G.E. P.A. I/P O/P G.C. P.D. NGC NPS (DBM) (DB) (DEG) (DBM) (DB) (DEG) (DB) (DEG) -12.00 .01 .14 -22.00 21 .40 - .01 -.14 -0.00 -0.00 -11.00 .01 .18 -21.00 22.40 -.01 -.18 -0.00 -0.00 -10.00 .02 .22 -20.00 23.40 - .02 -.22 -0.00 -0.00 -9.00 .02 .28 -19.00 24.40 -.02 -.28 -0.00 -0.00 -8.00 .03 .35 -18.00 25.40 -.03 - .36 -0.00 -.01 -7.00 .03 .44 -17.00 26.40 -.03 -.45 -0.00 -.01 -6.00 .04 .56 -16.00 27 .40 -.04 - .57 -0.00 -.01 -5.00 .05 .70 -14.90 28.30 -.05 -.72 -0.00 -.02 -4.00 .07 .88 -13.90 29.30 - .07 - .91 -0.00 - .03 -3.00 .08 1.10 -12.90 30.30 - .09 -1.15 -0.00 - .05 -2.00 . 1 1 1 .39 -11.90 31 .30 -.11 -1 .46 -.01 -.07 -1 .00 .13 1 .74 -10.90 32.30 -.14 -1 .86 -.01 - .12 0.00 .17 2.18 -9.83 33.20 -.18 -2.36 -.01 -.18 1 .00 .21 2.73 -8.79 34.20 -.23 -3.02 -.02 -.29 2.00 .26 3.42 -7.74 35.10 - .29 -3.87 -.03 - .46 3.00 .32 4.26 -6.63 36.00 -.38 -4.99 -.05 -.73 4.00 .40 5.31 -5.60 36.90 -.49 -6.47 -.09 -1 .16 5.00 .50 6.60 -4.50 37.80 -.64 -8.44 -.14 -1 .85 6.00 .62 8.16 -3.38 38.50 - .85 -11.10 -.23 -2.95 7.00 .77 10.10 -2.23 39.30 -1 .15 -14.80 -.38 -4.72 8.00 .95 12.30 -1.05 39.80 -1.58 -19.80 -.63 -7.53 121 10 ! 20 J PROGRAM ROOT 30 ! WRITTEN BY M. WLODYKA 40 ! 50 ! LAST REV JUL 85 60 ! 70 ! PROGRAM FOR FINDING THE ROOT OF THE 80 ! NONLINEAR ANTIPARALLEL DIODE EQUATION 90 f 100 DIM Va(IOO).Vb(IOO),Pa(100).Pb(IOO) 110 S-2.5E-8 120 N=31 130 Rl=50 140 PRINTER IS 701 150 PRINT "NEWTON-RAPHSON METHOD FOR FINDING" 160 PRINT "A ROOT OF A SIMPLE DIODE EQUATION" 170 PRINTER IS 1 180 PRINT "ENTER VALUE OF RP" 190 INPUT Rp 200 PRINT "ENTER VALUE OF LINEAR COEF B1, THIRD ORDER B3" 210 INPUT B1,B3 220 PRINT "ENTER VALUE OF PAD AFTER RESISTOR/DIODE NETWORK IN DB" 230 INPUT Pad 240 PRINT "PAD VALUE IS ";Pad:"DB" 250 Bb= 10'-(-Pad/20) 260 B1=B1/Bb 270 B3=B3/Bb 280 PRINT "NEW B1=";B1:"NEW B3=";B3 290 PRINT "DO YOU WANT TO CHANGE PARAMTERS" 300 PRINT "FROM IS=":S;"N=":N:"RL=";R1:"OHMS?(Y/N)" 310 INPUT P$ 320 IF P$="N" THEN Omit 330 PRINT "ENTER VALUE OF PARAMETERS" 340 PRINT "(IS,N,RL)" 350 INPUT S.N.Rl 360 Omit: PRINT 370 PRINTER IS 701 380 PRINT "PAD VALUE=";Pad;"DB" 330 PRINT "NEW B1=":B1,"NEW B3=":B3 400 PRINT "PARAMETER VALUES ARE:" 410 PRINT "IS= ".S."AMPS" 420 PRINT "N= ".N 430 PRINT "RP= ".Rp,"OHMS" 440 PRINT "RL= ".Rl."OHMS" 450 PRINT "WHERE IS= REVERSE SATURATION CURRENT" 460 PRINT " N= Q/KTn" 470 PRINT " RP= RESISTANCE IN PARALLEL TO" 480 PRINT " DIODES" 490 PRINT " RL= LOAD RESISTANCE" 500 Rt=(B1*R P)/(1-B1-(B1*Rp/Rl)) 510 PRINT "RT=",DROUND(Rt,4)."OHMS" 520 PRINT "WHICH IS THE CORRECT DIODE TERMINATION" 530 PRINT "FOR CORRECT B1 VALUE".B1 540 R f = ( R t * R l ) / ( R t + R l ) 550 PRINT "R EFFECTIVE^',DROUND(Rf,4)."OHMS" 560 C=0 570 PRINTER IS 1 580 PRINT "ENTER RMS POWER OR PK VOLTAGE (P OR V ) " 590 INPUT FS 600 IF F$="V" THEN V o l t 610 PRINT "ENTER INPUT POWER RANGE" 122 620 PRINT "LOWER, UPPER. (DBM)" 630 INPUT P l . P h 640 PRINT 650 PRINT "ENTER STEP SIZE (DB)" 660 INPUT Pd 670 PI-PI-3 680 Ph-Ph-3 690 Ns=((Ph-Pl)/Pd)+1 700 Ns=INT(Ns) 710 FOR 1=1 TO Ns 720 Pa(I)=Pl+Pd*(I-1) 730 Va(I)=SQR(10~((Pa(I)-30)/10)*Rl)*SGR(2) 740 V=Va(I) 750 CALL Root(V,G,S.N,R P.Rf) 760 Gp=10*LGT((C-2/Rl))+30-3 770 Vb(I)=G 780 Pb(I)=G P 790 Pin=Pa(I) 800 Pout=Pb(I) 810 Gain=Pout-Pin 820 Vg=Vb(I)/Va(I) 830 PRINTER IS 701 340 PRINT "P IN=".Pin,"P OUT=" , DROLIND(Poat, 5) , "DBM GAIN=" . DROUND(Gai n.5)."DB" 850 Vin=Va(I) 860 Vout=Vb(I> 870 PRINT "V IN = " .DR0UND(Vin,4)."V OUT = ".DROUND(Vout.4),"VOLTS GAIN= ".DROUND( Vg,5) 380 NEXT I 890 STOP 900 V o l t : PRINTER IS 1 910 PRINT "ENTER I/P VOLT LIMITS" 920 PRINT "LOWER, UPPER. (VOLTS)" 930 INPUT V I , V h 940 PRINT " ENTER VOLTAGE STEP SIZE" 950 INPUT Vd 960 Ns=(Vh-Vl)/Vd+1 970 V1=V1/S0R(2) 980 V h = V h/S0R(2) 990 Ns=INT(Ns) 1000 FOR 1=1 TO Ns 1010 Va(I)=Vl+Vd*(I-1 ) 1020 V=Va(I) 1030 CALL Root(V.G.S,N.Rp.Rf> 1040 Gv=G 1050 Vb(I)=Gv 1060 Vgain=Vb(I)/Va(I) 1070 PRINTER IS 701 1080 PRINT "V IN = ".Va(I),"V OUT = ".DROUND<Vb<I).5)."VGAIN = ".DROUND(Vgai n.5) 1090 NEXT I 1100 STOP 1110 END 1120 DEF FNSinh(X) 1130 Hyps=.5*(EXP(X)-EXP(-X)) 1140 RETURN Hyps 1150 FNEND 1160 DEF FNCosh(X) 1170 Hypc=.5*(EXP(X)+EXP(-X)) 1180 RETURN Hype 1190 FNEND 1200 SUB Root<v\G.S,N,Rp,Rf > 1210 V2=V 1220 V1=V 1230 K=1 1240 Start:E1=FNSinh(N-(V1-V2)) 1250 E2=FNCosh<N*<V1-v,2>> 12G0 F=2*S*(E1)+<V1-v2)/Rp-V2/Rf 1270 D=-2*N*S*E2-1/Rp-1/Rf 1280 G=V2-F/D 1290 IF ABS<G-V2)<1.E-8 THEN E x i t l 1300 K=K+1 1310 IF K>1000 THEN E x i t 2 1320 V2=G 1330 GOTO S t a r t 1340 Exit2:ST0P 1350 E x i t l : P R I N T 1360 SUBEND NEWTON-RAPHSON METHOD FOR FINDING A ROOT OF A SIMPLE DIODE EQUATION PAD VALUE= 7 DB NEW B1= .0564157726919 NEW B3= .146049689638 PARAMETER VALUES ARE: IS= 1.E-8 AMPS N= 33.25 RP= 420 OHMS RL= 50 OHMS WHERE IS= REVERSE SATURATION CURRENT N= Q/KTn RP= RESISTANCE IN PARALLEL TO DIODES RL= LOAD RESISTANCE RT= 50.45 OHMS WHICH IS THE CORRECT DIODE TERMINATION FOR CORRECT B1 VALUE .0564157726919 R EFFECTIVE= 25. 1 1 OHMS P IN = -18 P 0UT = -42.959 DBH GAIN=-24 .959 DB V IN = .03981 V 0UT = .002247 VOLTS GAIN= .056435 P IN = -16 P OUT = -40.958 DBM GAIN=-24 .958 DB V IN = .05012 V OUT = .002829 VOLTS GAIN= .056438 p IN = -14 P OUT = -38.958 DBM GAIN=-24 .958 DB V IN = .0631 V OUT= .003561 VOLTS GAIN= .056442 p IN = -12 p OUT = -36.956 DBM GAIN=-24 .956 DB V IN = .07943 V OUT = .004484 VOLTS GAIN= .056452 p IN = -10 p OUT = -34.953 DBM GAIN--24 .353 DB V IN = . 1 V OUT = .005647 VOLTS GAIN= .05647 p IN = -8 p 0UT = -32.947 DBM GAIN=-24 .947 DB V IN = .1259 V OUT = .007115 VOLTS GAIN= .056513 p IN = -6 p OUT = -30.929 DBM GAIN=-24 .929 DB V IN = .1585 V OUT = .003975 VOLTS GA1'N = - .056631 p IN = -4 p 0UT = -28.867 DBM GAIN=-24 .867 DB V IN = . 1995 V OUT = .01138 VOLTS GAIN= .057035 p IN = -2 p OUT = -26.593 DBM GAIN=-24 .593 DB V IN = .2512 V OUT = .01479 VOLTS GAIN= .058861 p IN = 0 p OUT = -23.!27 DBM GAIN--23 . 127 DB IN = .3162 V OUT = .02204 VOLTS GAIN= .069635 P IN = 2 p OUT = -16.232 DBM GAIN=-!8. .232 DB V IN = .3981 V OUT = .04874 VOLTS GAIN= .12243 10 20 30 40 50 60 70 80 90 100 110 120 130 140 150 * * * * * * * * ** PROGRAM POLY1****** * * * * * 125 PROGRAM FOR CURVE FITTING WITH ODD ORDER POLYNOMIAL OF ORDER 3 WRITTEN BY WLODYKA ACIMOVIC LAST REV JUL 19/85 PRINTER IS 1 DIM Y(100),X(100).Sum(19) DIM A<2,3),Kr<2000> C O O ) Eps=1 PRINT -32 LEAST TERMS" SQUARE POLINOMIAL APPROX. FOR FIRST AND THIRD ORDER 160 170 180 190 200 210 220 230 240 250 260 270 280 290 300 310 320 330 340 350 360 370 380 390 400 410 420 430 440 450 460 470 480 490 500 510 520 530 540 550 560 570 580 590 600 N PRINT INPUT Broj=N M=2 Red = M PRINT " PRINT " PRINTER PRINT PRINT " PRINT " PRINTER FOR 1=1 PRINT " INPUT X ( I ) PRINTER IS ENTER NUMBER OF DATA PAIRS Enter th< Format i< IS 701 3RD ORDER X ) i d a t a i n p a i r s ; X < i n p u t ) . Y C o u t p u t ) " ODD POLYNOMIAL CURVE FIT-IS 1 TO N Enter d a t a Y d ) 701 p a i r number " : I ;Y(I) 1 PRINT X ( I ) PRINTER IS NEXT I FOR 1=1 TO N Sumd )=Sum<1 ) + ( X ( I ) ) Sum<2)=Sui7i(2) + ( X d ) ) Sum<3)=Sum(3)+(X(I)) Sum<4)=Suir.<4) + Y< I)*X<I) Sum(5)=Sum(5)+Y(I)*(X<I))'3 NEXT I -2 '4 "6 A d A d A d A<2 1 )=Sumd ) 2) =Sum(2) 3) =Sum<4) 1>=Sum<2> A(2.2>=Sum<3) A<2.3>=Sum<5) PRINT A(*) M = 3 N = 2 Kk=0 J i = 0 FOR K=1 TO N Jj=Kk+1 L l = J j Kk=Kk + 1 PRINT PRINTER IS 1 PRINT J j IF ABS(AcJj.Kk))-Lps>0 AND KkOM THEN GOTO 650 610 IF Kk =M THEN A<Jj,Kk>=0 620 IF Kk = M THEN GOTO 860 630 Jj=Jj+1 640 GOTO 600 650 IF L l - J j = 0 THEN GOTO 710 660 FOR Mm=1 TO M 670 Atemp=A<Ll,Mm> 680 A(L1,Mm)=A<JjfMm> 690 A( J j,Mm)=Atemp 700 NEXT Mm 710 FOR L j = l TO M 720 J=M+1-Lj 730 A(K,J)=A<K.J)/A<K,K) 740 NEXT L j 750 FOR 1=1 TO N 760 FOR Lj=1 TO M 770 J=M+1-L; 780 IF I-K=0 THEN GOTO 800 790 A<I,J)=A<I,J>-A<I,K>*A<K.J) 800 NEXT L j 810 NEXT I 820 NEXT K 830 PRINTER IS 701 840 PRINT " COEFFICIENTS ARE:" 850 PRINT 860 FOR 1=1 TO L I 870 C(I)=A(I.M) 880 PRINT "C(";I-1;">=":DROUND(C<I),4) 890 NEXT I 900 GINIT 910 ALPHA OFF 920 M=3 930 Red=3 940 GRAPHICS ON 950 FOR J=1 TO B r o j 960 Kr(J)=Cn)*X<J)+C<2)*<X(J))"3 370 PRINT " Approximation i s Y'= ":DROUND(Kr(J).5):" f o r X= ";X<J):"Y = ":Y<J) 980 PRINT "ERROR = ";DROUND(Y(J)-Kr<J>,5) 990 PRINT 1000 NEXT J 1010 INPUT YS 1020 C(4)=C(2) 1030 C(3)=0 1040 C<2)=C(1) 1050 C(1)=0 1060 MOVE X(1)*10.Kr<1)*10 1070 FOR W=2 TO B r o j 1080 B=<Kr(W)-Kr<W-1 ) )*10 1090 R = X<W)-X(W-1 ) 1100 R=R*10 1110 NEXT W 1120 Najmx=ABS(X<1)) 1130 Najmy=ABS(Y(1)) 1140 Le=X<1) 1150 Ri=X(1) 1160 Bo=Y<1) 1170 U P=Y(1) 1180 FOR 1=2 TO N . 1190 IF X<IXLe THEN Le = X<I) 1200 IF X(I)>Ri THEN Ri=X(I) 1210 IF Y d X B o THEN Bo = Y( I ) 1220 IF Y(I)>Up THEN U P=Y(I) 1230 IF ABS<X<I)><Najmx THEN Najmx=ABS<X<I>> 1240 IF ABS<Y(I>XNajmy THEN Najmy=ABS(Y( I) ) 1250 NEXT I 1260 IF Najmx=0 THEN GOTO 1320 1270 IF Najmy=0 THEN GOTO 1320 1280 Le=<Le/Najmx) 1290 Ri=(Ri/Najmx) 1300 Bo=<Bo/Najmy) 1310 Up=<Up/Najmy) 1320 Cx=<Ri-Le>/1000 1330 Cy=(U P-Bo)/1000 1340 IF Le<0 AND Bo<0 THEN SHOW Le/Cx,Ri/Cx.Bo/Cy,Up/Cy 1350 IF Le<0 THEN SHOW Le/Cx,Ri/Cx,0,U P/Cy 1360 IF Bo<0 THEN SHOW 0,Ri/Cx.Bo/Cy,U P/Cy 1370 IF Le>=0 AND Bo> = 0 THEN SHOW 0.Ri/Cx,0,Up/Cy 1380 AXES Najmx/Cx,Najmy/Cy.0.0 1390 FOR 1=1 TO B r o j 1400 MOVE X(I)/Cx,Y<I)/Cy 1410 IMOVE -10,-10 1420 IDRAW 0,20 1430 IDRAW 20.0 1440 IDRAW 0.-20 1450 IDRAW -20,0 1460 NEXT I 1470 MOVE X<1)/Cx,Kr<1)/Cy 1480 FOR I=<X(1)/Cx) TO Ri/Cx 1490 Kr(I)=X( 1 )/Cy 1500 FOR M=2 TO Red+1 1510 Kr(I>=C(M)*<I*Cxr <M-1 >+KrU> 1520 NEXT M 1530 IF I=X(1)/Cx THEN GOTO 1560 1540 G=Kr(I)-Kr<I-1 ) 1550 IDRAW 1.G/Cv 1560 NEXT I 1570 END 3RD ORDER ODD POLYNOMIAL CURVE FIT X Y 128 .0793 .004484 . 1 .005647 .1259 .007115 .1535 .008375 .1995 .01138 .2515 .01479 .3162 .02204 COEFFICIENTS ARE: C( 0 )= .05142 C( 1 )= .1695 Approximation i s Y': ERROR = .00032218 Approximation i s Y': ERROR = .0003359 Approximation i s Y': ERROR = .00030347 Approximation i s Y': ERROR = .00015064 Approximation i s Y': ERROR = -.00022334 Approximation i s Y': ERROR = -.0008375 Approximation i s Y'= ERROR = .00042367 .0041613 .0053111 .0068115 .0088244 .011603 .015627 .021616 f o r X = f o r X = f o r X = f o r X = f o r X = f o r X = f o r X = 0793 Y= .004484 1 Y= .005647 .1259 Y= .007115 1585 Y= .008975 1995 Y= .01138 2515 Y = 3162 Y= 01479 02204 10 20 30 40 50 fin 70 80 80 100 110 120 130 140 150 160 170 180 190 200 210 220 230 240 250 260 270 280 290 300 310 320 330 340 350 360 370 380 390 400 410 420 430 440 450 460 470 480 490 500 510 520 530 540 550 560 570 580 590 600 610 129 * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * TWT LINEARIZER INTERMOD MEASUREMENT * * * * PROGRAM "INTERMOD" **** ******************************************* AUTO ATTENUATOR ADJUSTMENT AUTOMATIC 3RD AND 5TH ORDER IM MEASUREMENT AND TWO VOLTMETER MEASUREMENTS WITH SELECTABLE LO FREQUENCY MULTICARRIER AND WIDE BAND MEASUREMENTS BY M, P. W. WLODYKA ACIMOVIC STREET LAST REVISION: NOV 6/85 GCLEAR « PRINTER IS 1 **** INITIALIZE MEMORY AND VARIABLES * * * * * COM R1,Spn.Spnd,Cfr.Anc.Pmlc,Pm2c,Soa,Spap,Noise,Lo COM Cf1,Cf2,Nc DIM Nim(12),Freq(12>.Nim2(12>,Cs(12>.Cs2<12) DIM J<6,50),S<5,50),Mag<7),B<5,50). T<1:6,1:50) DIM St(3.50> DIM D(21) » COL 1.2,3 ARE ANAL,PM1, AND PM2 READINGS ASSIGN @Path5 TO "PLT1" ENTER «Path5;TO> ALLOCATE Remark$<6)[80] ASSIGN GPIB ADDRESSES TO EXTERNAL DEVICES SPECTRUM ANALYZER IS 18 POWER METER #1 IS 29 POWER METER #2 IS 13 POWER METER #3 IS 12 SWEEPER #1 IS !9<HP> SWEEPER *2 IS 25<AILTECH) ATTENUATOR IS 16 MULTIMETER 1 IS 14 MULTIMETER 15 ASSIGN ASSIGN ASSIGN ASSIGN ASSIGN ASSIGN ASSIGN ASSIGN ASSIGN OUTPUT OUTPUT OUTPUT OUTPUT OUTPUT 7 713 719 C«iSa TO 718 <siPm1 TO 729 TO TO TO TO 725 TO 716 TO 714 TO 715 "9D-V" "9D-V" $Pm3:"9D-V" <*Mm1 :"FIR0S0T0Y1" QMm2:"F1R0S0T0Y1" <S'Pm2 @Pn>3 S>SOJ 1 «?Sw2 @Att €>Mm1 'S<Mm2 ®Pm1 ; *Pm2; 620 630 640 650 660 670 680 690 700 7J0 720 730 740 750 760 770 780 790 300 810 320 830 840 850 860 870 880 890 900 910 920 930 940 950 960 970 980 990 1000 1010 1020 1030 1040 1050 1060 1070 1030 1090 1100 1110 1120 1130 1140 1150 1160 1170 1 180 1190 1200 1210 1220 ENTER @Mm1;Volt1;V1$ ENTER @Mm2;Volt2:V2S> PRINT V o l t ! , V 1 $ . V o l t 2 . V 2 $ ******* ASSIGN PATHS TO DATA FILES ******** P1 IS DATA FILE FOR ABSOLUTE POWER MEASUREMENTS OF FUNDAMENTAL. THIRD AND FIFTH ORDER POWER LEVELS. ALONE IS DATA FILE FOR TWT ALONE. ATT IS DATA FILE FOR ATTENUATOR SETTINGS RLCFSP IS DATA FILE FOR INITIAL VALUES OF SPECTRUM ANALYZER. REFERENCE LEVEL CENTRE FREQUENCY MAX FREQ SPAN MIN FREQ SPAN ANAL COUPLING FACTOR PM1 COUPLING FACTOR PM2 COUPLING FACTOR SINGLE TONE SAT POWER (ANAL) SINGLE TONE SAT POWER (PM2) 130 PRODUCT OF LIST FOR MULTICARRIER INTERMOD I=NUMBER !NIM<1)=0 !NIM(2)=0 !NIM(3)=3 !NIM(4)=12 !NIM(5)=30 !NIM(6)=60 !NIM(7>=105 !NIM(8)=168 !NIM(3)=252 !NIM(10)=360 !NIM(11)=495 !NIM(12)=660 t ASSIGN @Path3 ASSIGN QPath6 ASSIGN @Path7 ASSIGN @Path11 ASSIGN GPath.2 TO "INT" FOR 1=1 TO 12 ENTER ePa th.2;Nim<I).Nim2<I>.Cs(I) NEXT I CARRIERS (NO 1=1 TO 1 NIM2(1)= 0 CS(1)= 0 ^CS2(1)= 0 NIM2(2)= 2 CS(2)= 0 CS2(2)= 0 NIM2(3)= 6 CS(3>= 1 CS2(3)= 0 NIM2(4)= 12 CS(4)= 2 CS2(4)= 1 NIM2(5)= 20 CS(5)= 4 CS2(5)= c NIM2(6)= 30 CS(6)= 7 CS2(6)= 2 NIM2(7)= 42 CS(7)= 11 CS2(7)= c NIM2(8)= 56 CS(8>= 15 CS2(8)= 3 NIM2(9)= 72 CS(9)= 20 CS2(9)= 4 NIM2(10) =90 CS(10) =26 CS2(10) = 4 NIM2(11) = 110 CS(11) = 33 CS2(11) = 4 NIM2(12) = 132 CS(12) = 40 CS2C12) = 5 TO "RLCFSP" TO "MC" TO "FREQ" TO "DBTV" ( I ) * READ INITIAL SPECTRUM ANALYZER SETTINGS OUTPUT $Sa;"IP" OUTPUT $Sa:"KSX" ENTER @Path3:Rl,Cfr.Spn.Spnd.Anc.Pm1c.Pm2c.Spa,Spap,Noise,Lo PRINT "Reference L e v e l i s ":Ri:"dBm" PRINT "Center Frequency i s ":Cfr:"GHz" PRINT "LO FREQUENCY IS ":Lo:"GHz" PRINT "Span wide i s ":Spn;"MHz" PRINT "Span narrou is";Sond;"KHz" PRINT 1230 PRINT "ANAL COUPLING=";Anc:"DB" 1240 PRINT "PM1 COUPLING=";Pm1c:"DB" 1250 PRINT "PM2 COUPLING=":Pm2c:"DB" 1260 PRINT 1270 PRINT 1280 PRINT "SAT SINGLE TONE POWER ON ANAL";Spa;"DBM" 1290 PRINT "SAT SINGLE TONE POWER ON PM2";Spap;"DBM" 1300 PRINT "ENTER MAX NOISE POWER ON PM2";Noise:"DBM" 1310 PRINT 1320 PRINT " I f you want to change a n y t h i n g type Y" 1330 INPUT Y1£ 1340 IF Y1$="Y" THEN GOSUB E t e r 1350 ! WIDE BAND SECTION 1360 ! 1370 PRINTER IS 1 1380 PRINT "WIDE BAND MEASUREMENTS" 1390 INPUT WS 1400 PRINT "ENTER NUMBER OF RUNS" 1410 INPUT Mode 1420 IF W$="Y" THEN 1430 PRINT "ENTER FREO STEP SIZE IN MHZ" 1440 INPUT Fstep 1450 ELSE 1460 BEEP 1470 END IF 1480 F]ag0=0 1490 Mult=0 1500 Nc=2 1510 Noise=Noise+Pm2c 1520 ! 1530 !**INPUT TITLE AND COMMENTS TO APPEAR ON *** 1540 !*********** INTERMOD PLOT *************** 1550 ! 1560 ******* MULTICARRIER OR TWO TONE ********** 1570 »*********MEASUREMENT SELECTION *********** 1580 ! 1590 PRINT 1600 PRINT "MULTICARRIER OR TWO TONE MEASURMENT (M/T)" 1610 INPUT MIS 1620 IF M1$="M" THEN Mult=1 1630 IF Mult=0 THEN GOTO T i t 1640 IF Qq>1 THEN GOTO T i t 1650 ENTER €>Path6 j Cf 1 , Cf 2 ,Nc. Spnd 1660 PRINT 1670 PRINT "SLOT FREO=";Cf2:"GHZ" 1680 PRINT "NUMBER OF CARRIERS=":Nc 1690 PRINT "SPAN";Spnd;"KHZ" 1700 PRINT 1710 Spndw=Spnd 1720 FOR 1=1 TO Nc 1730 ENTER ®Path7.I:F<I) 1740 PRINT "FREQ(":I;">=":F<I>:"GHZ" 1750 F i ( I ) = (Fa)-<Lo+.21 )>*1000 1760 NEXT I 1770 PRINT 1780 PRINT "CHANGE ANYTHING?* Y/N>" 1790 INPUT C$ 1800 IF C$="Y" THEN GOSUB E t e r 2 1810 Tit:FOR Oq=1 TO Mode 1820 PRINTER IS 1 1830 ASSIGN §Path1 TO "P1" 1840 ASSIGN QPath2 TO "ATT" 1850 ASSIGN GPath4 TO "ALONE" 1860 ASSIGN @Path5 TO "PLT1" 1870 Remark$<1)="TWTA W/O LIN. TWO-TONE 3RD/5TH ORDER IM" 1880 IF W$="Y" THEN GOSUB Wide 1890 IF Mult-T THEN Remarks*1)="TWTA W/O LIN. MULTICARRIER 3RD ORDER I M * * 1900 IF Mult=0 THEN Nc=2 1910 IF Qq>1 THEN GOTO 1940 1920 PRINT "TWTA LINEARIZED? <Y FOR LINEARIZED)" 1930 INPUT S$ 1940 IF S$="Y" THEN Flag0=l 1950 IF FlagO=t THEN Remark$<1)="LINEARIZED TWTA TWO-TONE 3RD/5TH ORDE R IM" 1960 IF Mult=1 AND Flag0=1 THEN Remark$<1)="LINEARIZED TWTA MULTICARRI ER 3RD OR DER IM" 1970 ! 1980 IF FlagO=0 THEN GOTO N l i n 1990 IF Qq>1 THEN GOTO Rep 2000 PRINT "INPUT NUMBER OF DATA POINTS IN NON-LINEAR FILE" 2010 INPUT Count 2020 N l i n : ! 2030 FOR 1=2 TO 3 2040 PRINT 2050 PRINT "Do You have some remark <freq . . le v . )" 2060 PRINT "Remark No. " ; I 2070 INPUT RemarkS(I) 2080 NEXT I 2090 Rep:! 2100 ! 2110 OUTPUT $Sa;"CR" 2120 Freq: ! 2130 ! 2140 LOOP FOR VARIOUS ATTENUATOR SETTINGS ** 2150 ! 2160 A t t e n : ! 2170 FOR Br=1 TO 50 2180 P o s l e d n j i ^ B r 2190 ON END ®Path11 GOTO Stop 2200 ! 2210 OUTPUT ftSa;"KSoKSmA4KSnKSpA1" 2220 ! 2230 «********AUT0 ATTENUATOR ******************* 2240 ! 2250 ENTER C«>Path2;G 2260 OUTPUT ®Att:"B123456" 2270 Z=0+1 2280 ENTER QPathl1,Z:D<Q) 2290 IF D(O)=0 THEN 2250 2300 OUTPUT <s>Att:"A";D<Q> 2310 WAIT 1 2320 S t a r t : ! 2330 ! 2340 !** HERE YOU CAN DETERMINE YOUR MES. SET *** 2350 ! 2360 IF Mult=1 THEN 2380 2370 GOSUB Korak 2380 GOSUB Merenje — 2390 ! 2400 ! STORE ABSOLUTE MAIN. THIRD. AND FIFTH IM 2410 2420 2430 2440 2450 2460 2470 2480 2490 2500 2510 2520 2530 2540 2550 2560 2570 2580 2590 2600 2610 2620 2630 2640 2650 2660 2670 2680 2690 2700 2710 2720 2730 2740 2750 2760 2770 2780 2790 2800 2810 2820 2830 2840 2850 2860 2870 2880 2890 2900 2910 2920 2930 2940 2950 .21 >* ;"MHZ 2960 2970 2980 2990 ! TONE POWER LEVELS AT THIS ATTEN. SETTING t Beck: ! FOR K=1 TO 6 J(K,Br>=Mag<K> PRINT "J<K,Br>=J(":K:".":Br;")" OUTPUT @Path1;J(K,Br> NEXT K NEXT Br ********* END OF ATTENUATOR LOOP ********* Stop: ! NORMALIZATION OF ABSOLUTE PWR MEASUREMENTS OF FUNDAMENTAL AND IM TONES RELATIVE CARRIER TO INTERMOD RATIOS ARE CALCULATED Normali z a c i j a : ! Naj=-100 Flag9=0 Flag10=0 FOR Ik=1 TO P o s l e d n j i - 1 FOR Ii=2 TO Var1 PRINT J<1,Ik>,J<Ii,Ik> J < I i , I k ) = J ( 1 , I k ) - J ( I i . I k ) ! CALCULATION OF 3RD AND 5TH IM IF J ( I i , I k ) > 7 0 AND I i = 2 THEN Flag9=1 IF Flag9=1 THEN J(2,Ik)=70 IF J ( I i . I k ) > 7 0 AND Ii=3 THEN Flag10=1 IF Flag10=1 THEN J<3.Ik)=70 NEXT I i NEXT Ik t FOR P=1 TO P o s l e d n j i - 1 IF J<1.P)>Naj THEN Naj=J(1.P) ! CALCULATION OF MAX CARRIER POWER NEXT P Sp=Naj ********* PRINT TABLE OF IM VALUES ******** PRINTER IS 701 PRINT RemarkSd ) PRINT PRINT Remark$<2) PRINT Remark$(3) PRINT "VOLTAGE#1 = " : V o l t 1 : V I S ; " Spc=Sp+Anc Spac=Spa+Anc IF Mult=0 THEN Nc=2 IF W$="Y" THEN PRINT "RF1=":F<1>:"GHZ RF2=":F<2);"GHZ" PRINT "IF1=";<F<1)-Lo-.21>*1000:"MHZ 1000 133 VOLTAGES = ";Volt2:V2S lF2=";<F(2)-Lo-ELSE FOR 1=1 TO Nc PRINT "RF<";I:"> = ":F<I>:"GHZ I F ( " : I;") =":Fi(I>;"MHZ" NEXT I 134 3000 END IF 3010 PRINT " NUMBER OF CARRIERS=":Nc:"TEST#";Oq 3020 PRINT "MAX. MEASURED SINGLE CARRIER POWER = ";DR0UND(Spac,4);"DBM * • 3030 PRINT "MAX. MEASURED POWER PER CARRIER ON ANAL = ":DR0UND(SP,4>;" DBM" 3040 PRINT "MAX. MEASURED POWER PER CARRIER = ";DROUND<Spc,4);"DBM" 3050 PRINT 3060 PRINT 3070 PRINT "SPECTRUM ANALYZER MEASUREMENTS POWER METER MEASUREM ENTS" 3080 PRINT 3090 PRINT "O/P B/O 3RD ORDER 5TH SSPA (DBM) (DB)" 3100 PRINT " (DB) IM(DBC) IM(DBC) INPUT OUTPUT GAIN OPBO IMP IMR " 3110 PRINT " 3120 FOR P=1 TO P o s l e d n j i - 1 3130 IF Mult=0 THEN Nc=2 3140 J(1,P)=S Pa-J(1.P)-10*LGT(Nc) 3150 ! 3160 ********** CALCULATION OF OUTPUT B/O ******** 3170 f * * * * * * * * * REL TO SINGLE CARRIER SAT.******** 3180 ! 3190 Im3P=0 3200 lmpu=0 3210 lmrc=0 3220 lmr = 0 3230 PRINTER IS • 1 3240 Im3=10~(-J(2,P>/10> 3250 Im5=1(T<-J<3,P>/10> 3260 Pmr=(10^(J(5.P)/10))-(10~(Noise/10>) 3270 Gain=10*LGT(Pmr)-J(4 tP) 3280 IF Mult=1 AND Nc>2 THEN 3290 Im3p=Im3*J(6.P) 3300 Imptd = Im3 P*Nim(Nc) + ( (Im3p/4)*Nim2(Nc>) 3310 Impuc=Im3p*Cs(Nc)+(Im3p/4)*Cs2(Nc) 3320 Im3c=In>puc/J(6,P) 3330 Imrc=10*LGT(Im3c) 3340 ! J(6,P) IS AVERAGE CARRIER POWER IN MILLIWATTS 3350 ! IM3P IS IM POWER OF MEASURED SLOT IN MILLIWATTS 3360 Im3=Impu/(J(6,P>*Nc) 3370 Imr=10*LGT(Im3) 3380 ELSE 3390 ! END MULTICARRIER CORRECTION 3400 END IF 3410 IF Mult=1 THEN Im5 = 0 3420 Cor=1+Im3+Im5 3430 Imc=10*LGT((Pmr/Cor)) 3440 PRINT Cor.Imc,Noise 3450 PRINTER IS 701 3460 0pbo=Spap+Pm2c-Imc 3470 ! IMR = TOTAL IM RATIO IN DB 3480 ! IMPW = IM POWER IN MILLIWATTS 3490 ! IMRC = IM RATIO IN CENTRE SLOT IN DB 3500 ! J(2.P) BECOMES CENTRE SLOT IM RATIO IN MULTICARRIER SYSTEM 3510 Im P=J(2.P) 3520 IF Nc>2 THEN J(2,P)=-Imrc 3530 PRINT USING "4D.2D":J(1,P).J(2,P>.J(3.P),J(4,P>.J(5,P).Gain,Opbo . Imp.Imr 3540 ! J<1.P)=0 Pbo 3550 NEXT P 3560 IF Flag0=0 THEN GOSUB Alone 3570 PRINT CHR$<12) 3580 ! 3590 ********** END OF NORMALIZATION ROUTINE 3600 ! 3610 ! 3620 PRINTER IS 1 3630 Yy$="N" 3640 IF Yy$= MN" THEN GOTO 3670 3650 GOTO 4580 3660 ! 3670 j ********* PLOTTING ROUTINE *********** 3680 ! 3690 GIIMIT 3700 DUMP DEVICE IS 701,EXPANDED 3710 WINDOW -100,250,-150,650 3720 ! 3730 !********* DRAW X AND Y AXIS ********** 3740 ! 3750 MOVE -30.0 3760 DRAW 200,0 3770 MOVE 0,0 3780 DRAW 0.600 3730 LINE TYPE 4 3800 ! 3810 j * * * * * * * * * DRAW DIVISIONS ON GRAPH **** 3320 ! 3830 FOR 1=0 TO 200 STEP 10 3840 MOVE 1.0 3850 DRAW 1.600 3860 NEXT I 3870 LINE TYPE 1 3880 FOR 1=0 TO 200 STEP 50 3890 MOVE 1 .0 3900 DRAW 1,600 3910 NEXT I 3920 LINE TYPE 4 3930 FOR 1=0 TO 600 STEP 20 3940 MOVE 0 , 1 3950 DRAW 200 . 1 3960 NEXT I 3970 LINE TYPE 1 3980 FOR 1=0 TO 600 STEP 100 3990 MOVE 0 , 1 4000 DRAW 200 . 1 4010 NEXT I 4020 LINE TYPE 1 4030 MOVE -20.0 4040 CSIZE 3,.6 4050 ! 4060 !********* PRINT LABELS *************** 4070 ! 4080 FOR 1=100 TO 600 STEP 100 4090 MOVE -20 . 1 4100 LABEL 1/10 4110 NEXT I 4120 FOR 1=0 TO 200 STEP 50 4130 MOVE 1-10,-30 4140 LABEL 1/10 4150 4 1 GO 4170 4180 4190 4200 4210 4220 4230 4240 4250 4260 4270 4280 4290 4300 4310 4320 4330 4340 4350 4360 4370 4380 4390 4400 4410 4420 4430 4440 4450 4460 4470 4480 4490 4500 4510 4520 4530 4540 4550 4560 4570 4580 4590 4600 4610 4620 4630 4640 4650 4660 4670! 4680 4690 4700 4710 4720 4730 4740 4750 NEXT I ********* DRAW THIRD ORDER IH CURVE ******* MOVE J<1,1>*10.J<2,1)*10 FOR 1=2 TO P o s l e d n j i - 1 IF J(2.I>>70 THEN GOTO 4250 DRAW J(1 .I)*10.J<2.D*10 NEXT I IF FlagO=0 THEN GOTO F i f t h *** DRAW NON-LINEARIZED THIRD ORDER ******* ***** CURVE FOR COMPARISON *************** GOSUB A l o n e r e t MOVE B(1,1>*10,B<2,1)*10 FOR 1=2 TO Count IF B<2,I)>70 THEN GOTO F i f t h DRAW B<1,I>*10.B<2,I>*10 NEXT I 136 * * * * * * * ** DRAW FIFTH ORDER IM CURVE ******^ F i f t h : IF Mult=1 THEN GOTO Lab MOVE J<1,1>*10,J<3.1)*10 FOR 1=2 TO P o s l e d n j i - 1 IF J<3.1>>60 THEN GOTO 4480 DRAW J ( 1 . I ) * 1 0 . J ( 3 . I ) * 1 0 NEXT I * * * * * * * * PRINT TITLES AND COMMENTS Lab: MOVE 0,610 LABEL Remarks(1> MOVE 0.-70 LABEL RemarkS(2> MOVE 0,-140 LABEL Remark$(3) MOVE 0.-115 LABEL "MAX. POWER GOTO End :DR0UND(SP.4);"DBM CARRIERS=";Nc ********** HP PLOTTER SECTION ************** » PRINT "LOAD PROGRAM PLOT" GOTO End ******************************************* ********* SUBROUTINE SECTION ************** ******************************************* ********** SUBROUTINE KORAK ***************** t Korak: •! L = 0 FOR R=1 TO 5 OUTPUT GSa: "RL".: R1: "DMCF" : Cf r ; "GZSP" : S Pn: "MZTSM1 TSCR" OUTPUT @Sa;"ElSP":Spnd:"KZ" OUTPUT $Sa;"TSM1E1M3KSK0A" ENTER QSajCfss 4760 4770 4780 4790 4800 4810 4820 4830 4840 4850 4860 4870 4880 4890! 4900 4910 4920 4930 4940 4950 4960 4970 4980 4990 5000 5010 5020 5030 5040 5050 5060 5070 5080 5090 5100 51 10 5120 5130 5140 5150 5160 5170 5180 5190 5200 5210 5220 5230 5240 5250 5260 5270 L":R1:"DB i f 5280 ";R7:"DB" 5290 5300 5310 5320 5330 IF Cfss<0 THEN G= + 1 IF Cfss>0 THEN G=-1 L=L+G NEXT R G=SGN<L> Cfss=ABS<Cfss> OUTPUT »Sa;"SS";Cfss:"HZ" OUTPUT @Sa;"CF";Cfr+G*Cfss*1.E-9/2:"GZ" OUTPUT $Sa:"E3" Brp = 0 Fleg2=1 RETURN i *********** SUBROUTINE MERENJE ************** i Merenje: ! IF Mult=1 THEN GOTO Mwl Fleg4=0 Fleg5=0 Fleg6=0 Fleg3=0 Fleg2=0 Brp = 0 Poc: IF Mult=1 THEN GOTO Mwl OUTPUT @Sa;"M1TSE1E2TS" Brp=Brp+1 » i******** CHECK TO SEE IF IN TROUBLE WITH**** ********* LEVEL MEASUREMENT. IF YES **** ,******** G 0 T 0 KORAK SUBROUTINE AGAIN **** t IF Brp=5 THEN GOSUB Korak Fleg1=0 OUTPUT @Sa;"MlElMT1SP":Cfss/2;"HZCRRBUP" OUTPUT @Sa;"TSMT0" Mwl: Fleg1=0 i IF Mult=1 THEN var1=2 ELSE Var1=3 END IF Mactut=0 IF Mult=0 THEN Nc=1 FOR N-1 TO Nc Mt = 0 ASSIGN @Path7 TO "FREQ" FOR 1=1 TO Var1 IF Mult=1 THEN OUTPUT $Sa;"CR" Looping: ! IF Mult=1 AND 1=1 AND Qq--1 THEN ENTER @Path7 .N;F<N) IF Mult-1 AND 1=1 THEN OUTPUT §Sa;"CF";F(N);"GZSP";Spnd;"KZR IF Mult=1 AND 1=2 THEN OUTPUT «?>Sa; "CF" : Cf 2; "GZSP" ; Spnd: "KZRL IF Mult=1 AND L=1 THEN OUTPUT ®Sa;"TSE1E2RBUPTSE1" OUTPUT $Sa:"TSE1E2" OUTPUT (!>Sa;"MA"„ OUTPUT @Sa;"PP" ENTER »Sa:Ma 5340 IF 1=1 THEN OUTPUT @Sa;"RL";Ma+5:"DB" 5350 R7=Ma+5 5360 Gogo: ! 5370 ! PRINT I . F l e g l , F l e g 2 , F l e g 3 . F l e g 4 5380 IF 1=1 THEN OUTPUT @Sa;"DT<s>DA3072DW1026.450,2548D3LBMAIN TON E0DW1O28" 5390 IF 1=2 THEN OUTPUT @Sa:"DTQDA3072DW1026,450,2548D3LBTHIRD IN TERM0D«DW 1028" 5400 IF 1=3 THEN OUTPUT @Sa:"DT®DA3072DW1026,450,2548D3LBFIFTH IN TERMODSDW 1028" 5410 GOTO Cupa 5420 Opa: ! 5430 OUTPUT @Sa;"RC1" 5440 Mt=Gp 5450 Cupa: ! 5460 IF Fleg1=0 THEN Fleg3=1 5470 IF Fleg3=1 THEN GOTO Ntest 5480 Okok: ! 5490 IF Fleg5=1 THEN OUTPUT ®Sa:"RC1" 5500 IF Fleg6 THEN OUTPUT @Sa:"RC1" 5510 IF Fleg6=1 THEN GOTO Nena 5520 IF Fleg4=1 THEN OUTPUT @Sa:"RC1" 5530 IF Fleg1=1 THEN GOTO Poc 5540 Nena: IF G=1 THEN OUTPUT @Sa;"CFUP" 5550 IF G=-1 THEN OUTPUT $Sa;"CFDN" 5560 Fleg6=0 5570 NEXT I 5580 ! 5590 ! 5600 Pupa: ! 5610 OUTPUT @Sa:"DT@DA3072DW1026,450.2548D3LB SDW10 28" 5620 OUTPUT @Pml:"9D-V" 5630 ENTER @Pm1;Mag(4) 5640 Mag(4)=Mag<4>+Pm1c 5650 LOCAL 729 5660 OUTPUT ®Pm2;"9D-V" 5670 ENTER @Pm2;Mag<5) 5680 Mag<5>=Mag<S>+Pm2c 5690 LOCAL 712 5700 LOCAL 719 5710 RETURN 5720 ! 5730!***** SUBROUTINE ETER - OPTIONAL SUBROUTINE** 5 7 4 0 ? * * * * * TO CHANGE INITAL SPEC. ANAL. VALUES ** 5750 ! 5770 ASSIGN #Path5 TO "RLCFSP" 5780 PRINT "Input Ref Lev i n dBm" 5790 INPUT R l 5800 PRINT "Input Cen. F r . i n GHz" 5810 INPUT C f r 5820 PRINT "INPUT LO FREQ IN GHZ" 5830 INPUT Lo 5840 PRINT "Input Span enough to i n s p e c t s i g n a l i n MHz" 5850 INPUT Spn 5860 PRINT "Input Span narrow enough to measure d i f f e r e n c e betwee n two pea ks i n KHz" 5870 INPUT Spnd 5880 PRINT "INPUT ANAL COUPLING FACTOR IN DB" 5890 INPUT Anc 5900 PRINT "INPUT PM1 AND PM2 COUPLING FACTORS" 5910 INPUT Pm1c,Pm2c 5920 PRINT "INPUT SINGLE TONE SAT POWER ON ANAL IN DBM" 5930 INPUT Spa 5940 PRINT "INPUT SINGLE TONE SAT POWER ON PM2 IN DBM" 5950 INPUT Spap 5960 PRINT "INPUT MAX NOISE POWER ON PM2 IN DBM" 5970 INPUT Noise 5980 OUTPUT @Path5:R1.Cfr.Spn,Spnd,Anc,Pm1c.Pm2c,Spa,Spap.Noise,L o 5990 PRINT R l ,Cfr,Spn,Spnd,Anc,Pn.1c.Pm2c,Spa.Spap,Noise,Lo 6000 RETURN 6010 ! 6020 ! 6030!***********MEASURE FOR NOISt FLOOR*********** 6040 ! 6050 N t e s t : Kuku.ruz = 0 6060 Fleg5=0 6070 OUTPUT @Sa:"SV1" 6080 N t t : ! 6090 Kukuruz = KLLkuruz + 1 6100 IF Kukuruz=2 THEN Fleg6=1 6110 IF Fleg6=1 THEN GOTO Okei 6120 Fleg3=0 6130 Fleg4=0 6140 OUTPUT @Sa;"RB3KZ" ! CHANGED RB TO 3 KHZ 6150 IF 1=1 THEN OUTPUT @Sa:"LG1DB"! CHANGED TO 1 DB 6160 OUTPUT @Sa:"KSG10HZ" 6170 WAIT 3 6180 OUTPUT @Sa;"A3KSHE1MA" 6190 ENTER @Sa;Mact 6200 OUTPUT @Sa;"M2DNDNMA" 6210 ENTER @Sa;Mc1 6220 OUTPUT @Sa;"M2UPUPUPUPMA" 6230 ENTER €>Sa;Mc2 6240 OUTPUT @Sa;"A1" 6250 IF ABS(Mc1-Mact)<2 OR ABS(Mc2-Mact)<2 THEN Fleg4=1 6260 OUTPUT @Sa;"LG10DB" 6270 IF Fleg4=0 THEN GOSUB Caravg 6280 IF Fleg4=0 THEN GOTO Okej 6290 Tupa: Gp=0 6300 FOR Gaga=1 TO 3 6310 OUTPUT @Sa;"SPDNRBDN" 6320 OUTPUT $Sa;"TSE1MA" 6330 ENTER @Sa:Gr 6340 Gr=10'(Gr/10) 6350 Gp=GP+Gr 6360 NEXT Gaga 6370 IF Maq<1>-55>10*LGT(Gp/3> THEN Fleg6=1 6380 GOTO Ntt 6390 Okej: Fleg5=1 6400 NEXT N 6410 Mactut=0 6420 GOTO Okok 6430 ! 6440 ! 6450 Alone: ! 6460 FOR 1=1 TO P o s l e d n j i - 1 6470 FOR K=1 TO 3 6480 OUTPUT @Path4;J(K,I) 6490 T(K,I)=J<K,I> 6500 NEXT K 6510 NEXT I 6520 RETURN 6530 ! 6540 !**********SUBROUTINE ALONE RETURN********* 6550 ! 6560 A l o n e r e t : ! 6570 FOR 1=1 TO Count 6580 FOR K=1 TO 3 6590 ENTER @Path4;B(K.I> 6600 NEXT K 6610 NEXT I 6620 RETURN 6630 ! 6640 End: ! 6650 FOR 1=1 TO 50 6660 T<4,I)=J(1 , 1 ) 6670 T(5.I)=J<2.I) 6680 T ( 6 . I ) = J ( 3 , I ) 6690 NEXT I 6700 OUTPUT @Path5;T(*) 6710 PRINTER IS 701 6720 DUMP GRAPHICS 6730 PRINT CHR$<12) 6740 NEXT Qq 6750 STOP 6760 ! 6770'************SUBROUTINE ETER2***************** 6780'! 6790 E t e r 2 : ! 6800 ASSIGN SPathG TO "MC" 6810 ASSIGN §Path7 TO "FREQ" 6820 PRINT 6830 PRINT "INPUT SLOT FREQ IN GHZ" 6840 INPUT Cf2 6850 PRINT 6860 PRINT "INPUT NUMBER OF MULTICARRIERS" 6870 INPUT Nc 6880 PRINT 6890 PRINT "INPUT FREQ SPAN IN KHZ" 6900 INPUT Spnd 6910 Spndu=S Pnd 6920 FOR 1=1 TO Nc 6930 PRINT 6940 PRINT " FREQ(";I;"> IN GHZ: "; 6950 F(I>=0 6960 INPUT FCI) 6970 IF F(I)=0 THEN 6990 6980 OUTPUT @Path7,I;F(I) 6990 ENTER «Path7.I:F<I) 7000 PRINT F<I> 7010 NEXT I 7020 PRINT 7030 FOR X=1 TO Nc /040 ENTER 's»Path7.X:F<X> 7050 PRINT " F";X:"=":F(X):"GHZ" 7060 NEXT X 7070 PRINT 7080 OUTPUT @Path6;Cf1.Cf2,Nc.Spnd 7090 RETURN 7100 !******CARRIER AVERAGING SUBROUTINE********** 7110 Caravg: ! 7120 Mactui=10^<Mact/10> 7130 PRINT "MACTW F(" ;N:">= ";Mactw;"mW" 7140 Mactwt = Mactwt+r1actw 7150 IF N<<Nc) THEN RETURN 7160 PRINT "MACTWT = ":r1actwt 7170 Mactii.a=Mact(jt/<Nc> 7180 IF 1=1 THEN Mag(6)=Mactua 7190 IF 1 = 1 THEN Mag< 1 > = 1 0*LGT<Mactu.a> 7200 IF 1 = 2 THEN Mag<2) = 1 0«LGT<r1actii»t) 7210 PRINT "MAG(2>=";Mag(2);"DBM" 7220 IF 1=3 THEN Maq<3)=10*LGT<Mactwt> 7230 IF 1=1 THEN PRINT "MAG<1)=";Mag<1> 7240 RETURN 7250 Wide: ! 7260 IF Wd=0 THEN GOTO Fre 7270 F<1)=F(1)+<Fstep/1000) 7280 F(2)=F(2)+(Fstep/1000) 7290 Cf2=Cf2+<Fste P/1000) 7300 F r e : F1=F<1>-<Lo+.21> 7310 F2=F(2)-(Lo+.21) 7320 Wd=Wd*1 7330 OUTPUT @Sw2;"FR";F1:"GZ" 7340 OUTPUT @Su2:"AP":19;"-DBM" 7350 OUTPUT @Sui1 : "FR" ;F2: "GZ" 7360 OUTPUT @Su1:"AP";15.5:"-DB" 7370 Bp: BEEP 7380 RETURN 7390 END 10 20 30 40 50 60 70 80 30 100 110 120 130 140 150 160 170 180 190 200 210 220 230 240 250 260 270 280 290 300 310 320 330 340 350 360 370 380 390 ILTER T" 400 410 EWING 50KHZ 420 430 440 450 460 470 480 430 500 510 520 530 540 550 560 142 ****** PLTNPRAUTO ********** ******************************************* ****** PROGRAM TO MEASURE NPR ********** ****** OF THE LINEARIZER RACK ********** ****** USING THE SPECTRUM ANAL- ********* ****** ZER AT 14 GHz. ********** ****** ********** ****** WRITTEN BY: W. STREET ********* ****** M. WLODYKA * * * * * * * * * ( ****** LAST REVISION: JULY 3/85 ********** ******************************************* ******************************************* ******** DIMENSION VARIABLES ************ ******************************************* DIM 0 Pbo(21) DIM Npr<21) DIM D<21 ) DIM B<21> ASSIGN GPath2 TO "ATT" ASSIGN @Path11 TO "DBTV" ASSIGN @Sa TO 718 ASSIGN §>Pm2 TO 713 ASSIGN @Att TO 716 OUTPUT @Pm2:"9D-V" OUTPUT @Att:"A123456" ENTER @Path2;Ss Flag1=0 Flag2=0 ******************************************* ******** INITIAL INSTRUMENT SET-UP ******* ******************************************* PRINTER IS 1 OUTPUT QSa;"IP" OUTPUT @Sa;"CF14013988KZSP50KZRL0DM" PRINT "SET UP SPECTRUM ANALYZER TO THE CENTRE FREQUENCY OF THE F SLO PRINT PRINT "ENTER FREQUENCY OF THE CENTRE OF THE SLOT WHILE ON SPAN" INPUT Cf PRINT "INPUT SATURATED SINGLE CARRIER POWER ON POWER METER-INPUT Sat PRINT "LINEARIZED" INPUT Y5> IF Y$="Y" THEN Flag1=1 ALPHA OFF OUTPUT @Sa:"CF";Cf;"GZ" OUTPUT @Sa:"SP50KZCFUPUPUPUPV630HZLG5DBSTOA" ENTER «Sa:Sup WAIT Sup+1 OUTPUT <9>Sa:"E1MA" ENTER @Sa:Pk OUTPUT $Sa:"RL";Pk +2:"DB" t * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * VI 143 580 590 600 610 620 630 640 650 660 670 680 690 700 710 720 730 740 750 760 770 780 790 800 810 820 830 840 850 860 870 880 890 900 910 920 930 940 950 960 970 980 990 1000 1010 1020 1030 1040 1050 1060 1070 1080 1090 1100 1110 1120 1130 1 140 1150 1160 1170 1180 »***»***» MEASUREMENTS ***************** ******************************************* OUTPUT @Sa:"CTv"B3HZS2ST0A" ENTER @Sa:Sup1 OUTPUT @Sa:"ATUP" FOR N=0 TO 16 STEP Ss OUTPUT <j>Sa;"A1S2" IF N=10 THEN OUTPUT @Sa;"ATDN" Sup1=80 WAIT Sw P1/1 .5 OUTPUT <?>Sa:"A3E1 M2" :Cf ; "GZMA" OUTPUT @Sa;"M2KSNMA" ENTER <§>Sa:Y1 OUTPUT @Sa:"M2UPUPUPMA" ENTER @Sa;Y2m OUTPUT @Sa:"M2UPUPMA" ENTER @Sa:Y2n Y2=<Y2ni + Y2r>)/2 N Pr(N)=Y2-Y1 GOSUB C a l c PRINT "Npr(N>=":NPr<N> * * * * * * * * * * AUTO ATTENUATOR * * * * * * * * * * * * * * S t t : ENTER <s>Path2:I ON END €>Path2 GOTO 910 OUTPUT <5>Att;"B123456" 2 = 1 + 1 ENTER @Path11,Z;D<I) IF D<I)=0 THEN GOTO S t t OUTPUT @Att:"A";D(I> NEXT N GOSUB P i t GOTO F i n i s h xxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxx xxxxxxxxxxxx SUBROUTINE SECTION xxxxxxxxx xxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxxx * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * CALCULATIONS h ADJUSTMENTS * * * * * * * * ****** SUBROUTINE ******** * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * * C a l c : ! ENTER @Pm2:CtiiT. Cim=10~<Ctim/10> F=10'<(Npr(N>-2.30)/10> Opbo(N)=Sat-<10*LGT(Cim-(Cim/(F + 1> >>) RETURN ******************************************* ********* PLOTTING SUBROUTINE ************* ******************************************* P i t : GINIT ASSIGN SPath3 TO "OPBO" ASSIGN @Path4 TO "NPR" GRAPHICS OFF DUMP DEVICE IS 701.EXPANDED WINDOW -100.250.-250.680 1190 1200 1210 1220 1230 1240 1250 1260 1270 1280 1290 1300 1310 1320 1330 1340 1350 1360 1370 1380 1390 1400 1410 1420 1430 1440 1450 1460 1470 1480 1490 1500 1510 1520 1530 1540 1550 1560 1570 1580 1590 1600 1610 1620 1630 1640 1650 1660 1670 1680 1690 1700 1710 1720 1730 1740 1750 1760 1770 1780 1790 144 j i * * * * * * * * * DRAW X AND Y AXIS *************** i MOVE 0,0 DRAW 200.0 MOVE 0,0 DRAW 0.600 LINE TYPE 4 DRAW DIVISIONS ON GRAPH ********* FOR 1=0 TO 200 STEP 10 MOVE 1,0 DRAW 1,600 NEXT I LINE TYPE 1 FOR 1=0 TO 200 STEP 50 MOVE 1,0 DRAW 1,600 NEXT I LINE TYPE 4 FOR 1=0 TO 600 STEP 20 MOVE 0 , 1 DRAW 200 . 1 NEXT I LINE TYPE I FOR 1=0 TO 600 STEP 100 MOVE 0 , 1 DRAW 200 . 1 NEXT I LINE TYPE 1 MOVE -20,0 CSIZE 4,.5 I I * * * * * * * * * PRINT LABELS ******************** t FOR 1=100-15 TO 600-15 STEP 100 MOVE -20 , 1 LABEL<I+15>/10 NEXT I FOR 1=0 TO 200 STEP 50 MOVE 1-10.-40 LABEL 1/10 NEXT I MOVE -30,200 LDIR 3.14/2 LABEL "CARRIER/IM <dB>" LDIR 0 MOVE -35.-75 LABEL "OUTPUT BACKOFF RELATIVE TO SINGLE CARRIER SATURATION" MOVE 90,-110 LABEL " ( d B ) " MOVE -10,640 CSIZE 5..6 LABEL "CARRIER/IM vs OUTPUT BACKOFF" t *********** PRINT DATA ************* f ALPHA ON — PRINTER IS 701 PRINT "OPBO NPR" 1800 FOR N = 0 TO 20 STEP Ss 1810 PRINT DROUND(Opbo<N>,3);NPr(N) 1820 ! ]830 I ********** PLOT DATA ************* 1840 ! 1850 IF N=0 THEN MOVE Opbo(N)*10,Npr<N)*10 1860 IF 0 Pbo(N>=0 OR N Pr(N)=0 THEN 1880 1870 DRAW 10*<Opbo(N)),10*(N Pr(N)) 1880 NEXT N 1890 IF Flag2=1 THEN F i n i s h 1900 FOR N=0 TO 20 STEP Ss 1910 IF Flag1=0 THEN 1920 OUTPUT @Path3,N+1;Opbo(N> 1930 OUTPUT ®Path4.N+1;Npr(N> 1940 END IF 1950 IF Flagl=1 THEN 1960 ENTER @Path3,N+1:0 Pbo(N) 1970 ENTER @Path4,N+1;Npr<N) 1980 END IF 1990 NEXT N 2000 IF Flaq1=1 THEN Flag2=1 2010 IF Flag1=1 THEN GOTO 1800 2020 RETURN 2030 F i n i s h : END 

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