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UBC Theses and Dissertations

A spread spectrum modem for use on electrical power lines Van der Gracht, Peter Kenneth 1982

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A SPREAD SPECTRUM MODEM FOR USE ON ELECTRICAL POWER LINES by Peter Kenneth jvan der Gracht B.A.Sc. University of British Columbia, 1977 A THESIS SUBMITTED IN PARTIAL FULFILLMENT OF THE REQUIREMENTS FOR THE DEGREE OF MASTER OF APPLIED SCIENCE i n THE FACULTY OF GRADUATE STUDIES (DEPARTMENT OF ELECTRICAL ENGINEERING) We accept this thesis as conforming to the required standard THE UNIVERSITY OF BRITISH COLUMBIA January, 1982 ° Peter Kenneth van der Gracht, 1982 McGraw-Hill Book Company 1221 Avenue of the Americas New York, New York 10020 Telephone 212/997-2613 Copyrights and Permissions Department February 1 7 , 1 9 8 2 Peter van der Gracht Dept. of E l e c t r i c a l Engineering Library-Special Collections 1 9 5 6 Main M a l l University of B r i t i s h Columbia Vancouver, B.C., Canada V6T 1Y3 Dear Mr. Van der Gracht: Permission i s hereby granted for the use of the following material for the purpose specified i n your l e t t e r of January 2 9 provided the material i s o r i g i n a l i n the book and appears without c r e d i t to any other source. Baumeister: STANDARD HANDBOOK FOR ELECTRICAL ENGINEERS This permission i s granted with the understanding that your use of the material i s l i m i t e d to the s p e c i f i e d purpose and that a c r e d i t l i n e w i l l be footnoted on the f i r s t page of each quotation covered by t h i s permission or on the copyright page of the volume i n which the material appears. Where i l l u s t r a t i o n s are used, the c r e d i t l i n e should appear a f t e r the legend. The acknowledgments should read: "From ( t i t l e of book) by (author). Copyright (c) (date and owner). Used with the permission of McGraw-Hill Book Company". Figure no. 15-12 Sincerely, Pat Colomban Permissions Supervisor PC/km Peter van der Gracht Dept. of E l e c t r i c a l Engineering University of B r i t i s h Columbia January 29, 1982 McGraw-Hill Book Co. 1221 Avenue of the Americas New York, N.Y. 10020 Dear S i r s : As a graduate student i n E l e c t r i c a l Engineering at the U n i v e r s i t y of B r i t i s h Columbia, I am preparing my M.A.Sc. t h e s i s . The thesis w i l l be filmed by the National L i b r a r y , Ottawa, and copies of the f i l m l e n t or s o l d . May I have permission to use i n my t h e s i s , and the National L i b r a r y to f i l m , Figure no. 15-12 from the Standard Handbook for E l e c t r i c a l Engineers, 10 th ed., published by McGraw-Hill ? I should be very g r a t e f u l f or your favorable consideration of the request. Kindly send your permission to Joan Selby at the following address: L i b r a r y - S p e c i a l C o l l e c t i o n s 1956 Main M a l l University of B r i t i s h Columbia Vancouver, B.C., Canada V6T 1Y3 Sincerely, Peter van der Gracht In p r e s e n t i n g t h i s t h e s i s i n p a r t i a l f u l f i l m e n t o f the requirements f o r an advanced degree a t the U n i v e r s i t y o f B r i t i s h Columbia, I agree t h a t the L i b r a r y s h a l l make i t f r e e l y a v a i l a b l e f o r r e f e r e n c e and study. I f u r t h e r agree t h a t p e r m i s s i o n f o r e x t e n s i v e copying of t h i s t h e s i s f o r s c h o l a r l y purposes may be granted by the head of my department o r by h i s o r her r e p r e s e n t a t i v e s . I t i s understood t h a t copying or p u b l i c a t i o n o f t h i s t h e s i s f o r f i n a n c i a l g a i n s h a l l not be allowed without my w r i t t e n p e r m i s s i o n . Department of El.Ec TR/C/tc E/JtjiNEEKlMGf The U n i v e r s i t y of B r i t i s h Columbia 2075 Wesbrook P l a c e Vancouver, Canada V6T 1W5 Date DE-6 (2/79) ABSTRACT The design, implementation, and te s t i n g of a spread spectrum modem f o r use on e l e c t r i c a l power l i n e s are described i n t h i s t h e s i s . It i s shown that f o r data communication over power l i n e s , spread spec-trum s i g n a l l i n g has performance advantages over s i g n a l l i n g formats which use a narrow frequency band. Synchronization f o r the modem i s based on the 60 Hz power l i n e s i g n a l and s i g n i f i c a n t l y reduces the complexity of the modem. A spread spectrum transmitter and receiver are implemented and performance r e s u l t s on some representative power l i n e c i r c u i t s are presented. The modem performed well and appears to be a v i a b l e candidate f o r data communications over power l i n e s . i i TABLE OF CONTENTS Page ABSTRACT i i TABLE of CONTENTS . . . . . . . . . . . . . . . . . i i i LIST of TABLES . . . . . . . . v LIST of ILLUSTRATIONS . . . . . . . . . . . . . . . v i ACKNOWLEDGEMENTS . . . . . . . . . . . x 1 . INTRODUCTION TO POWER LINE CARRIER 1 1.1 Uses of Power Line Carrier (PLC) 1 1 .1 .1 Uses on High Voltage Lines 1 1 . 1 . 2 Uses on D i s t r i b u t i o n Lines 1 1 . 1 . 3 Uses on Intrabui l d i n g Lines 4 1.2 PLC Modem Requirements 6 1 . 2 . 1 Data Rates Required f o r PLC 6 1 . 2 . 2 Error Rates Tolerable f o r PLC 9 1 . 2 . 3 Switching A l t e r n a t i v e s f o r PLC 10 1.3 Outline of the Thesis . . . . . . . . . . . . . . 10 . 2 . REVIEW OF PREVIOUS WORK 12 2.1 Power System Transmission C h a r a c t e r i s t i c s . . . . 12 2 . 1 . 1 Power System Topology . . . . . . . . . . 12 2 . 1 . 2 Power Line Noise 20 2 . 1 . 3 Signal Propagation at Low Frequencies 26 2 . 1 . 4 Signal Propagation at High Frequencies 29 2 . 1 . 5 Received Signal to Noise Ratio Variations . . . . . . 32 2.2 PLC S i g n a l l i n g Techniques . . . . . . . . . . . . 35 2 . 2 . 1 Very Low Frequency S i g n a l l i n g . . . . . . . . . . . . 35 2 . 2 . 2 Ripple Control . . . . . . ...... . . . . . . . . . . . 36 2 . 2 . 3 Baseband S i g n a l l i n g 39 2 . 2 . 4 C a r r i e r S i g n a l l i n g . . . . . . . . . 41 2 . 2 . 5 Two-way S i g n a l l i n g near 50 KHz. . . . . . . . . . . . 41 2 .3 Spread Spectrum Communications 42 2 . 3 . 1 Direct Sequence Spread Spectrum . . . . . . . . . . . 43 2 . 3 . 2 Performance C h a r a c t e r i s t i c s of Spread Spectrum . . . 43 2 . 3 . 3 Disadvantages of Spread Spectrum f o r PLC. '. . . . . . 47 3 . PLC MODEM SYSTEM DESIGN . . . . . . . . . . . . . . . . . . . . . 48 3.1 Block Diagram f o r a Spread Spectrum PLC Modem 48 i i i Page 3.2 Synchronization 50 3.2.1 Code Synchronization i n Spread Spectrum Systems . . . 50 3.2.2 Code Synchronization by Power Line Zero Crossings . . 52 3.2.3 C a r r i e r and B i t Synchronization . . . . . 52 3.2.4 Implications of Line Synchronization 54 3.3 Frequency Band of Operation . . . . . 68 3.4 Line Coupling Network 72 3.5 The Code Generator . . . , 78 4. PLC MODEM IMPLEMENTATION 81 4.1 The Transmitter 81 4.1.1 The Transmitter Block Diagram . . 81 4.1.2 The Power Supply and Protection Network 81 4.1.3 The Code Generator . 84 4.1.4 The Data Spreader and Modulator 90 4.1.5 The Transmit Amplifier . 94 4.1.6 The Coupling Network 94 4.2 The Receiver . . . . . . . . . . . . . . . . . . . 101 4.2.1 The Receiver Block Diagram . . . . . . . . . . . . . 101 4.2.2 The Receiver Amplifier 105 4.2.3 The Data Demodulator and Despreader 105 4.2.4 The Data Detector . . . 108 5. MODEM PERFORMANCE 116 5.1 Test Procedure 116 5.2 Performance i n a Controlled Environment . . . . . 117 5.3 Performance i n a Large Multi-Use Building 123 5.4 Performance i n a Small R e s i d e n t i a l Apartment Complex . . . . 126 5.5 Summary of Results 126 6. CONCLUSIONS 131 6.1 Summary 131 6.2 Cost Estimate of the Modem . . 132 6.3 Suggestions f o r Future Work 133 REFERENCES 134 i v LIST OF TABLES Table Page 1 . 1 I n t r a b u l l d i n g communications s e r v i c e s . . . . . . . . . . . . . . 5 1 . 2 Data rates required to implement some PLC s e r v i c e s . 7 3 . 1 Modem c h a r a c t e r i s t i c s . . . . . . . . . . . . . . . o . . . . . . 7 0 4 . 1 Transfer function of the coupling network 9 8 5 . 1 Measured performance i n a white noise environment. . . . . . . . 1 2 3 LIST OF ILLUSTRATIONS Figure Page 1.1 A monthly meter reading message . . . o . . . . . . . 3 2.1 The high voltage network i n British Columbia . . . . . . . . . 13 2.2 The ele c t r i c a l equivalent of a high voltage line . . . . . . . 14 2.3 The distribution 2.4 The el e c t r i c a l equivalent of a distribution line . 17 2.5 Residential wiring between houses and the distribution 18 transformer 2.6 Typical residential wiring . . . . . . . . . . . . . . . . . . 19 2.7 Transient recorded during lightning storm . . . . . 21 2.8 Transient recorded during starting of a furnace blower . . . . 21 2.9 Transient power line noise 22 2.10 Narrowband power line noise 23 2.11 Power line conducted noise . . . . . . 25 2.12 Power line impedance at low frequencies . . . . . . . . . . . . 28 2.13 The measured power line impedance 30 2.14 Typical transmission versus frequency characteristic . . . . . 31 2.15 Characteristics of a channel loaded with some specific 33 devices . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.16 Narrowband channel drop-outs due to standing waves . . . . . . 34 2.17 A VLF transmitter and the resulting distribution line voltage . 37 2.18 A VLF receiver . . . . . . . . . . . . . . . o . . . . « > « . 37 2.19 A ripple control transmitter . . . . . . . . . . . 38 2.20 A ripple control receiver . . . . . . . . . . o . . . . . . . . 38 v i Figure Page 2.21 A baseband transmitter and receiver . . . . . . 40 2.22 A direct sequence spread spectrum communications system . . . . 44 2.23 D S - S S PSK time domain waveforms . . . . . . . . . . . . . . . . 45 2.24 DS-SS PSK frequency domain power spectrums . . . . 46 3.1 Transmitter and receiver block diagram for the DS-SS PLC modem. 49 3.2 Autocorrelation of an m-sequence of N bits 51 3.3 Code synchronization by zero crossings 53 3.4 A coherent, line synchronized spread spectrum system . . . . . 55 3.5 The six possible power line voltages . . . . . . . . . . . . . 58 3.6 Zero crossing j i t t e r due to noise on the 60 Hz voltage . . . . 60 3.7 Instantaneous frequency variation i n e l e c t r i c i t y generators . . 61 3.8 The effect of unequal loads on zero crossing j i t t e r . . . . . . 62 3.9 Measured zero crossing j i t t e r . . . . . . . . . . . . . . . . . 64 3.10 Maximum processing gain 65 3.11 Effective processing gain . . . . . . . . . . 67 3.12 Spectral occupancy of the spread spectrum signal . . . . . . . 71 3.13 Polarity ambiguity In modem connection . . . . . . . 73 3.14 Impedances seen by a PLC transmitter . . . . . . . . . . . . . 74 3.15 Equivalent impedances seen by a PLC transmitter . . . . . . . . 75 3.16 The coupling network . . . . . . . . . . . . . . . . . . . . . . 77 3.17 Ideal autocorrelation function of the spreading code . . . . . 79 3.18 Cross-correlation between two m-sequences . . .. ... . . . . . . . 79 4.1 The transmitter block diagram 82 4.2 The power supply . . . . . . . . . . . . . . . . . . . . . . 83 vii Figure Page 4.3 The code generator block diagram 85 4.4 The zero crossing detector 86 4.5 Output of the zero crossing detector 87 4.6 The clock generator schematic 88 4.7 The clock generator timing diagram . . . . . . . 89 4.8 The code generator 91 4.9 Operation of the code generator 92 4.10 The data spreader and BPSK modulator 93 4.11a Frequency spectrum of the spread data . . . . . 95 4.11b Frequency spectrum of the modulated data 95 4.12 The transmit amplifier . 96 4.13 The line coupling network 97 4.14 Transfer function of the coupling network 99 4.15 Data transmission through the coupling network 100 4.16 The spectrum of the transmitted signal 102 4.17 The receiver block diagram 103 4.18 Circuits similar to the PLC transmitter 104 4.19 The receive amplifer 106 4.20 Block diagram ofthe data demodulator and despreader 107 4.21 A simplified data demodulator and despreader 109 4.22 Schematic of the simplified data demodulator and despreader . . 110 4.23 The data demodulator and despreader I l l 4.24 Expanded view of the data demodulator and despreader . . . . . 112 v i i i Figure Page 4.26 The data detector schematic 114 4.27 Performance of the data detector 115 5.1 The white noise tests . . . . . . . . . . . . . . . . . . . . . 118 5.2 The receiver's noise bandwidth . . . . . . . . . . . 121 5.3 Predicted and measured performance i n a white noise environment 122 5.4 The equipment used for tests In a large multi-use building . . 124 5.5 Performance in a large multi-use building 125 5.6 Performance i n a small apartment complex . . . . . . 127 5.7 Effect of the data despreader on impulse noise . 128 ix ACKNOWLEDGEMENTS I would l i k e to thank Dr. R.W. Donaldson f o r h i s suggestions and encouragement during t h i s p r o j e c t . In addition I would l i k e to thank Dean L.M. Wedepohl for his i n t e r e s t and advice and Dr. C.S.K. Leung f o r the use of equipment which enabled me to carry out the modem t e s t s . F i n a n c i a l support during t h i s work was provided by the Natural Science and Engineering Research Council. x 1 1. INTRODUCTION TO POWER LINE CARRIER 1.1 Uses of Power Line C a r r i e r (PLC) Power l i n e c a r r i e r involves transmission of data using l i n e s whose primary purpose i s the transmission or d i s t r i b u t i o n of e l e c t r i c power. 1.1.1 Uses of High Voltage Lines High voltage (HV) l i n e s have been used to carry information as well as power f o r some time. In some cases the l i n e s carry long distance t e l e -communication services such as voice or low b i t rate data channels. Using the HV l i n e s f o r communications i s p a r t i c u l a r l y a t t r a c t i v e f o r developing countries which do not have widespread land l i n e s or microwave l i n k s . In t h i s case the desired information bandwidth i s large and almost any number of channels made a v a i l a b l e could be put to use. However, the channel character-i s t i c s (see Section 2.1.4) normally l i m i t the information bandwidth. Another service which i s sometimes implemented using HV l i n e s as a communications channel i s power system telemetry. In t h i s case data on generating plants, remote substations, and the HV l i n e s themselves are sent over the HV l i n e s to a c e n t r a l monitoring s t a t i o n . This c e n t r a l s t a t i o n i s then able to monitor the state of the power system. 1.1.2 Uses on D i s t r i b u t i o n Lines The low voltage d i s t r i b u t i o n wiring has also been used f o r communica-t i o n s . PLC on d i s t r i b u t i o n wiring i s used to provide meter reading, 2 load shedding, and other services [1], The motivation for automated meter reading i s to reduce the costs associated with manual reading [2]. In addition, remote meter reading allows p r i c e r a t i o n i n g , thereby c o n t r o l l i n g e l e c t r i c i t y demand during peak periods [3], This c o n t r o l l i n g of e l e c t r i c power reduces the requirements f o r a u x i l i a r y generators and hence costs. S i m i l a r l y , automatic shedding of non-essential loads reduces a u x i l i a r y generator requirements and p o t e n t i a l brown-out problems. Caldwell has done a cost/benefit analysis of these services f o r the London area [4]. His figures Indicate that automatic load shedding and meter reading are economically v i a b l e provided that meter reading c a p a b i l i t i e s are used to provide p r i c e r a t i o n i n g (time of day t a r i f f s ) . Both of these services require the communication of data between the customer's premise and the l o c a l Hydro o f f i c e . Meter reading requires e l e c t r i c power consumption data to be transmitted from the customer to the Hydro o f f i c e , while load shedding requires shedding commands to be transmitted i n the opposite d i r e c t i o n . The data rates required f o r these services can be estimated as f o l l o w s . For meter reading (time of day b i l l i n g ) , assume four b i l l i n g periods per day. A "smart" meter keeps a running t o t a l of the number of kilowatt-hours used during each time-of-day period. Once a month a s i n g l e message i s sent to the Hydro o f f i c e containing the previous month's e l e c t r i c i t y usage. Figure 1.1 shows a possible format f o r t h i s message which contains 124 b i t s and includes a s t a r t f i e l d , an address f i e l d f o r over one m i l l i o n customers, an e r r o r control f i e l d , the e l e c t r i c i t y usage, and a stop f i e l d . 3 NO.OF BITS 20 20 -A-20 20 20 8 8 KILOWATT KILOWATT j KILOWATT KILOWATT ERROR CONTROL | S T A R T ADDRESS HOURS AT HOURS AT HOURS AT HOURS AT STOP RATE 1 RATE 2 RATE 3 RATE 4 124 BITS/MESSAGE Figure 1.1 A monthly meter reading message. Assuming one substation for every 100,000 customers the data rate required for meter reading i s calculated using equation 1.1. R • 10 5 customers x ^ M t S x 1 m ° f h 4 b i t / s e c (1.1) customer-month 2 > 6 x l Q 6 s e c For loading shedding assume each shedding command a f f e c t s ten customers (except s p e c i a l emergency shedding commands which would e f f e c t many hundreds of customers). Also assume that one command (msg) i s needed to turn non-e s s e n t i a l loads o f f , another message to turn them back on, and that t h i s on-o f f cycle occurs an average of twice a day. F i n a l l y , l e t the message be the same length as the message shown i n Figure 1.1. then, the data rate required f o r load shedding i s given by (1.2). •,/%5 4 msg 124 b i t s 1 day / r t . / R = 10 customers x — — x x — « 4 0 b i t s / s e c s 10 customers-day msg _ n5 (1.2) These data rates are estimates only but are close to the data rates used on some commercial PLC communication systems [1]. 1.1.3 Uses on I n t r a b u i l d i n g Lines Table 1.1 l i s t s some of the communication services which could be provided within a home, o f f i c e or factory [ 5 ] . There are a number of communication channels which could be used to provide these services such as CATV cable [6], telephone l i n e s , i n f r a r e d r a d i a t i o n [7] or power l i n e c a r r i e r (PLC). The advantages of using PLC are that the channel i s already In place, it has a standard mechanical i n t e r f a c e i n the wall socket and power plug, and 5 Security Monitoring [8] Intercom [9] Stereo d i s t r i b u t i o n [10] Video d i s t r i b u t i o n [11] Computer to Terminal Links [12] Extension Phones [13] Appliance Control [14] Environmental Management [15] Table 1.1 Intrabuilding Communications Services 6 i t runs almost everywhere Inside and outside homes, o f f i c e s , and f a c t o r i e s . Its disadvantages are that i t has l i m i t e d bandwidth compared with f i b r e optics or CATV cable, i t i s noisy, and i t s impedance, attenuation, and topology are not well defined. (See Section 2.1). 1.2 PLC Modem Requirements Modem operational parameters of i n t e r e s t include data rate and e r r o r r a t e s . Another f a c t o r which impacts modem design i s the switching scheme used. These matters are discussed i n t h i s s e c t i o n . 1.2.1 Data Rates Required f o r PLC The requirements of a PLC system w i l l vary depending on the type of services c a r r i e d . Data rates of a few b i t s / s e c would s a t i s f y the require-ments of meter reading while d i g i t a l voice would require over 10 Kbits/sec. Table 1.2 l i s t s the data rates of some PLC services which could be provided. Notice that many of the services require fewer than 60 b i t s / s e c , a few require rates near l K b i t / s e c , while few others require data rates i n excess of 100 Kbit/sec. Table 1.2 l i s t s some services only. Many others e x i s t and even more would be proposed i n the future. It seems then that any PLC system must be f l e x i b l e enough to handle a wide v a r i e t y of information r a t e s . Some of the data rates l i s t e d i n Table 1.2 need c l a r i f i c a t i o n . D i g i t a l voice can be encoded i n a v a r i e t y of ways; companded PCM requires 64 K b i t s / -sec [15] while adaptive delta-modulation can provide adequate q u a l i t y using 16 Kbits/sec [16]. Some of the phoneme-based speech synthesizers Meter reading Pwr systems telemetry Load shedding Voice ( d i g i t a l ) Voice (analog) Bldg s e c u r i t y Bldg intercom Stereo d i s t r i b u t i o n Terminal-to-computer l i n k Computer-to-terminal l i n k Computer-to-computer l i n k Video d i s t r i b u t i o n Appliance co n t r o l Environmental management ~ 4 b i t s / s e c (See Section 1.1.2) ~ 10 b i t s / s e c (See Section 1.1.1) ~ 40 b i t s / s e c (See Section 1.1.3) ~ 10 Kbits/sec ~ 4 KHz ~ 10 b i t s / s e c ~ 4 KHz or 10 Kbits/sec (per channel) ~ 40 KHz or 1 Mbit/sec ~ 50 b i t s / s e c ~ 100 b i t s / s e c ~ 300 b i t s / s e c to 64 Kbits/sec ~ 6 MHz or 80 Mbits/sec ~ 10 b i t s / s e c ~ 10 b i t s / s e c fable 1.2 Data rates required to Implement some PLC s e r v i c e s . 8 require approximately 1 Kbit/sec [17], [18], Hence, the 10 Kbits/sec quoted f o r voice and intercom i s a compromise. Analog voice as c u r r e n t l y provided on both telephone and radio voice channels uses approximately 4 KHz of bandwidth. Services such as b u i l d i n g s e c u r i t y , appliance c o n t r o l , and environmental management can be implemented i n a v a r i e t y of ways. T y p i c a l l y these services might be implemented as f o l l o w s . A maximum of 1024=2 1 0 devices might be s p e c i f i e d which could be used as sensors f o r s e c u r i t y and environmental monitors or c o n t r o l l e d switches f o r appliances. The message format would contain 10 b i t s of address, 8 b i t s f o r command or status information, an error control f i e l d of eight b i t s , and 4 b i t s f o r s t a r t , stop, and overhead. A t y p i c a l message would therefore contain around 30 b i t s . Some of the devices such as temperature sensors or appliance switches would not be accessed more than a few times a day. Other devices such a se c u r i t y sensor may be polled every few seconds or may Independently send alarm messages. Depending on the number of a c t i v e devices we could expect a few messages per hour i n a l i g h t l y loaded s i n g l e family home to over ten messages per second In a f a c t o r y with many sensors* The f i g u r e of 10 b i t s / s e c given In Table 1.2 i s a compromise between these extremes. Stereo d i s t r i b u t i o n requires an analog bandwidth of 20 kHz per channel. D i g i t a l stereo with high f i d e l i t y must be sampled at 40 kHz, with 12 bits/sample, f o r a t o t a l of about 800 Kbits/sec on two channels. Data communications between computers and terminals i s highly v a r i a b l e . A s k i l l e d operator can type approximately 10 characters per second (120 9 words/min). Using 8 bits/character gives a peak data rate from terminal to computer of 80 b i t s / s e c . However, t h i s rate i s r a r e l y sustained, except possibly i n word processing. Thus, 50 b i t s / s e c seems to be a safe estimate. Computer to terminal information rates are also highly v a r i a b l e . Individuals can read at about 250 words/min (» 150 bits/ s e c ) but r a r e l y sustain that rate during computer transactions. The rate of 100 b i t s / s e c represents about 180 words/min. Computer to computer communication can be at almost any r a t e . 300 b i t s / s e c used to be standard over telephone l i n e s but recently 1200 b i t s / s e c or higher has become commonplace. Some s p e c i a l i z e d data communication services such as packet switched networks o f f e r data rates of 64 Kbits/sec. F i n a l l y video d i s t r i b u t i o n requires 6 MHz of bandwidth using the NTSC standard while d i g i t a l video can range from 1 Mbit/sec f o r picturephone [19] to 80 Mbits/sec f o r 8 b i t PCM video [20], 1.2.2 Error Rates Tolerable f o r PLC The error rates t o l e r a b l e f o r the services l i s t e d i n table 1.1 are d i f f i c u l t to estimate. Customers using meter reading and load shedding equipment seem s a t i s f i e d with a b i t error rate of 10 - 1* [21]. Modem manufacturers t r y to d e l i v e r a b i t error rate of better than 1 0 - 6 while users of d i g i t a l voice t o l e r a t e an error rate of 10~ 2 [22]. A PLC modem might provide the user with the c a p a b i l i t y to trade o f f e r r o r rates against data rates [23]. One way to accomplish t h i s objective i s to make the code rate v a r i a b l e , which involves a scheme to co n t r o l undetected 10 errors by trading off the number of information bits against the number of error control bits [23]. The type of errors (bursty, random, etc) must be known i f the most effective error control strategy is to be selected [23], Unfortunately, there is not much data available on PLC error s t a t i s t i c s . 1.2.3 Switching Alternatives for PLC The switching requirements for a PLC communications system are varied. Some services, such as computer to terminal communications, are effectively handled by packet switching. Other services which include f i l e transfers are effic i e n t l y handled using a ci r c u i t switched network. Sometimes point-to-point communications is required, such as when a message from an individual customer's hydro meter i s sent to the b i l l i n g o f f i c e . In other cases broadcast capability i s required, for example when a load shedding command is to be sent to several hundred customers. Finally, some analog services, such as home intercom, may use the PLC channel. It appears that a multipurpose PLC modem should allow for different switching alternatives. 1.3 Outline of the Thesis Chapter two Is a discussion of previous work which Impacts spread spectrum modem design. The communications characteristics of the power system are examined in Section 2.1. In Section 2.2 PLC signalling techniques which have already been tried by others are summarized. Section 2.3 is an Introduction tb spread spectrum communications. 11 Chapter three deals with the system design of the spread spectrum power l i n e modem. Sixty Hertz synchronization and i t s implications are outlined i n Section 3.1. Section 3.2 explains the transmitter and receiver structures required to implement a d i r e c t sequence spread spectrum PSK modem. The choice of the bandwidth selected for communication i s explained i n Section 3.3. The network which coupled the modem in t o the power system i s discussed i n Section 3.4 while Section 3.5 discusses the code generator. Chapter four documents the modem implementation. Included are block diagrams, c i r c u i t schematics, and c i r c u i t d e s c r i p t i o n s . Chapter f i v e contains the r e s u l t s of tests c a r r i e d out with the modem. In Section 5.1 background information on modem te s t i n g i s given. Section 5.2 describes the re s u l t s of tests done i n a co n t r o l l e d environment. Section 5.3 and Section 5.4 are the r e s u l t s of tests done on some representative power c i r c u i t s . A summary and discussion of the r e s u l t s Is included i n Section 5.5. Conclusions and suggestions f o r future work are given i n Chapter s i x . 12 2. REVIEW OF PREVIOUS WORK 2.1 Power System Transmission C h a r a c t e r i s t i c s The transmission c h a r a c t e r i s t i c s of power transmission and d i s t r i b u t i o n systems have been, and continue to be, the subject of extensive study. Items of i n t e r e s t include the propagation of power system t r a n s i e n t s , the propagation of conducted power l i n e noise, and the propagation of power l i n e c a r r i e r signals within the power system. The f a c t o r s which influence the transmission c h a r a c t e r i s t i c s of the power system are discussed i n t h i s s e c t i o n . 2.1.1 Power systems topology E l e c t r i c a l power i s d i s t r i b u t e d from generating plants to users over a s e r i e s of d i s t i n c t networks. These networks operate at d i f f e r e n t voltages and are interconnected by transformers. F i r s t the high voltage network c a r r i e s power from generating plants to to e l e c t r i c a l substations. Figure 2.1 i s a map, published by B.C. Hydro, of the high voltage l i n e s e x i s t i n g or planned i n B.C. as of 1974. These l i n e s vary g r e a t l y i n length from a few to hundreds of kilometres. Figure 2.2 i s the e l e c t r i c a l equivalent of one high voltage l i n e showing i t s generator, branches, loads (which vary with time and frequency), and the l i n e i t s e l f . Next, the d i s t r i b u t i o n network d e l i v e r s e l e c t r i c power from a substa-t i o n to a number of d i s t r i b u t i o n transformers. Figure 2.3 i s a map of part of t h i s d i s t r i b u t i o n network for Vancouver. These d i s t r i b u t i o n l i n e s vary i n length, too. In a d d i t i o n they have a large number of branches and loads, presented by the d i s t r i b u t i o n transformers, with complex Impedances. These loads, l i k e the l i n e s , have impedances which vary with frequency. The load 13 Figure 2.1 The high voltage network i n B r i t i s h Columbia. PORTAGE MOUNTAIN GENERATING STATION AND SWITCHYARD 14 GENERATORS 227 MW 0.93 PF GENERATOR BREAKERS 2000 MVA 10.000 AMP. GENERATOR TRANSFORMERS 240MVA (3 X80MVA) 900/138/12 KV TRANSFORMERS 230 MVA 900KV CIRCUIT BREAKERS 33.000 MVA 2300AMR 2-I23MVAR REACTORS n 1 i i 1 n J J 3 0 0 KV BUS J i i s . is < O K • .—«->l><>'. I2KV-7SMVAR REACTOR 300/230/12 KV TRANSFORMERS 300 MVA SYNCHRONOUS CONDENSER G 73 MVAR m TO BULKLEY ANO. SKEENA VALLEYS 230KV RING BUS Dp SERIES COMPENSATION -CK, ,^Q^J -2-123 MVAR REACTORS o <" ° K 132 KV BUS o PRINCE GEORGE SUBSTATION J t I2KV-75MVAR REACTOR 230KV-IOOMVAR REACTOR 900/230/12 KV TRANSFORMERS 900 MVA 230KV-IOOMVAR REACTOR SYNCHRONOUS CONOENSER 73 MVAR 8 a 3-123 MVAR REACTORS SERIES COMPENSATION •-9 230KV RING BUS £ £ 2-I23MVAR REACTORS A J 1 j 88 t KELLY LAKE SUBSTATION 2-123MVAR REACTORS SERIES COMPENSATION MERIDIAN SUBSTATION K>0MVAR REACTOR tOOMVAR CAPACITOR faff 230KV BUS 300/ 230/12 KV TRANSFORMERS 600MVA 2-73 MVA REACTORS 1 T 230KV B U S 1 I i BONNEVILLE •TO POWER AOMMISTRATKW INGLEOOW  SUBSTATION 3-73 MVAR REACTORS EXISTING B C HYDRO SYSTEM EXISTING SC.HYDRO SYSTEM Figure 2.2 The electrical equivalent of a high voltage line [24]. 15 p i t a s u e : o-- l ^ Q — "o -i — Q Q — — v — FrM] D Vw*. i i—s^n IE.7+ C3N .i.l/'.U.t.lOtS .... s ID D VUH3M SU». DC. I T M ™ i c DDE 1 "™|-J . (y .^V**—(T,ao- »•*•) . . . ^ 3D 0 c rz \~- it ******* rwrt " 12.82 CSMK M O O esoo z e o o MOO 3000 H 5 31 oo 3 t O O s s o o 3 4 0 0 3 6 0 0 3 6 0 0 H » C N O O t»A»«»C»MC» g) T I U V O I W I ktult » ». or ?cu*',ro»«t«« • w Ig O l S T f t t B u T O * U H0C4QNOUH0 C > K . I LATEST ISSUE' B.C. HYDRO a POWER AUTHORITY CITY OF VANCOUVER MAP 3 12 Kv P R I M A R Y D I S T R I B U T I O N a » . « » q j C gy- i i 50. * SC* w C 400' I Figure 2.3 The d i s t r i b u t i o n network. 16 impedances, however, also change with time as e l e c t r i c a l loads (such as home appliances) are switched on and o f f . In some cases i n d i v i d u a l d i s t r i b u t i o n l i n e s connect to each other forming a loop back to the d i s t r i b u t i o n transformer. (These loops provide redundancy. If a s i n g l e f a u l t occurs at any point i n the loop users w i l l s t i l l receive power.) Figure 2.4 i s the e l e c t r i c a l equivalent of part of the d i s t r i b u t i o n network. Resi d e n t i a l wiring i s s i m i l a r to the d i s t r i b u t i o n network [25]. Each d i s t r i b u t i o n transformer supplies power f o r a number of houses. Figure 2.5 shows a t y p i c a l connection from a d i s t r i b u t i o n transformer to the houses. Although a single-phase transformer i s shown, three-phase transformers are also used. In turn, each house contains many Individual loads. Figure 2.6 shows the wiring i n a t y p i c a l house. Notice that each household load may connect to e i t h e r or both phases. A l s o , there are few r e s t r i c t i o n s on the actual number of loads which may be connected at a given time In a given household. (There are some r e s t r i c t i o n s - each branch must contain no more than twelve d i f f e r e n t loads. In a d d i t i o n , some c i r c u i t s , including those which supply appliances, are f u r t h e r r e s t r i c t e d as to loading [26].) O f f i c e and i n d u s t r i a l wiring resemble r e s i d e n t i a l wiring i n that a v a r i e t y of loads can be connected to an i n d i v i d u a l c i r c u i t although three-phase power and d i f f e r e n t voltages are sometimes used. A wide v a r i e t y of loads can be connected to the power l i n e . On the high voltage and d i s t r i b u t i o n l i n e s these loads are usually r e s t r i c t e d to transformers, which can present v i r t u a l l y any impedance, and power f a c t o r Figure 2.4 The electrical equivalent of a distribution line 3 0 DISTRIBUTION LINE FEEDER j?B 0C 10 DISTRIBUTION TRANSFORMER 10 SECONDARY • « • • • e <H * 8 * • n T Figure 2.5 Res i d e n t i a l wiring between houses and the d i s t r i b u t i o n transformer. 10 SECONDARY 240 v 12Qv I120v| DRYER CIRCUIT 0,-1 VPLU6 H PLUG •0-CIRCUIT 0t -2 HLIGHT t r l PLUG ri LIGHT •4 LIGHT 7_ PLUG h LIGHT 7 CIRCUIT 02-1 STOVE CIRCUIT 02-2 PLUG -j| LIGHTL JPLUG | - | • -8 ' CIRCUIT 0.-M CIRCUIT 02-N Figure 2.6 Typical residential wiring. 20 correction loads. Residential and commercial wiring, however, can supply loads which are time-varying i n number and type. These loads may be induc-tive, capacitive, resistive, or nonlinear. They may draw currents ranging from milliamps to hundreds of amps. Finally, except for regulations covering new equipment [27 ], they can conduct and/or radiate any amount of e l e c t r i c a l noise back on to the power l i n e . 2.1.2 Power line noise Power lines carry el e c t r i c a l noise along with the 60 Hz power. This noise is generated from a large number of sources in a large number of ways. Power line noise can be short lived. This transient noise can arise from events such as lightning strikes (Figure 2.7) or powering on an induc-tive load (Figure 2.8). Power line noise can be transient i n the short term and periodic in the long term. This type of noise is typically generated phase controlled loads (see Figure 2.9). Power line noise can be coupled In externally from radio and T.V. stations. Power line noise can be narrowband (Figure 2.10). This type of noise i s often produced by d i g i t a l equipment where many internal devices w i l l turn on and off (producing transient noise) i n response to some master clock [28]. Finally, power line noise can be wideband. This type of noise can arise from spark gaps or corona discharges [29]. Although i t i s instructive to look at sources and characteristics of power line noise, i t i s the noise seen by a receiver which affects the per-formance of PLC systems. The transmitted and received noise are not the 21 WOO Us ' ' . • H i m 2000 m M H i m 111111 -VOLTS 0 1 2000 m im i i WOO m •amuna • B ||ffp HI HI ' • v : - • BB 0 5 10 20 30 10 75 100 MICROSECONDS Figure 2.7 Transient recorded during l i g h t n i n g storm. 150 100 50 VOLTS 0 50 100 150 0 8.5 18 32 53 10 MICROSECONDS Figure 2.8 Transient recorded during s t a r t i n g of a furnace blower. Figure 2.9 Transient power l i n e noise. Horizontal * 10 KHz/cm V e r t i c a l • 10 dB/cm Figure 2.10 Narrowband power l i n e noise. 24 same. The difference arises from the transmission characteristics of the power system. Smith [30] has done a survey of power line noise coupled into a 50ft receiver. His results are shown i n Figure 2.11 for various environments. These results show average levels only, although short term variations of 10 dB were observed. Notice how the noise levels vary with the environment. The rural levels are almost 30 dB below the urban office levels! Notice also that the noise is not white but f a l l s off at about 20 dB/dec. The r o l l - o f f results from both lower generated noise and higher attenuation at high f r e -quencies (see Section 2.1.4). Assuming most loads generate noise, we would expect that power line noise would decrease during periods of inactivity (typically at night). Similarly, periods of heavy ele c t r i c a l use (around dinner time) should also have increased noise. In other words, background noise should correlate well with the electrical load pr o f i l e . Some power line noise measurements were recorded, over a 24-hour period, in Room 325 of the Hector MacLeod building, U.B.C. Also, power line noise, again over a 24-hour period, was measured i n Suite 206 - 3680 West 7th Avenue, Vancouver, B.C. (A residential complex). The background noise i n the Hector MacLeod building was f a i r l y consistent varying less than 2 dB over the measurement period. However, the background noise in the residential complex varied by more than 6 dB during a 24-hour period. The highest noise levels occurred at night around 11:00 p.m. while the lowest levels were present i n the early morning. 25 Figure 2.11 Power l i n e conducted noise. 26 In addition to background noise l e v e l s , e l e c t r i c a l interference from common household appliances were also measured. The measurements i n d i c a t e that most loads do not s i g n i f i c a n t l y add to the background noise. The worst noise offenders were the vacuum cleaner and the l i g h t dimmer and these added less than 3 dB to the background noise. F i n a l l y , very l i t t l e appliance noise couples across from one phase to the other. 2.1.3 Signal propagation at low frequencies Power l i n e s contain many impedance d i s c o n t i n u i t i e s which cause s i g n a l s propagating down the l i n e to r e f l e c t and backscatter [31]. The backscattered signals w i l l combine with the forward s i g n a l to create areas of increased and decreased s i g n a l strength. In other words standing waves w i l l be set up along the power l i n e . These transmission l i n e e f f e c t s become pronounced when the impedance d i s c o n t i n u i t i e s are separated by more than approximately an eighth of the signal's wavelength. Therefore, the frequency at which transmission l i n e e f f e c t s become noticeable w i l l depend on the environment through which the s i g n a l passes. For example, i f a propagation v e l o c i t y of free space i s assumed then a signal's wavelength i s given by X » c/f (2.1) where X i s the s i g n a l ' s wavelength, C i s the speed of l i g h t , and f i s the signal's frequency. In r e s i d e n t i a l wiring i n d i v i d u a l c i r c u i t s r a r e l y run longer than three hundred metres, so from (2.1), PLC signals below 100 KHz w i l l not s u f f e r from transmission l i n e e f f e c t s . On the d i s t r i b u t i o n network c i r c u i t s can run tens 27 of kilometres so signals above 10 kHz w i l l be a f f e c t e d . F i n a l l y , on the high voltage power transmission network, power l i n e s can run f o r hundreds of k i l o -metres. Transmission l i n e e f f e c t s on l i n e s t h i s long occur even f o r the 60 Hz power s i g n a l . Ignoring transmission l i n e e f f e c t s permits a simple model f o r PLC s i g -nal propagation at low frequencies. At low frequencies power l i n e conductors have a very small s e r i e s impedance. For example common household wiring (number twelve gauge) has an equivalent ser i e s resistance of .006 Si/m and a seri e s inductance of approximately .43 uH/m depending on the material i n close proximity to the conductors [32]. At a frequency of 100 kHz, and a length of 30 metres, the ser i e s impedance of the conductors i s given by: |ZC0NDUCT0R| = / R g 2 + X g 2 = / (.006x30) 2 + (.43xl0 - 6x30x2Trxl0 5) 2 « i n (2.2) This impedance i s l e s s than one-tenth of the minimum load impedance which can be connected across a r e s i d e n t i a l c i r c u i t . Thus the conductor impedances (Z^,) are small compared to the load impedances (Z ) connected between them. v> Li This means that at low frequencies the equivalent load impedance presented to a PLC transmitter or r e c e i v e r , i s approximately equal to the p a r a l l e l com-bination of a l l the loads on the l i n e (see Figure 2.12). The actual power delivered to the receiver from the transmitter w i l l depend on t h i s equivalent l i n e impedance (Z^) • (At low frequencies i t does not matter where along the l i n e the transmitter and receiver are placed. The power delivered to the receiver i s independent of p o s i t i o n . ) The equivalent l i n e impedance w i l l be time varying, as loads are turned on and o f f , and 28 PLC TRANSMITTER POWER LINE AND LOADS PLC RECEIVER .A . h — 3 B N N B - B - * — • • B N Figure 2.12(a) Power l i n e impedance at low frequencies, POWER LINE PLC TRANSMITTER AND LOADS PLC REGEIVER-L L, L 2 L N Figure 2.12(b) Equivalent power l i n e impedance at low frequencies, 29 frequency varying, depending on the type of loads connected at any given instant. Some measurements of the line impedance were made at different times i n the Hector MacLeod building, U.B.C. The results of these measurements are shown in Figure 2.13. Notice that the line impedance varies with both time and frequency. At frequencies above 100 kHz there is increased variation i n Z^ because transmission line effects become noticeable. At one point Z^ dropped off sharply at 400 kHz. This decreased impedance was due to reflec-tions which occurred at that instant of time and frequency. (Reflections, or standing waves, w i l l cause the line impedance to vary with position along the line [33].) The attenuation of the PLC signal between the transmitter and receiver depends on the equivalent line impedance. Because this Impedance varies with time and frequency the attenuation w i l l also vary. Figure 2.14 shows the attenuation between a transmitter and receiver measured at different times within a building. 2.1.4 Propagation at high frequencies With increased frequency the skin effect increases the per unit r e s i s -tance and inductance of the line causing i t s characteristic impedance and i t s attenuation to increase [34]. The characteristic impedance of a transmission line depends on many factors, such as conductor size, spacing, height above ground, diel e c t r i c , etc. For power lines the characteristic Impedance can vary from 400 ft in widely separated overhead lines to 20 SI for underground lines [35]. IZL! tfl) 800 700 600 j— 500 H 400 300 200 30 # - X " X X c X c & X o o o o o o o o o Oo OOO AUG 24/81 1:00 PM AUG 24/81 2:30 PM AUG 24/81 4:00 PM AUG 25/81 8:30 AM AUG 25/81 10:30 AM 1 1 1 I I 1111 1 1 II1IMI .01 .02 .03 .04 .O5J06D7JC8O3.1 3 .4 5.6 7A.91. FREQUENCY (MHz) Figure 2.13 The measured power l i n e impedance. -4 » » 1 IOH<> •o no wo sco Figure 2.14 Typical transmission versus frequency characteristic [38] 32 When transmission line effects are included, the transmission charac-teristics of the power system become very d i f f i c u l t to predict because of branches in the line and the time and frequency varying loads. Work has been done on modelling the high voltage and distribution networks at high f r e -quencies [35], [36]. However, even though some general RF characteristics of the power system are known there are s t i l l many uncertanties [37]. The Impedance seen by power line transmitters and receivers w i l l vary with time and position as well as frequency. In general, however, the trans-mission line w i l l transform "distant" load Impedances to the characteristic line impedance and loads in close proximity w i l l therefore have the biggest impact [34]. Attenuation along the power line i s generally quite low because the conductors are large. On overhead lines the attenuation can be less than .1 dB/Km [35]. However, wideband signal attenuation due to capacitive loads [38] and distribution transformers [39] w i l l be present (see Figure 2.15). Finally, narrowband signal fades can occur because of standing waves and multimodal propagation [35]. These narrowband fades w i l l be position and time dependent (see Figure 2.16). 2.1.5 Received signal to noise variations The performance of any communication system is dependent on the signal to noise ratio (SNR) available at the receiver. In power line communication systems the channel characteristics discussed In this section make the received SNR highly variable and unpredictable. M i I — • • • » • • I (Uti) 80 . no so aoo Figure 2.15 Channel c h a r a c t e r i s t i c s of a channel loaded with some s p e c i f i c devices. I I I I •» » bH/> (0 as ao 300 Figure 2.16 Narrowband channel drop-outs due to standing waves. 34 The SNR w i l l vary with time as loads are connected and disconnected. More loads generally r e s u l t In more attenuation and more noise. Thus bet t e r performance would be expected at night when fewer loads are connected across the l i n e . The SNR w i l l vary with frequency. The frequency dependence a r i s e s both from loads and the l i n e i t s e l f . The l i n e , because of the skin e f f e c t , a t t e n -uates high frequencies more than low frequencies. D i f f e r e n t loads w i l l a f f e c t PLC signals d i f f e r e n t l y . Resistive loads, such, as heating elements, w i l l attenuate signals evenly. Inductive loads, such as motors, w i l l atten-uate low frequencies more than higher ones. Capacitive loads, such as power fa c t o r c o r r e c t i o n capacitors, w i l l attenuate high frequencies more than lower ones. The SNR w i l l vary with p o s i t i o n ; t h i s p o s i t i o n dependence i s due to a number of f a c t o r s . F i r s t , narrowband fades due to r e f l e c t i o n s and multimodal c a n c e l l a t i o n are p o s i t i o n dependent. Second, s i g n a l losses increase as the receiver Is moved fa r t h e r from the transmitter. Third, background noise w i l l be worse close to a noisy load. In summary, the s i g n a l l e v e l at a receiver depends on the following f a c t o r s : 1) how much transmitter power i s coupled i n t o the l i n e (the degree of coupling depends on the transmitter's source impedance and the i n s t a n -taneous l i n e impedance presented to the transmitter); 2) how much power i s coupled into the receiver ( t h i s coupling again depends on the receiver's input impedance and the l i n e impedance at the r e c e i v e r ) ; and 3) how much sig n a l power i s attenuated between the transmitter and receiver which i s influenced by the l i n e and i t s loads. The noise l e v e l , and power spectrum, 35 at the receiver w i l l depend mostly on the type of loads connected nearby. Generally, the noise i s approximately +10 dBm, in t o a 5 ohm load at 10 kHz, and f a l l s o f f at approximately 20 dB per decade. 2.2 PLC S i g n a l l i n g Techniques A large number of s i g n a l l i n g techniques f o r power l i n e c a r r i e r have been tested [40]. Information has been sent at baseband, at very low f r e -quencies, at audio frequencies, and at frequencies above 100 kHz. The v a r i e t y of e x i s t i n g systems indicates that there i s not yet a consensus on the best system f o r power l i n e c a r r i e r s i g n a l l i n g . This section discusses the most common power l i n e c a r r i e r systems together with t h e i r advantages and disadvantages. 2.2.1 Very low frequency s i g n a l l i n g Very low frequency (VLF) s i g n a l l i n g i s a data communications system where s i g n a l l i n g i s done at a very low rate, u s u a l l y a f r a c t i o n of the power-l i n e frequency. VLF s i g n a l l i n g circumvents many of the channel anomalies mentioned i n Section 2.1 by ensuring that no high frequency signals are generated. Thus, transmission l i n e and t o p o l o g i c a l e f f e c t s can be ignored and the power d i s t r i b u t i o n g r i d appears as a broadcast channel with n e g l i -g i b l e transmission l i n e l o s s e s . With VLF s i g n a l l i n g , only varying channel noise and load impedances need be considered. Forrest and Gray propose a VLF s i g n a l l i n g scheme i n which every f o u r t h zero crossing of the d i s t r i b u t i o n voltage waveform Is e i t h e r d i s t o r t e d ( i . e . transmitting a one) or undistorted (transmitting a zero) [41]. This d i s t o r -9 36 t i o n i s accomplished by purposely connecting a load at the appropriate zero crossings, or by purposely generating a switching transient (see Figure 2.17). Receivers connected to the l i n e monitor the l i n e zero crossings f o r these switching transients (see Figure 2.18) and assemble messages. VLF s i g n a l l i n g , however, does have drawbacks. F i r s t , only very low data rates can be transmitted (le s s than 100 b i t s / s e c ) . Generally, f o r meter reading, t h i s data rate r e s t r i c t i o n i s not unmanageable since meter data rates are on the order of a few hundred b i t s per month per household. Thus, tens of thousands of households could be accommodated. A second d i f f i c u l t y with VLF s i g n a l l i n g i s the uplink channel, or the channel from the customer to the hydro o f f i c e . The technique used by Forrest only provides a simplex l i n k from the o f f i c e to the customer. Thus, VLF s i g n a l l i n g i s applicable only to load shedding and other s i g n a l l i n g tech-niques must be employed to enable two-way communications. 2.2.2 Ripple c o n t r o l Ripple c o n t r o l has been used f o r power l i n e c a r r i e r since 1929 [42]. Since that time r i p p l e c o n t r o l has evolved Into a number of r e l a t e d systems. Ripple c o n t r o l i s b a s i c a l l y ASK s i g n a l l i n g with a c a r r i e r frequency between 110 Hz and 750 Hz. These frequencies are low enough so that transmission l i n e e f f e c t s can be s a f e l y ignored. Figure 2.19 shows a r i p p l e control transmitter. The data to be trans-mitted controls the f i r i n g of the SCR's i n the s t a t i c frequency converter. The audio frequency used i s chosen to l i e between two of the 60 Hz harmonics to minimize noise. The s e r i e s resonant c i r c u i t (CT and L„) blocks the 60 Hz 37 tXSTRlSUTION TRANSFORMER POINT OF COMMON Figure 2.17 A VLF transmitter and the r e s u l t i n g d i s t r i b u t i o n l i n e v oltage. P O W E R U N E =1> ZERO CROSSING RDATA MESSAGE ASSEMBLER AND DETECTOR INTERPRETER Figure 2.18 A VLF r e c e i v e r . 38 j~i_n_ TDATA STATIC FREQUENCY CONVERTER —TJIP-JUU1— FROM ^SUBSTATION TRANSFORMER TO DISTRIBUTION NETWORK Figure 2.19 A r i p p l e c o n t r o l transmitter. FROM — r v j -? POWERZJ I Z 3 U N E - L > T _ J f — n f T n — M -V " L P "PCP ENVELOPE DETECTOR "t> RDATA _ J T J T _ —w—w— Figure 2.20 A r i p p l e c o n t r o l r e c e i v e r . 39 while the parallel resonant circuit (Cp> L^) couples the audio power to the li n e . The transformer provides isolation. The parallel resonant c i r c u i t also f i l t e r s the higher audio harmonics generated by the frequency conver-ter. The ripple control receiver is shown i n Figure 2.20. A coupling net-work similar to the transmitter passes the carrier, rejects the 60 Hz and isolates the receiver from the power l i n e . A non-coherent AM detector f o l -lows and passes the received data to the decoding and control c i r c u i t s . The advantages of ripple control are i t s simplicity and Its low carrier frequency. The low carrier frequency means that transmission line effects can be ignored. Ripple control also has disadvantages, including a low data rate (about 2 bits/sec), simplex only transmission, and very high transmitter requirements (greater than 50 RVA!). The high transmitter power is required because the power line noise i s very high at low frequencies. 2.2.3 Baseband signalling Baseband signalling i s used to transmit and receive data i n the v i c i n i -ty of the zero crossings of the 60 Hz power [43]. During this interval both the power voltage and noise are low and the line (at least between trans-formers) looks l i k e a d.c. link. Noise i s low in this Interval because motors and switches draw (and hence generate) l i t t l e power. However, some loads such as switching power supplies and data processing equipment (which operate asynchronously from the 60 Hz) w i l l generate noise in this Interval. Figure 2.21 shows a typical baseband transmitter and receiver. In this system coupling to the line i s easy. A d.c. connection is made to the line TRANSMITTER 40 ^COMPUTER ZERO UART TDATA ZERO CROSSING DETECTOR POWER LINE RECEIVER POWER LINE —! ZERO CROSSING DETECTOR ZERO yu COMPUTERJ 1 ZERO SWITCH UART DATA Figure 2.21 A baseband transmitter and r e c e i v e r . 41 during the zero-crossing interval. Beside simple coupling, the system has other advantages. Two-way communication i s possible and the T^ power requir-ed is low (because the noise i s low). Finally, high data rates can be achieved (more than 10 Kbits/sec). The biggest disadvantage of this system is that the signals propagate poorly through the distribution transformer. For this reason baseband sys-tems normally operate only within a building. 2.2.4 Two-way signalling near 10 KHz This class of signalling techniques attempt to overcome the disadvan-tage of ripple control (see Section 2.2.2). F i r s t , the power line noise i s smaller at 10 KHz than 400 Hz. Second, the power line has a higher imped-ance. These two factors combine to reduce the transmitter power required. A lower transmitter power also makes multiple transmitters feasible with the result that two-way communication is possible. The actual signalling can be ASK [44], PSK [45], or FSK [43], a l l of which have been tried. One disadvantage of these systems i s that they s t i l l require moderate transmitter power, typically 400 watts, which makes the transmitter expen-sive. Another disadvantage i s that the signals are s t i l l attenuated somewhat by the distribution transformers and power factor correction capacitors. 2.2.5 Two-way signalling near 50 KHz At high carrier frequencies the transmitter power required i s around 1 watt [44] and coupling to the power lines can be done simply [9]. These facts make the communication equipment cheaper than most of the lower f r e -42 quency alternatives. However, transmission line effects (see Section 2.1.4) become more pronounced so performance i s less predictable. Even at these frequencies there i s wide dissagreement on which frequen-cies are best suited to power line carrier. Frequencies from 10 KHz to over 100 KHz have been tried [43], [44], and [46]. 2.3 Spread Spectrum Communications Spread spectrum communications is a signalling technique which increases the bandwidth (spreads the spectrum) of a signal to make i t less susceptible to inteference [47] and narrowband fades [48]. The spreading process can be accomplished by one of four techniques or combinations of them. Chirp spread spectrum systems [49] sweep a carrier, usually linearly, within a frequency band (f , f„). A 'one' could be transmitted by sweeping from f to f while a 'zero' could be transmitted by sweeping from f to f . IJ H H II Frequency hopped spread spectrum systems [50] shift the carrier fre-quency to a number of discrete frequencies within a frequency band ( f T , f ) . Ii H The shift from one frequency to another is usually done in a pseudo-random fashion. Time hopped spread spectrum systems [50] turn the carrier on and off in a pseudorandom order. Time and frequency hopped systems are often combined into a hybrid system [51]. 43 2.3.1 Direct sequence spread spectrum communications Figure 2.22 shows a direct sequence spread spectrum (DS-SS) communica-tions system. In DS-SS the information bit-stream d(t) Is f i r s t multiplied by a high rate pseudorandom bit stream c(t) before being PSK modulated by the carrier k ( t ) . Similarly, the receiver, after conventional demodulation, remultiplies by c(t) to recover the information bit stream. To gain more insight into the spreading and despreading processes Figures 2.23 and 2.24 show the various signals in the time and frequency domains. In this example the information rate = 1/T^ bits/sec and the chip rate (i.e., the pseudorandom bit stream rate) = 1/T = l O x l / T . In addition, c I the carrier frequency Is f = 1/Tc» Also included is wideband interference. In this case the processing gain = G p = 10 l o g 1 0 (T^/TC) = 10 dB. Notice that the SNR i s -5 dB at RF but +5 dB at baseband because of the pro-cessing gain. '2.3.2 Performance characteristics of DS-SS Spread spectrum signals are Insensitive to narrowband fades which occur in PLC as a result of reflections and multimode propagation (see section 2.1.4). Spread spectrum communication is not degraded severely by narrowband fades because the RF energy i s spread over a large number of frequencies. Thus a narrowband fade w i l l remove only a small portion of the RF signal so that.the total energy i s not significantly reduced [48]. Spread spectrum signals are insensitive to narrowband interference which can be present in PLC systems (see section 2.1.2) [52], Spread spec-trum signals can be successfully demodulated even with a negative received signal-to-noise ratio. Thus, with a power density below that of the noise 44 Figure 2.22 A d i r e c t sequence spread spectrum communications system. 45 «.>J - J L T L T U JLTLfli rLTUlf r r Us r Figure 2.23 DS-SS PSK time domain waveforms. Figure 2.24 DS-SS PSK frequency domain power spectrums. 47 floor SS systems can co-exist with other narrowband systems [53], such as analog voice intercom. Finally, more than one spread spectrum signal may be transmitted simul-taneously without excessive mutual interference. (The interference increases as the number of signals increase.) Other SS signals appear as wideband noise to an SS receiver whose despreading code does not match those of other signals; codes are chosen to have small cross-correlations [54], Thus some security of the information sent over power lines ensues, since information cannot be demodulated unless the code is known. 2.3.3 Disadvantages of spread spectrum communications The wide spectral occupancy of spread spectrum signals causes some d i f f i c u l t i e s . The power line has both amplitude and phase characteristics which are frequency dependent. In narrowband communications these non-linearities appear constant within the band of interest. However, wideband signals w i l l suffer distortion unless these effects are equalized [55]. Spread spectrum signals must be demodulated synchronously. The codes in the receiver and transmitter must match or the processing gain of the link w i l l be reduced [56], with the result that code synchronization must be ac-quired and kept for the duration of the message. 48 3. PLC MODEM SYSTEM DESIGN 3.1 Block diagram of a Direct Sequence Spread Spectrum PLC Modem The block diagrams of the transmitter and receiver are shown i n f i g u r e 3.1. In the transmitter the data i s f i r s t spread by the m u l t i p l i e r and then up-converted by the modulator. The r e s u l t i n g radio frequency (RF) s i g n a l i s then amplified and coupled onto the power l i n e . (The sequence of spreading and up-converting can be reversed without a f f e c t i n g the modem character-i s t i c s ) . The code generator which i s discussed i n Section 3.5, generates the high rate pseudorandom sequence which m u l t i p l i e s (spreads) the data stream. The spreading f a c t o r (processing gain) i s about 400 times (26 dB) and i s discussed i n Section 3.2. The PSK modulator converts the baseband s i g n a l to RF. The RF frequency band chosen, which i s discussed i n Section 3.3, i s a compromise between attenuation at high frequencies and noise at low frequencies. The transmit amplifier must be wideband to pass the spread RF s i g n a l and have l i n e a r phase to minimize d i s t o r t i o n . F i n a l l y , the l i n e coupling network, discussed i n Section 3.4, must provide i s o l a t i o n , surge protection, impedance matching and 60 Hz blocking. The receiver i s s i m i l a r i n structure to the transmitter, with addi-t i o n a l system units f o r synchronization at the c a r r i e r , code, and b i t l e v e l , and f o r detection. The d e t a i l s of synchronization are discussed i n Section 3.2. 49 CODE iGENERATORl CARRIER GENERATOR TRANSMITTER BLOCK DIAGRAM DATA> — 0 DATA SPREADER PSK MODULATOR LINE COUPLING NETWORK 3= R E C E I V E R BLOCK DIAGRAM CARRIER SYNCH LINE COUPLING NETWORK .—4 / e(t) PSK DEMODULATOR OESPREADER CODE SYNCH BIT SYNCH k(t) • i COHERENT r rtATA DETECTOR i ->RDATA Figure 3.1 Transmitter and receiver block diagrams for the DS-SS -PLC modem. 50 3.2 Synchronization 3.2.1 Code synchronization i n spread spectrum communications Code synchronization i s c r i t i c a l to spread spectrum communications. If synchronization i s l o s t there i s no processing gain and hence no i n t e r f e r e n c e r e j e c t i o n . Figure 3.2 shows the autocorrelation function of a pseudorandom (m-sequence) which i s N b i t s long. In the example shown N = 15. The autocorrelation function i s at a maximum at zero time s h i f t and then f a l l s o f f sharply as the time s h i f t increases. Note that i f the time s h i f t i s greater than one b i t period then a l l c o r r e l a t i o n i s l o s t . In general, synchronization between the transmitted code sequence and the l o c a l r e p l i c a , generated at the r e c e i v e r , should be within one h a l f of a b i t . A number of techniques have been developed to acquire and maintain t h i s synchronization. One procedure i s to generate a l o c a l r e p l i c a of the code and then " s l i d e " t h i s r e p l i c a and the received s i g n a l by each other u n t i l they c r o s s c o r r e l a t e . Because of the large number of positions to t r y , the s l i d i n g c o r r e l a t o r Is slow to acquire synchronization. As a r e s u l t modifications are often added to reduce synchronization delay. Sometimes a short code i s transmitted f i r s t to acquire synchronization and then the long code Is used to transmit the data. (The code must be as long as possible to spread the data uniformly). Other times s p e c i a l codes with p e c u l i a r c r o s s -c o r r e l a t i o n properties which speed synchronization are used [50]. Other techniques such as matched f i l t e r receivers or sequence estimators are quicker but have other disadvantages, including s e n s i t i v i t y to noise and high implementation c o s t s . Figure 3.2 Autocorrelation of an m-sequence of N bits (N=15). 52 3.2.2 Code synchronization by zero crossings Whichever process i s used, synchronization slows down and complicates the transmission of data through a spread spectrum modem. For this reason synchronization should be simplified i f possible. Some applications such as sat e l l i t e communications make use of universal timing [50]. In this case a l l transceivers operate off the same timing signal and only propagation delays determine the phase of the received signal. A universal timing signal i s already present in power line carrier communications - the a.c. power frequency. Code synchronization can be achieved automatically i f a l l transceivers synchronize to the 60 Hz zero crossings as shown i n Figure 3.3. In this example the code i s fifteen bits long and starts on every positive-going zero crossing of the power line voltage. Receivers generate local replicas of the code which, ignoring propagation delays, matches the received code generated by the same zero crossings at the transmitter. 3.2.3 Carrier and bit synchronization Synchronization by zero crossings can be extended to both carrier and bit synchronization as well as code synchronization. Synchronizing the carrier, code, and bits gives a coherent communications system which performs better than a non-coherent system [57]. Because zero crossings provide this coherence the cost and complexity disadvantages normally associated with coherent communications may be substantially reduced. Bit synchronization can be accomplished by limiting the data rate to an integer multiple of the power line frequency. This may seem restrictive at 53 Figure 3.3 Code synchronization by zero crossings. 54 f i r s t but most of the services lis t e d in Chapter One require data rates of less than 60 bits/sec. These services can therefore be accommodated by one bit every power line cycle. Higher data rate services can also be accommo-dated because the standard data transmission rates of 300, 1200, 2400 and 9600 bits/sec are a l l multiples of 60 Hz. Similarly carrier synchronization can be acheived by making the carrier frequency a multiple of the code rate and hence a multiple of the power line frequency. Again this may seem restrictive but there are many multiples to choose from. Thus the RF spectrum can be placed wherever desired. (See Section 3.3) Figure 3.4 illustrates the b i t , code, and carrier synchroniza-tion. The bit frequency i s 60 bits/sec, the code length is 15 bits, and the carrier frequency equals the code rate. Note that a l l the timing (synchroni-zation) is referenced to the positive going zero crossings of the power line voltage. The bottom waveform In Figure 3.4 is the spread spectrum signal which results from multiplying the data, the code, and the carrier together. 3.2.4 Implications of power line synchronization Power line synchronization i s simple to implement and hence is an inexpensive way to provide synchronization to a l l transceivers. However, power line synchronization does have a number of disadvantages. Propagation delays of both the 60 Hz and spread spectrum signals w i l l cause phase uncertainties at the receiver. Propagating at the speed of light the SS signal and the zero crossings of the 60 Hz line voltage move about 300 a every microsecond. Inside a building this delay is negligible because Figure 3.4 A coherent, l i n e synchronized, spread spectrum system. 56 transmission distances are short. However, on the d i s t r i b u t i o n or high v o l t -age networks the distance between transceivers may be many kilometres. As an example, consider the case of a receiver located at a substation receiving a message from a transmitter located at a house 10 Km away. The receiver sees a zero crossing and s t a r t s i t s l o c a l code generator. The same zero crossing then propagates 10 Km to the house where the transmitter senses i t and s t a r t s i t s l o c a l code generator. The 10 Km delay Is about 30 micro-seconds, assuming signals propagate at close to free space v e l o c i t y (the speed of l i g h t ) . The transmitter's s i g n a l then propagates the 10 Km back to the receiver making a t o t a l delay of about 60 microseconds. If the code rate Is 10 Kbits/sec then each chip ( b i t of the code) has a 100 usee duration. Thus a delay of 60 usee i s s i x tenths of a chip. From Figure 3.2, 6/10 of a chip reduces the processing gain by more than f i f t y percent. If the house had been 20 Km away there would be no processing gain at a l l . Fortunately the transceivers are not mobile so I t i s possible to pre-compensate f o r the propagation delay. Precompensation would be done at the transmitter as follows: For a message t r a v e l l i n g down the l i n e i n the d i r e c -t i o n of power flow, no compensation i s necessary because both the power and the SS s i g n a l w i l l t r a v e l at close to the same v e l o c i t y . For a message t r a v e l l i n g up the l i n e opposite the d i r e c t i o n of power flow the transmitter w i l l s t a r t the code and the data b i t s s l i g h t l y before the zero c r o s s i n g . The distance between the transmitter and the receiver w i l l determine the amount of precompensation such that the s t a r t of the code w i l l reach the receiver at the same time the zero-crossing does. 0 57 Another source of phase uncertainty e x i s t s because wall sockets, and therefore the transceivers attached to them, can be connected to d i f f e r e n t phases of the power system. In normal r e s i d e n t i a l w i r i n g , each house i s connected to one of the three phases of the d i s t r i b u t i o n network. In a d d i -t i o n , each wall socket i n the house connects to one of the two phases. Thus there are s i x possible phases as shown In Figure 3.5. In a d d i t i o n to the s i x possible phases, each transceiver can be plugged into one of two p o l a r i t i e s because the plugs are not normally p o l a r i z e d . The r e s u l t i n g phase ambiguity can be resolved by making the code repeat s i x times within each c y c l e . In t h i s way the code w i l l r e s t a r t at each of the s i x possible phases. I t doesn't matter i n t h i s instance to which phase the transceivers are connected. The p o l a r i t y ambiguity can be resolved by d i f f e r e n t i a l l y encoding the data [57]. In d i f f e r e n t i a l encoding, the r e c e i v -ed p o l a r i t y i s unimportant. Only the d i f f e r e n c e i n p o l a r i t y between the current b i t and the previous b i t determines the data. For example, a "one" would be encoded by making the current b i t opposite from the previous one while a "zero" would be the same as the previous b i t . A small penalty i s paid when the method i s used because each received error u s u a l l y generates two decoded e r r o r s . A f i n a l source of phase uncertainty i s phase noise ( j i t t e r ) . The phase noise w i l l cause the zero crossings to j i t t e r , with the r e s u l t that codes i n the transmitter and receiver w i l l move with respect to each other. The mag-nitude of t h i s j i t t e r w i l l determine the e f f e c t i v e processing gain of the system. For example, i f the j i t t e r i s l e s s than one tenth of the chip dura-t i o n , the e f f e c t i v e processing gain w i l l not be s i g n i f i c a n t l y l e s s than the POSSIBLE POWER LINE VOLTAGES Figure 3.5 The s i x possible power l i n e voltages. 5 9 maximum gain obtainable with no j i t t e r . If, however, the j i t t e r i s about equal to the chip duration then the effective gain w i l l be about one half of Its maximum value. J i t t e r in the zero crossing comes from three sources. Amplitude noise (AM) on the 60 Hz line voltage w i l l convert to phase noise (PM) when passed through a threshold detector as shown in Figure 3.6. This AM to PM conver-sion can be significantly reduced by f i l t e r i n g the line voltage to remove the amplitude noise before passing i t to the threshold detector. Phase noise is also produced at generating stations [32]. The com-posite load presented to the generators w i l l vary as individual loads are turned on and off. These varying loads cause the generators to vary their speed of rotation. As the speed of rotation changes so does the instantan-eous frequency as shown in Figure 3.7. This instantaneous frequency varia-tion i s equivalent to phase noise. This source of phase noise w i l l cause j i t t e r but a l l transmitters and receivers w i l l encounter the same j i t t e r . Thus, the zero crossings at different transceivers w i l l not move with respect to each other and as a result the processing gain w i l l not be reduced. The f i n a l source of j i t t e r i s d i f f i c u l t to eliminate. This j i t t e r Is also caused by varying loads but w i l l be different for different transmitters and receivers. The mechanism shown in Figure 3.8 represents two equal power line segments (Z T) terminated by two different loads (Z. and Z. ). It should be noted that the phases of the 60 Hz voltages (./V^  and Jy^) depend on the loads Z. and Z. . These loads w i l l be different and time varying so the 1 2 60 L I N E VOLTAGE Z E R O CROSSING D E T E C T O R O U T P U T t — J I T T E R — t Figure 3.6 Zero crossing jitter due to noise on the 60 Hz line voltage. Figure 3 . 7 Instantaneous frequency v a r i a t i o n s i n e l e c t r i c a l generators. 62 Figure 3.8 The e f f e c t of unequal loads on zero crossing j i t t e r . 63 60 Hz phases at the receiver and transmitter - and hence the zero crossings -w i l l vary with respect to each other. Fortunately, the line Impedances (Z ) Li are much smaller than the load Impedances, so the phase changes are very small. Figure 3.9 is an oscilloscope photograph which illustrates the zero crossing j i t t e r . Notice that peak one-sided j i t t e r i s about 10 usee. The amount by which the j i t t e r reduces the processing gain depends on the code rate. If the code rate is 10 Kbits/sec (chip duration = 100 usee), the j i t -ter w i l l be less than 10% of the chip duration, and the effective gain w i l l not be significantly reduced. If, however, the code rate Is 100 Kbits/sec (chip duration = 10 jjsec) the j i t t e r w i l l be close to 100%, and the effective gain i s reduced by about one half. The theoretical processing gain i s the ratio of spread to unspread bandwidth, or the ratio of code rate to data rate as shown in equation (3.1). Processing gain = G = c o d e (3.1) P Rdata For a fixed data rate, the processing gain increases as the code rate increases. However, as the code rate increases the j i t t e r becomes a larger percentage of chip duration, and the effective processing gain decreases. At some point, as shown in Figure 3.10, a maximum processing gain i s reached where any further increase In code rate causes a net decrease l n processing gain. It i s possible to solve for this maximum processing gain. Assume N code chips per 60 Hz cycle (R ^ =» N x 60) and a measured peak j i t t e r of J Horizontal = 20 ysec/div. Figure 3.9 Measured zero crossing PROCESSING GAIN (dB) to PROCESSING GAIN WITH • ZERO PHASE JITTER 30 20 10 > GAIN REOUCTION DUE TO JITTER EFFECTIVE PROCESSING GAIN CODE RATE DATA RATE 10 100 1000 10.000 Figure 3.10 Maximum processing gain. 66 sec. Redrawing Figure 3.2 i n the vi c i n i t y of zero time shift (see Figure 3.11) gives an equation for the effective processing gain as a function of N and J: G g f f(J,N) = G p [ l - J x N x 60] (3.2) The maximum processing gain (G^) is directly proportional to the chip rate and hence directly proportional to the number of chips per 60 Hz cycle (N). Thus, G can be written as follows: P G = K • N (3.3) p p Combining (3.3) and (3.2) gives: G ,-:(T,N) = K • N - K • 60 • J • N 2 (3.4) erf P p Differentiating (3.4) with respect to N gives: 3G f £(T,N) — S i i - K - K • 60 • J • 2 • • N . (3.5) 3N P P Equating (3.5) to zero and solving for N yields: N = Kp 1 , N ° p t K • 6 0 • 2 • J ~ 1 2 0 • J ( 3 ' 6 ) P Equation (3.6) shows that i f the j i t t e r i s small one can select a large value for N, in which case a high code rate and therefore a large processing gain results. Inserting (3.6) back into (3.4) yields: K K • 60 • J K . W T ) m a x = — - — ~ 2 = — C 1 " ^ <3'7) err max 1 2 Q J 1 2 Q c U Q # ^ 1 2 Q J z Thus at the maximum gain, the effective gain i s down by 1/2 (3 dB) from i t s theoretical value as a result of the j i t t e r . 6 7 Figure 3.11 E f f e c t i v e processing gain. 68 A measured value for J is about 10 usee. Using this value In (3.6) gives N Qp t » 830. At the minimum data rate of one bit every 60 Hz cycle, the theoretical maximum processing gain from (3.1) i s : R . N . x 60 G = -£2*i = °P t = N - 830 (3.8) p R, _ 1 x 60 opt •max data From (3.7) the effective maximum processing gain i s one half of this, o r G p e f f a 4 0 ° -max Converting this gain to a power ratio gives: G e f f ( d B ) = 1 0 X l o g 1 0 ( Geff ' ) " 2 6 d B ( 3 , 9 ) max max Therefore, the maximum processing gain i s limited to about 26 dB when zero crossings of the power line voltage are used for synchronization. In this application 26 dB is adequate, and line synchronization would therefore appear to be viable. 3.3 Frequency Band of Operation The transmission characteristics of the power system include high noise power at low frequencies and high attenuation at high frequencies (see Sec-tion 2.1). These characteristics indicate that the frequency band between 10 kHz and 100 kHz is most suitable for PLC. There are, however, special cases. On the HV network the conductors are large and therefore have lower attenuation at high frequencies than distribution or intrabuilding wiring. Also, most PLC communications on the HV network do not pass through power 69 transformers, which are a source of high frequency attenuation. Therefore, PLC on the HV network i s generally at a higher frequency. Within a b u i l d i n g , distances are so short that attenuation - even at higher frequencies - i s not too lar g e . It i s also f e a s i b l e to i n s t a l l high frequency bypass capacitors between the phases so transformer losses w i l l not occur at high frequencies. These f a c t o r s allow PLC communications at higher frequencies within a b u i l d i n g . In the case of narrowband PLC, the choice of c a r r i e r frequency i s c r i t i c a l . It would not be advisable to set the c a r r i e r frequency to 10 kHz, for example, because of interference from OMEGA, a marine l o c a t i o n wervice which uses high power transmitters operating at 10 KHz [58]. It would be better to set the c a r r i e r between two of the 60 Hz harmonics to reduce the noise l e v e l [42]. For SS communications a wide frequency band i s used and the above con-siderations are not as c r i t i c a l . For simple synchronization i t i s important instead to make the c a r r i e r frequency a multiple of the l i n e frequency (see Section 3.2). To make things even simpler the c a r r i e r frequency should be equal to the code r a t e . In that way only one frequency need be synthesized. Using these constraints the modem w i l l have the c h a r a c t e r i s t i c l i s t e d i n Table 3.1. The processing gain l i s t e d i n Table 3.1 i s an approximation which takes into account the gain reduction due to j i t t e r (see Section 3.2). Using these frequencies the modem w i l l occupy the spectrum as shown i n Figure 3.12. 70 Data rate (D T) 60 bits/sec. Code rate (C T) 14 Kbits/sec. Carrier frequency 14 KHz Processing gain 10 l o g 1 0 (C T/D T) - J i t t e r reduction » 20 dB. Table 3.1 Modem characteristics. S(f) Figure 3.12 Spectral occupancy of the spread spectrum s i g n a l . 72 3.4 The Line Coupling Network The l i n e coupling network connects the modem to the power system. I t provides i s o l a t i o n , surge protection, 60 Hz blocking, impedance matching and low Insertion loss of the PLC s i g n a l . I s o l a t i o n must be provided because most power plugs are not p o l a r i z e d , that i s the plug can be inserted i n two ways and the ground wire w i l l be connected to ei t h e r of the two modem wires as shown i n Figure 3.13. I s o l a -t i o n i s most simply achieved magnetically with a transformer. Surge protection i s necessary because high voltage transients are present i n the power system and would otherwise damage the modem. The coupling network must simultaneously block the 60 Hz power and pass the PLC s i g n a l . This can be accomplished by a high pass f i l t e r . The ^ f i l t e r ' s cut-off frequency should be as low as possible so I t w i l l pass most of the PLC s i g n a l while s t i l l providing s u f f i c i e n t attenuation of the power s i g n a l . F i n a l l y , the coupling network must match impedances between the modem and the power l i n e , to maximize the s i g n a l to noise r a t i o . Unfortunately, t h i s matching i s d i f f i c u l t to achieve because both the noise and power l i n e impedances are h i g h l y v a r i a b l e . (See Section 2.1) Figure 3.14 shows the s i t u a t i o n at the transmitter. The voltage on the l i n e (V^) w i l l contain both the transmitted s i g n a l (V ) and noise (V ). Figure 3.14 can be furth e r s i m p l i f i e d by combining the two noise sources and t h e i r impedances as shown i n Figure 3.15. In t h i s case the s i g n a l power P s i s given as GND HOT 2 WIRE 2 IS HOT 1 WIRE 1 IS GND A , . . _ 2 WIRE 2 IS GND 9—i - 1 J 1 WIRE 1 IS HOT Figure 3.13 P o l a r i t y ambiguity i n modem connection. Figure 3.14 Impedances seen by a PLC transmitter. 75 Figure 3.15 Equivalent impedances seen by a PLC transmitter. 76 Z„ 2 , rv *L_1 ps = " t = * z" T.z" N (3'10> Z T + Z N While the noise power P^ i s given by 2 Tv ' - - 1 V „ Z N • Z + Z„J Z T + Z N Thus the SNR i s P V 2 Z 2 SNR = / = tf) • (^) (3.12) N N T D i f f e r e n t i a t i n g (3.12) with respect to the transmitter's impedance (Z^,) and equating t h i s to zero gives the Z T which maximizes the transmitted SNR. KSMO = _ I x l 2 x z-3 = 0 (3 13) azT 2 x Lv^  . z NJ x  Li  u u * 1 3 ; Unfortunately, (3.13) i s only zero i f Z^ i s zero. This value can not be obtained p r a c t i c a l l y but equation (3.13) i n d i c a t e s that should be made as small as possible to make the SNR as large as p o s s i b l e . A s i m i l a r analysis can be done at the PLC receiver and leads to the same conclusions. The receiver's impedance should be as small as p o s s i b l e . Figure 3.16 shows the f i n a l configuration of the coupling network. « TRANSMITTER R T HIGH PASS FILTER T \ RECEIVER Figure 3.16 The coupling network. 78 3.5 The Code Generator The code, generator provides a b i t stream k(t) which must have a number of c h a r a c t e r i s t i c s . F i r s t the code should be part of a family so that a transmitter can send to a s p e c i f i c receiver by generating the receiver's code sequence. Each receiver has a unique code sequence which Is a member of the family. Codes of t h i s type allow privacy. An unauthorized receiver cannot decode the message unless i t knows the p a r t i c u l a r code used. (However, a user with s u i t a b l e processing equipment could learn the code i d e n t i t y ) . Second, the c r o s s - c o r r e l a t i o n s between the code sequences should be low. Codes of t h i s type allow code d i v i s i o n multiple access (CDMA) i n that a number of messages can be transmitted simultaneously. Any given receiver can only decode (despread) a message whose code matches. A l l other messages -provided t h e i r codes have low c r o s s - c o r r e l a t i o n with the receiver's code -w i l l appear as background noise. F i n a l l y , the codes should have a low autocorrelation except at zero time s h i f t . (See Figure 3.17.) This r e a l l y means the codes should be as random as p o s s i b l e . Truly random code sequences w i l l have no d i s c r e t e l i n e s i n t h e i r power spectrum, making them the l e a s t susceptible to fades and inte r f e r e n c e . In other words, random codes have no concentrations of s i g n a l power which could be removed by a fade or i n t e r f e r e d with by a jamming s i g n a l . Unfortunately, a t r u l y random s i g n a l could not be reproduced at the receiver so pseudo-random codes must be used as a compromise. M-sequences have t h i s i d e a l autocorrelation property. Unfortunately, m-sequences have poor c r o s s - c o r r e l a t i o n properties because members are j u s t time-shifted versions of each other. Thus, a l l c r o s s - c o r r e l a t i o n functions •NT NT Figure 3.17 H e a l autocorrelation function of the spreading code. *KM.K(M-L) ( T ) / E C N4-«NT -N-UT NT Figure 3.18 Cross-correlation between two m-sequences. 80 between m-sequences w i l l have a "peak" where the time s h i f t s cancel. (See Figure 3.18.) This peak i n the c r o s s - c o r r e l a t i o n function can cause d i f f i -c u l t i e s when a receiver i s attempting to acquire synchronization. The receiver may f a l s e l y "lock" to the inc o r r e c t transmitter, i n a SDMA system, because the s l i d i n g c o r r e l a t o r w i l l lock onto the f i r s t c o r r e l a t i o n peak i t encounters. Because the PLC modems use uni v e r s a l timing they can not f a l s e l y acquire synchronization. Therefore, the PLC modems can use simple m-sequences. 81 4. PLC MODEM IMPLEMENTATION 4.1 The Transmitter The PLC transmitter accepts data which i s then spread, modulated, amplified, and coupled into the power system. The circuits which implement these functions are discussed in this section. 4.1.1 The transmitter block diagram Figure 4.1 is a block diagram of the PLC spread spectrum transmitter. Synchronization for the modem i s based on the power line frequency and allows a simple implementation for the code generator which is discussed in Section 4.1.4. The data spreader and binary phase-shift-keyed (BPSK) modulator are also examined in Section 4.14. The transmit amplifier, built from an inte-grated circuit (IC) developed for the audio industry, i s examined i n Section 4.1.5. The power supply and protection circuits are discussed in Section 4.1.2 while the coupling network i s detailed In Section 4.1.6. 4.1.2 The power supply The ci r c u i t diagram for the power supply Is shown in Figure 4.2. The transformer provides isolation and two phases of low voltage alternating current (AC). The bridge r e c t i f i e r and capacitors Cl and C2 generate posi-tive and negative unregulated direct current (DC) voltages. These two volt-ages are regulated by UI and U2 which are fixed five volt positive and nega-tive series-pass regulators. Capacitors C^, C^, C^, and C^ provide addi-tional f i l t e r i n g . 82 PROTECTION CIRCUIT ZERO CROSSING DETECTOR ZERO CLOCK GENERATOR 2ER0 POWER SUPPLY -4-5V -i-SV -OAIA COOE GENERATOR CODE DATA SPREADER CARRIER COUPLING NETWORK ADATA Tx AMP 'DATA _ MDATA BPSK MODULATOR *C TDATA ^ Figure 4.1 The transmitter block diagram. 83 Fl - 1/2 Amp Cl , C2 = 2200 uF <? 16V C3, C4 - 10 uF @ 25V D2 » GE Transorb #V130LA20A UI - uA7805 U2 «= uA7905 Tl = Hammond #161K12 (6.3V to centre tap) DI - EBR //VS048 (2 Amps) Figure 4.2 The power supply. 84 Also shown i n Figure 4.2 i s the protection c i r c u i t . I t consists of a fuse and a transorb which protect the modem from the high voltage transients which occur on the power l i n e . 4.1.3 The code generator Figure 4.3 Is a d e t a i l e d block diagram of the code generator. The code generator accepts the power l i n e voltage as i t s input and generates the pseudorandom sequence and the c a r r i e r , which are both synchronized to the power l i n e frequency. The zero-crossing detector generates a pulse (ZERO) on every p o s i t i v e going zero-crossing of the power l i n e voltage. This pulse i s then used to synchronize both the c a r r i e r and the code sequence. Figure 4.4 i s the schematic of the zero-crossing detector [59], while Figure 4.5 Is an o s c i l l o -scope photograph showing i t s performance. The clock generator i s driven by a colour-burst (3.58 MHz) c r y s t a l and synchronized by the ZERO pulse. Figure 4.6 Is the schematic of the clock generator and Figure 4.7 i s i t s timing diagram. The c i r c u i t d e l i v e r s a c a r r i e r which i s synchronized to the power l i n e voltage. The 3.58 MHz o s c i l l a t o r [60] generates a high frequency clock which i s then divided down to produce the c a r r i e r . The two d i v i d e r s together divide by 256 so the c a r r i e r frequency i s about 14 KHz. Line synchronization i s provided by the CLR s i g n a l , which i s derived from the ZERO pulse as shown i n the timing diagram (Figure 4.7). The CLR pulse c l e a r s the two counters so they always s t a r t counting from zero at every zero-crossing of the power l i n e voltage. Thus, the c a r r i e r goes through about 233 cycles before being POWER LINE ZERO CROSSING ZERO DETECTOR Hi/60 SEC [ — " J ~ Y ~ 3.58 MHz CLOCK CLR GENERATOR (T256) • CLK CODE M-SEQUENCE CLK GENERATOR (127 BITS) CLR CARRIER (U KHz) Figure 4.3 The code generator block diagram. POWER LINE Figure 4.4 The zero crossing detector. Horizontal = 2 ms/div. Top trace = power line voltage. Bottom trace = output of zero crossing detector Figure 4 . 5 Output of the zero crossing detector 88 ZERO [ > CARRIER CARRIER ZERO «5V 13 121 10 A7on«7on 10-60pF l-^ -|[|f4 3.58MHz 5 J 7 •5/ 1lTl0T9 , V A CO QAQBQC QDET LD | ) 74LS163 (T16) S R Q K D O D I D 2 D 3 EPGNd 1 Up M L 12 ) 74LS74 C L R D C K P S Q Q GNDl A A A A A A *5V •5V •5V |15f^ 1l3fl2fllTloT9 y^COOAQBQCODEr LD p 7ALS163 (T16) QROK DO D1D2D3 E P G N D |1 |2jL3M5j6l7l8 • • • « *S/ • Figure 4.6 The clock generator schematic. 89 l i I I l i CLR CARRIER Figure 4.7 The clock generator timing diagram. 90 restarted by the CLR pulse at the next zero-crossing. The code generator accepts the carrier signal and the ZERO pulse. It then generates the code sequence which i s used to spread the data. The code generator Is simply an m-sequence generator [61] which i s restarted by the ZERO pulse so that code synchronization i s simplified (see Section 3.2.2). Figure 4.8 is the schematic of the code generator and Figure 4.9 i s an oscilloscope photograph showing i t s operation. The two f l i p s flops (74LS74) format the ZERO pulse so i t can be used by the eight bit shift register (74LS299). Taps one and seven of the shift register are brought out to an exclusive NOR gate (74LS266). The gate's output then feeds back to the input of the shift register. This feedback arrangement generates a pseudorandom m-sequence which i s 127 (2 7-l) bits long. Because there are 233 carrier cycles within every 60 Hz cycle the code sequence repeats almost twice between each restart pulse. 4 . 1 . 4 The data spreader and BPSK modulator The data spreader is simply a di g i t a l multipler which multiplies the low speed data by the high speed code. This multiplying process spreads the spectrum of the data. The multiplier can be implemented by an exclusive NOR gate as shown in Figure 4.10. Also shown in Figure 4.10 is the BPSK modulator. It too can be imple-mented by an exclusive NOR gate. The carrier i n this case i s a square wave so the BPSK modulator is sometimes called a biphase modulator. 91 CARRIER [ > ZERO CARRIER CODE <J-12 1lTl0l9 18 ) 74LS74 °^A D A C KA^A QA 5A G N D : «5V 3F •9/ •5V T 2 1 1 1 Vcc S1SL0HH F D BCLKSR ^ 74LS299 (SHIFT REGISTER) SO G1 G2 G £ C A Q / O R G N D j »5V 7 7 9 j p •Sv* 1KX1 1/4 OF 74LS266 Figure 4.8 The code generator. Horizontal Top trace Bottom trace Code rate Code length 2 ras/div. restart pulse the spreading code 14 Kbits/sec. 127 b i t s Figure 4.9 Operation of the code generator. 93 •5V •5Y 1KA 1KT1 1/4 OF 74LS266 1/4 OF 74LS266 CODE £>" CARRIER^-v*-DATA SPREADER 3 I > M DATA N V ' BPSK MODULATOR Figure 4.10 The data spreader and BPSK modulator. 94 Figure 4.11 i s a spectrum analyzer photograph which shows the s p e c t r a l occupancy of the data (a) a f t e r i s has been spread by the m u l t i p l i e r and (b) a f t e r i t has been modulated by the BPSK modulator. 4.1.5 The transmit a m p l i f i e r The transmit a m p l i f i e r accepts the modulated data (M^^^,^) and d e l i v e r s an amplified d i f f e r e n t i a l version to the coupling network. The actual power delivered to the coupling network depends on the l i n e impedance, which i s time varying. Under normal circumstances, however, the transmitted power i s about .5 watts. Figure 4.12 i s a schematic of the transmit a m p l i f i e r . 4.1.6 The l i n e coupling network The l i n e coupling network i s shown i n Figure 4.13. The c i r c u i t con-s i s t s of an audio matching transformer, which provides i s o l a t i o n and imped-ance matching. Also included i s a t h i r d order Butterworth high pass f i l t e r , which blocks the 60 Hz power s i g n a l while passing the high frequency PLC s i g n a l . The transmit a m p l i f i e r has a low output impedance (See Section 4.1.6) which i s lowered s t i l l further by the matching transformer. The equivalent source Impedance, on the secondary side of the transformer, i s l e s s than one ohm. This low source Impedance Is required f o r a high transmitted SNR (See Section 3.4). The high pass f i l t e r i s designed f o r a nominal cut-of f frequency of 6 KHz. This frequency i s adequate to pass most of the PLC spectrum (see Figure 3.12) while providing more than 100 dB of attenuation at 60 Hz. However, the 95 V e r t i c a l = 10 dB/cm. Horizontal = 10 KHz/cm. Figure 4.11(a) Frequency spectrum of the spread data (^DATA^ * V e r t i c a l = 10 dB/cm. Horizontal = 10 kHz/cm. Figure 4.11(b) Frequency spectrum of the modulated data ( M n A T A ) . 96 100KA |Z2KnJ MDATA!> \h 100KQ -vW ? -5V -5V -5V J J -V rcV -V "t> ADATA BIAS LM1877 (DUAL POWER AMPS) -V -V 1 1 1 1 -5V -5V -5V c 3.311 1 «5V <3.3A -vW— 100KXI DATA Figure 4.12 The transmit a m p l i f i e r . W A [ > WTA [> £ > L D A T A £> L DATA C5.C6=13.2/JF (250 V) L1 = 2.5mH ( 2A ) T2= 10011:3.3X1 ( 1.5W) Figure 4.13 The l i n e coupling network. 98 line impedance w i l l vary (see Figure 2.13) so the cut-off frequency w i l l vary as well. Figure 4.14 and Table 4.1 i l l u s t r a t e the transfer function of the coupling network. The load Impedance is varied with frequency to simulate the power line Impedance which varies with frequency. The primary side volt-age (Vp) is kept constant to simulate the signal which would come from the low impedance transmit amplifier. Notice that the signal level i s f l a t between about 6 KHz and over 100 KHz. The low frequency losses are due to the high pass f i l t e r while the high frequency attenuation is due to trans-former loss. FREQ (KHz) 3 6 10 20 30 40 60 100 300 600 1,000 V p(volts p-p) V(volts p-p) 10 .5 1 10 1.6 2 10 1.8 3 10 1.6 7 10 1.6 10 10 1.6 15 10 1.6 20 10 1.6 40 10 1.5 90 10 1.3 100 . 10 1.0 100 Transfer function of the coupling network. Figure 4.15 Is an oscilloscope photograph of the PLC signal at the input (top) and the output (bottom) of the coupling network. The load imped-ance was set here to ten ohms as an approximation of the line impedance in the frequency band of the PLC signal. 99 LINE VOLTAGE (VOLTS P-P) 7 6 5 i. J 6 5 A .1 JD9 £8 JD7 K £6 JW 03 02 X X X m t i t i I I III r I ' l l m n i t i u m i 3 « 5 6 7 8910 20 30 40 50 60738090130 2D0 300 «X) 603 8001000 FREQUENCY (KHz) Figure 4.14 Transfer function of the coupling network. Horizontal = Top trace = Bottom trace = Line impedance = 100 usec/div. ^DATA LDATA ion Data transmission through the coupling network. 101 The output signal i s only slightly distorted. The distortion is a result of the high pass f i l t e r in the coupling network which removes frequency components below 5 KHz. In addition there is negligible phase shift through the network. Figure 4.16 i s a spectrum analyzer photograph which shows the spectral occupancy of the signal delivered by the trans-mitter. 4.2 The Receiver The PLC receiver accepts the PLC signal from the power l i n e . The signal i s then amplified, demodulated, despread and f i n a l l y detected. The circuits which implement these functions are discussed in this section. 4.2.1 The receiver block diagram Figure 4.17 i s a block diagram of the PLC spread spectrum receiver. The circuits which implement the protection network, coupling network, power supply, zero crossing detector, clock generator and code generator are simi-lar to those found in the transmitter (see Section 4.1) Only the operating voltages are different: ±15 volts for the receiver and ± 5 volts for the transmitter. The higher voltages in the receiver are required by the data despreader (see Section 4.2.3). Figure 4.18 is a schematic of these c i r -cuits . The receive amplifier is descussed in Section 4.2.2. This ci r c u i t amplifies the received PLC signal before passing i t on to the demodulator and data despreader, which are examined in Section 4.2.3. Finally, Section 4.2.4 concerns the data detector which generates the received data (Rp^-p^) • V e r t i c a l Horizontal Line impedance Figure 4.16 The spectrum = 10 dB/cm. = 10 KHz/cm. = ion of the transmitted s i g n a l . 103 PROTECTION NETWORK "lINE Z E R O C R O S S I N G DETECTOR Z E R O P O W E R S U P P L Y t-sv I-1SV COUPLING NETWORK Rp L C CLOCK GENERATOR ZERO CLOCK C A R R I E R PEMOOULATCF [OESPREADEF C O O E G E N E R A T O R C O O E DATA OETECTOR Figure 4.17 The receiver block diagram. 104 Figure 4.18 C i r c u i t s s i m i l a r to the PLC transmitter. 105 A.2.2 The receive a m p l i f i e r The receive amplifier i s a d i f f e r e n t i a l a m p l i f i e r which boosts the received s i g n a l (Rp^) l e v e l before passing i t on to the m u l t i p l i e r . The schematic diagram of t h i s a m p l i f i e r i s given i n Figure A.19. 4.2.3 The data demodulator and despreader Once the received s i g n a l has been amplified I t must be demodulated and despread before being passed to the data detector. The amplified s i g n a l (R^) i s demodulated by multiplying i t by the l o c a l l y generated c a r r i e r . R D E M = R A X R C A R ( 4 a )  RCAR * 8 ^ e ^ o c a ^ ^ v generated c a r r i e r and Rpg^ i s the demodulated s i g n a l . Next the demodulated s i g n a l i s despread by multip l y i n g i t by the l o c a l l y generated code sequence (R ). hzS = ^EM X RC0DE ( A * 2 ) Rpgg represents the despread s i g n a l . These two m u l t i p l i c a t i o n s are shown i n Figure A.20. Because the amplified s i g n a l (R^) i s analog, both the m u l t i p l i e r s i n Figure A.20 must be four-quadrant analog m u l t i p l i e r s . These are more c o s t l y and les s accurate than an exclusive OR-gate (XOR) which can multiply two d i g i t a l s i g n a l s . However, combining (A.l) and (4.2) r e s u l t s i n a t r i p l e m u l t i p l i c a t i o n i n which two of the terms (RQ^JJ) and (RcoDE^ a r e <*i2ita-'- s i g n a l s . By chang-ing the order to m u l t i p l i c a t i o n , which i s commuative, one of the analog 106 LPLC [> yW- •vw-•15V R P L C [ > R A s R 2 / R l " ( R P L C " R P L C } Figure 4.19 The receiver a m p l i f i e r . 107 DEM - £ > R D E S 'CAR CODE CARRIER GENERATOR CODE GENERATOR ZERO I> Figure 4.20 The data demodulator and despreader. 108 m u l t i p l i e r s can be replaced by a d i g i t a l one as shown i n Figure 4.21. Figure 4.22 i s the schematic of t h i s s i m p l i f i e d data demodulator and despreader. The performance of the data demodulator and despreader i s shown i n Figure 4.23 and Figure 4.24. Figure 4.23 i s an o s c i l l o s c o p e photograph show-ing the inputs and output of the analog m u l t i p l i e r . The top trace i s the transmitted data (T ^j,^) • Th e next trace i s the output of the XOR gate, which Is a product of the l o c a l l y generated c a r r i e r and code. The t h i r d trace i s the amplified s i g n a l (R^), while the bottom trace i s the output of the m u l t i p l i e r - the demodulated and despread s i g n a l (RJJES^* F^-Sure 4.24 i s an expanded view of Figure 4.23 at the point where the transmitted data changes p o l a r i t y . This shows more c l e a r l y the operation of the analog m u l t i p l i e r . 4.2.4 The data detector The data detector i s of the integrate-and-dump v a r i e t y . The block diagram of the detector Is shown i n Figure 4.25 while i t s schematic i s de t a i l e d i n Figure 4.26. The integrator integrates the despread s i g n a l (Rpgg) over one b i t period. This s i g n a l i s then applied to the threshold detector (comparator) the output of which i s sampled at the end of the b i t period. Figure 4.27 shows the performance of the data detector. The top trace i s the despread s i g n a l (Rjjug)* The next trace i s the ZERO s i g n a l , or the sampling point. The next trace i s the output of the i n t e g r a t o r . The bottom trace shows the output of the sampler, which i s the received data s i g n a l (Rp^j.^) • 1 0 9 ANALOG MULTIPLIER — R CAR * RCODE 6 R CAR IT DIGITAL MULTIPLIER RCODE CARRIER GENERATOR CODE GENERATOR DES ZERO t> Figure 4.21 A s i m p l i f i e d data demodulator and despreader. 110 1/4 CD4077 CAR *C0DE |E>7 R C A R " R C 0 D E DIGITAL MULTIPLIER -T5V 1/JF IOKH. 4hVvV-A N A L O G M U L T I P L I E R # l*15V 7.5KQ 7.5KO. IOKH . 0 « 7 I M F IOKH R A > — 1 £ — V V \ A 10 11 5 MCU95 -(X«Y) (X»Y) 10KA «15V| \ W - f [OUTPUT. P F 5 E T 5 - 2 0 K A AAA/ I-15V 10KA | 20KQ. L KKft 1.8KQ ^.1.8KA V D E S Figure 4.22 Schematic of the simplified data demodulator and despreader. Figure 4.23 Operation of the data demodulator/despreader. 112 Figure 4.24 Expanded view of the data demodulator/despreader. 113 R D E S > SAMPLER DATA CLOCK (DUMP) ZERO £ > -(SAMPLE) Figure 4.25 The data detector. 114 I.5M0. *DES1>—M*-INTEGRATOR 100A ANALOG SWITCH M*-J0047/L.F HI * ZERO [> ZERO[> (SAMPLE) SAMPLER •15V 'ISv 45V ]Jo[l2|llllXT8 ) CD4013 "C R DATA ) Figure 4.26 The data detector schematic. 1 1 5 116 5 . MODEM PERFORMANCE 5.1 Test Procedure Because the transmission c h a r a c t e r i s t i c s of the power system vary over time, p o s i t i o n , and frequency i t i s d i f f i c u l t to determine how we l l a PLC modem w i l l work i n any given s i t u a t i o n . One approach to the t e s t i n g and comparison of PLC modems would be to construct a communications channel with the transmission c h a r a c t e r i s t i c of a t y p i c a l power l i n e . A l l modems could then be tested on th i s channel and compared against each other. Unfortunate-l y , a sui t a b l e channel model i s not a v a i l a b l e . An a l t e r n a t i v e t e s t proce-dure, which would serve well to define the general performance c h a r a c t e r i s -t i c s of the PLC modem, would involve actual operation on representative power l i n e c i r c u i t s . In t h i s regard, the spread spectrum PLC modem was exercised i n three separate surroundings. F i r s t , the modem was tested on a communications chan-n e l which had a constant impedance, constant attenuation, and a d d i t i v e white Gaussian noise. Although t h i s channel i s not t y p i c a l of actual power l i n e transmission c h a r a c t e r i s t i c s , t h i s t e s t does f a c i l i t a t e a comparison of the actual modem performance against the t h e o r e t i c a l l y predicted performance. Any s i g n i f i c a n t differences between the two would i n d i c a t e that the modem was operating improperly. Second, the modem was operated i n a large multi-use b u i l d i n g where a wide v a r i e t y of loads were present, Including i n d u s t r i a l machinery, computers and computer terminals, and o f f i c e equipment. This environment approximated that of i n d u s t r i a l and o f f i c e b u i l d i n g s . 117 In the third set of experiments, the modem was tested in an apartment complex of approximately thirty individual apartments. An apartment building this size provides a typical residential environment - with more el e c t r i c a l loads than a single family dwelling but fewer than a large high rise apart-ment complex. The results of these tests are presented i n Sections 5.2, 5.3, and 5.4. Section 5.5 summarizes and discusses the results. 5.2 Performance in a Controlled Environment Figure 5.1 shows the equipment used for tests with additive white noise. To test the synchronization procedure both the transmitter and receiver are synchronized by the power line zero crossings. However, the PLC signal is sent over a separate channel. The channel in this case consists of a load resistor, which roughly approximates the power system impedance, and an additive white noise source. In addition to the channel, the transmitter, and the receiver, other equipment i s necessary for the test. A data source is needed to generate pseudorandom data bits (T, ) at the rate of 60 bits/sec. (In other words, data one bit every power line cycle.) The data error counter compares the receiv-ed data bits (R <j a t a) against the transmitted bits and increments an error counter whenever there i s a difference. Because data errors occur randomly, the error rate is d i f f i c u l t to measure precisely. For example, a string of 10** data bits may have ten errors within the f i r s t one hundred bits sent. If the measurement i s termi-nated after one hundred bits the error rate would be 10"*1. If, however, the 118 P O W E R . S Y N C H TRANSMITTER! 'DATA D A T A S O U R C E T R U E R M S VOLT M E T E R DATA C H A N N E L P O W E R . S Y N C H [WHITE NOISE GENERATOR LOAD I *w—1 R E S I S T O R 6 ± RECEIVER *OATA DATA ERROR COUNTER Figure 5.1 The white noise t e s t . 119 measurement i s made over 10** b i t s then the error rate would be 10" 3 . For t h i s reason i t Is important to allow the modem tests to run f o r a long period of time so that the error rate can be measured accurately. In the case of the PLC modem a l l t e s t s were run u n t i l close to 100 errors were accumulated to ensure an accurate P( SL) reading. The true RMS voltmeter i s used to measure the power of the PLC s i g n a l and the noise which are present at the rec e i v e r . The t h e o r e t i c a l l y predicted performance of an integrate-and-dump detec-tor operating i n an addit i v e white noise environment i s PU) = i- x e r f c JE^/NQ (5.1) where P(l) i s the p r o b a b i l i t y of a received b i t being detected i n c o r r e c t l y , e r f c ( ) i s the erro r function complement, E^ Is the energy i n each data b i t as measured at the rece i v e r , and N q i s the noise power s p e c t r a l density as measured at the receiver [62]. The p r o b a b i l i t y of error i s measured by the data e r r o r counter and i s the r a t i o of the number of b i t s received i n erro r r e l a t i v e to the t o t a l num-ber of b i t s sent. The received b i t energy Is given by where v D i s the RMS voltage with the noise generator o f f and T. i s the data b i t ' s duration. The b i t duration (T^) i n (5.2) needs further explanation. In a d i r e c t sequency spread spectrum modem, each b i t i s m u l t i p l i e d by the spreading code. Thus each b i t consists of a large number of code chips which have chip dura-tions (T ) much shorter than the o r i g i n a l data b i t (see Figure 3.A). How-c 120 ever, the chip duration and the bit duration are related to the spread spec-trum processing gain (G^) T f b = c (5.3) Therefore, from (5.2) and (5.3) the energy per chip (E £) i s : E = (v n ) 2/f = E./G (5.4) W C b p Equation (5.4) indicates that either the bit energy (E^), or the chip energy (E ) multiplied by the spread spectrum processing gain (G ), can be used i n c P equation (5.1). The noise power spectral density i s N Q = (v N ) 2 / £ R (5.5) RMS where v„ i s the RMS voltage measured with the transmitter off and f_ i s RMS R the receiver's noise bandwidth. The receiver's noise bandwidth (f_) i s determined by the line coupling network. The noise bandwidth was measured by applying white noise to the receiver and observing the signal, after i t has passed through the coupling network, on a spectrum analyzer. Figure 5.2 is a spectrum analyzer photograph of this signal and hence the receiver's noise bandwidth. As shown in Figure 5.2, the half power (3 dB) bandwidth is about 300 KHz. Table 5.1 i s a l i s t of the data taken i n the white noise environment. Figure 5.3 is a graph of this data along with the predicted performance as given by (5.1). Note that there i s close agreement between the measured and predicted performance of the modem. This agreement indicates the modem is operating properly. V e r t i c a l - 10 dB/cm. Horizontal - 100 KHz/cm. Figure 5.2 The receiver's noise bandwidth. 122 P U ) 10 x L O G 1 0 ( E b / N Q ) Figure 5.3 Predicted and measured performance. 123 (volts-KMb) VNOISE (volts-RMS) P( £) .0875 .0875 .0875 .0875 .0875 .0875 .0875 .0875 5.1 4.1 3.6 3.25 3.05 2.8 2.55 2.25 730 xlO"1* 287 xlO"1* 85 xl0_,t 48x10" 4 285x10" 5 162x10" 5 85x10" 5 26x10" 5 TABLE 5.1 Measured performance i n a white noise environment. 5.3 Performance i n a Large Multi-Use Building Figure 5.4 shows the equipment used f o r the t e s t s which were c a r r i e d out i n the Hector MacLeod Building - a large multi-use b u i l d i n g on the U n i v e r s i t y of B r i t i s h Columbia campus. Synchronization i s again provided by the power l i n e . In t h i s case, however, the receiver accepts not only the PLC s i g n a l generated by the transmitter but also the background noise which i s present on the power l i n e . In a d d i t i o n , both the transmitter and the receiver are connected across the power l i n e impedance, which i s both time and frequency dependent. The measurements were taken over a 30-hour period on Sunday, November 11 and Monday, November 12, 1981. During t h i s time the background noise l e v e l , as measured at the r e c e i v e r , remained constant. There was, however, a period of two hours, between 2:00 p.m. and 4:00 p.m., Sunday afternoon, when many voltage transients occurred, presumably from loads switching on and o f f . During t h i s period the error rates increased by more than a fac t o r of 10. Figure 5.5 i s a plot of the data. Shown i n Figure 5.5 are curves which show the best, worst, average, and white noise performances of the modem. 124 DATA GENERATOR 'DATA TRANSMITTER PLC COUPLING NETWORK J 4 POWER LINE DATA ERROR COUNTER TRUE RMS VOLTMETER DATA RECEIVER 1 ? RPLC*RNOISE COUPLING NETWORK Figure 5.4 The equipment used f o r te s t s i n a large multi-use b u i l d i n g . 126 5.4 Performance i n a Small Residential Apartment Complex The equipment used in this test was identical to that shown in Figure 5.4. The test was conducted in a small residential apartment complex, located in a residential d i s t r i c t of Vancouver, B.C., containing approximate-ly thirty apartments. The measurements were made over a 24-hour period on Tuesday, November 12, and Wednesday, November 13, 1981. In this case the background noise level varied by about 10 dB over the 24-hour period. The highest noise level occurred in the evening around 11:00 p.m. while the lowest noise level was observed in the early morning at about 4:00 a.m.. Figure 5.6 is a plot of the measurements taken. Included in Figure 5.6 are four curves showing the best, worst, average, and white noise performance of the modem. 5.5 Summary of Results Observations made during the modem tests indicate that most of the errors occurred as a result of impulse noise as shown in Figure 2.9. Impulse noise presents a problem because the duration of the impulse is less than the duration of the code chips. For that reason, the impulses w i l l pass almost unaffected through the data despreader as shown in Figure 5.7. After passing through the data despreader the impulses w i l l cause step changes in the inte-grator's output which may cause i t to cross over the comparator's threshold resulting i n a bit error. From Figures 5.5 and 5.6 i t i s apparent that there is a large variation in error performance for the same SNR, especially at low power levels. This is a result of the impulse noise. The impulses result from the switching of Figure 5.6 Performance i n a small apartment complex. 1 2 8 INTEGRATOR OUTPUT •~ «• ! T i n J ~ l J i n _ r COMPARATOR THRESHOLD Figure 5.7 E f f e c t of the data despreader on impulse noise. 129 e l e c t r i c a l loads and therefore occur randomly. For a constant received s i g n a l power the error performance w i l l be l a r g e l y determined, at l e a s t f o r low signal power l e v e l s , by the number of impulses which occur during the t e s t . Figures 5.5 and 5.6 also i n d i c a t e that the erro r performance becomes more consistent when the SNR i s high. This improved consistency occurs because the noise impulses w i l l have less energy than the received s i g n a l . At high power l e v e l s , the output of the integrator w i l l be higher and the step changes which r e s u l t from the impulse noise w i l l be less l i k e l y to cause a decoding e r r o r . Instead, most errors w i l l occur from the background noise which i s r e l a t i v e l y constant. I t also became clear during the tests that the performance of the modem varies widely over a 24-hour day, because background e l e c t r i c a l i n t e r f e r e n c e , which a f f e c t s the modem's performance varies with the time of day. The modem had the worst error performance i n the evening, from 6:00 p.m. to 11:00 p.m., when e l e c t r i c a l noise l e v e l s were highest. Conversely, the modem worked best i n the early morning hours from 3:00 a.m. to 6:00 a.m. when there was l i t t l e e l e c t r i c a l noise. The differe n c e between the best and worst performance over two orders of magnitude f o r a constant SNR or, equivalently, over 5 dB i n SNR for the same error r a t e . When the test r e s u l t s are averaged the modem per-forms as though i t were i n a white noise environment with a performance l o s s of approximately 3 dB. From Figures 5.5 and 5.6 a 12 dB SNR i s required f o r an error rate of better than one i n 10,000 (10 - 1*). Using the peak noise l e v e l s observed during tests the plus 12 dB SNR corresponds to a received s i g n a l power of 130 about one quarter of a milliwatt (-6 dBm). Therefore, one quarter of a milliwatt or more must be delivered to the receiver to obtain acceptable error performance. A typical PLC transmitter would deliver about 1 watt (+30 dBm) of power so that transmission lines losses between the transmitter and receiver must be kept below 36 dB for reliable communications. As a compari-son, measurements made during the performance tests show an attenuation of 6 dB between the transmitter and receiver when they are both on the same side of the distribution transformer. The losses increased to 16 dB when the transmitter and receiver were on opposite sides of the transformer. The results summarized in this section are based on measurements made on some typical power lines. A spread spectrum PLC modem should have a more consistent performance than a PLC modem using narrowband signalling. This i s because many of the transmission impairments of the power system affect only a narrow frequency band and therefore w i l l not significantly degrade the wideband SS signal. However, additional tests carried out on a wide variety of power lines w i l l be necessary to f u l l y assess the SS PLC modem's perform-ance, 131 6. CONCLUSIONS 6.1 Summary This thesis has documented the design and testing of a spread spectrum modem for communication over e l e c t r i c a l power lines. The ava i l a b i l i t y of a power line modem would allow many communications services within buildings, within c i t i e s , and between c i t i e s to be implemented inexpensively because the communications channel would be available without cost. In spite of i t s advantages, power line communications has not enjoyed widespread use. this is partially because there i s not yet a concensus on the best signalling format for power line communications channels. It i s apparent from examining the transmission characteristics of power lines that many impairments affect only a narrow range of frequencies. For this reason a signalling format which uses a broad range of frequencies should perform better and more consistently than a signalling format which uses a narrow frequency range. Spread spectrum modulation provides such a broadband signalling format. During the course of this work one other author [38] has suggested that spread spectrum communications should work well on power lines. However, no Implementation details or performance results were published. In order to test the performance of spread spectrum modulation on power lines a transmitter and receiver were designed and b u i l t . As a result of this design a novel synchronization procedure based on the 60 Hz power signal, was developed. This synchronization procedure significantly reduced the complexity of the modem. In addition, explicit relationships between the modem's processing gain and the synchronization accuracy were estab-l i s h e d . The modem was f i r s t tested i n a con t r o l l e d environment, i n order that i t s actual performance could be compared to that t h e o r e t i c a l l y predicted. The two re s u l t s were very close which indicates that the synchronization procedure and other modem c i r c u i t s operate properly. Further t e s t s were car r i e d out i n a large multi-use b u i l d i n g and a r e s i d e n t i a l apartment com-plex. The r e s u l t s of these tests Indicate that the modem performs well on actual power l i n e c i r c u i t s . The modem's performance varied s u b s t a n t i a l l y with the time of day as a r e s u l t of a 10 dB v a r i a t i o n In background noise l e v e l s during a twenty-four hour period. I t was established that the modem requires a received s i g n a l to noise r a t i o of 12 dB to d e l i v e r a b i t error rate below 10 _ 1*. Typ i c a l background noise l e v e l s and t y p i c a l transmitted power Indicate that the modem's performance w i l l be adequate with channel losses as high as 36 dB. 6 . 2 Cost of the Spread Spectrum Power Line Modem A spread spectrum modem i s more complex and hence more c o s t l y that a narrowband modem. However, for power l i n e communications the modem's syn-chronization can be based on the 60 Hz power s i g n a l . In t h i s case complexity and cost are s i g n i f i c a n t l y reduced and only a small cost penalty Is paid f o r the increased performance. Because the implementation i s not f i n a l i z e d , the modem's cost i s d i f f i c u l t y to estimate. Parts costs f o r the transmitter and receiver were about t h i r t y d o l l a r s each. However, i f power l i n e communications becomes 133 widespread volume discounts and custom integrated c i r c u i t s would s i g n i f i c a n t -l y reduce these c o s t s . 6.3 Suggestions f o r Future Work Observations made during the modem tests i n d i c a t e that impulse noise on the power l i n e i s a major source of b i t e r r o r s , e s p e c i a l l y at low s i g n a l to noise r a t i o s . Impulse noise i s also a problem i n other communications receivers such as FM demodulators. As a r e s u l t impulse f i l t e r s have been developed. Further work could investigate the e f f e c t of an impulse f i l t e r placed ahead of the spread spectrum r e c e i v e r . Phase noise on the 60 Hz power s i g n a l l i m i t s the maximum processing gain of the modem. Additional work on phase averaging the power s i g n a l could reduce t h i s phase noise, improve the processing gain, and hence the modem's performance. Although the transmission c h a r a c t e r i s t i c s of the high voltatge and d i s t r i b u t i o n networks have been studied, very l i t t l e work has been done on i n t r a b u i l d i n g wiring. Future work could examine transmission c h a r a c t e r i s t i c s of i n t r a b u i l d i n g wiring e s p e c i a l l y at high frequencies. Standard t e s t procedures would be us e f u l i n comparing the performance of d i f f e r e n t PLC modems. Work could be done to develop a simulation of the transmission c h a r a c t e r i s t i c s of the power system. This would allow modems to be tested i n i d e n t i c a l environments by operating on i d e n t i c a l simulated power l i n e s . 134 REFERENCES [1] B.D. Rusell, "Communication alternatives for distribution metering and load management", IEEE Trans. Pwr. 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