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UBC Theses and Dissertations

Decoupling capacitor design issues in 90nm CMOS 2006

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D E C O U P L I N G C A P A C I T O R D E S I G N I S S U E S IN 9 0 N M C M O S by X I O N G F E I M E N G B . A . S c , The University of British Co lumb ia , 2004 A T H E S I S S U B M I T T E D IN P A R T I A L F U L F I L M E N T O F T H E R E Q U I R E M E N T S F O R T H E D E G R E E O F M A S T E R O F A P P L I E D S C I E N C E in T H E F A C U L T Y O F G R A D U A T E S T U D I E S ( E L E C T R I C A L A N D C O M P U T E R E N G I N E E R I N G ) The University of British Co lumb ia Apri l 2006 ©X iong fe i Meng , 2006 ABSTRACT On-chip decoupling capacitors (decaps) are widely used to reduce power supply noise. Typically, designs use NMOS decaps between standard-cell blocks and NMOS+PMOS decaps within the blocks. Starting at the 90nm CMOS technology node, the traditional decap designs may no longer be suitable due to increased concerns regarding thin-oxide gate leakage and electrostatic discharge (ESD) reliability. This thesis investigates new decap design approaches that address gate leakage and ESD. A cross-coupled design is described that has been recently introduced by cell library developers to handle ESD problems. Three modifications of the cross-coupled design are introduced here and the tradeoffs among transient response, gate leakage and ESD performance are analyzed. The modifications offer designers greater flexibility in decoupling capacitor design for 90nm and below. To improve the power-grid noise reduction capability in the areas between blocks, two versions of a switched-decap design are proposed. One provides excellent decap performance but consumes large power, whereas the other saves power but suffers from excessive delay. A novel low-power voltage regulator using switched decaps is proposed to better balance performance and power consumption. 11 T A B L E OF CONTENTS ABSTRACT II TABLE OF CONTENTS Ill LIST OF TABLES V LIST OF FIGURES VI ACKNOWLEDGMENTS VIII CHAPTER 1 INTRODUCTION 1 1.1 MOTIVATION 1 1.2 DECOUPLING CAPACITORS ISSUES AT 90NM 5 1.3 RESEARCH OBJECTIVES 7 1.4 ORGANIZATION OF T H E THESIS 8 CHAPTER 2 BACKGROUND 10 2.1 INTRODUCTION 10 2.2 DECOUPLING CAPACITOR MODELING 11 2.3 G A T E TUNNELING L E A K A G E 12 2.4 ELECTROSTATIC DISCHARGE RELIABILITY IN DECAP DESIGN 15 2.5 STANDARD-CELL DECAP L A Y O U T AND P L A C E M E N T 17 CHAPTER 3 EXISTING DECOUPLING CAPACITOR DESIGN APPROACHES 22 3.1 INTRODUCTION 22 3.2 CROSS-COUPLED DECAP 22 3.3 G A T E D DECAP 25 3.4 THICK OXIDE DECAP 28 3.5 HIGH-K G A T E DIELECTRIC 30 3.6 METAL-INSULATOR-METAL CAPACITOR 32 3.7 SUMMARY 33 CHAPTER 4 PASSIVE DECOUPLING CAPACITOR DESIGNS 34 iii 4 . 1 I N T R O D U C T I O N 3 4 4 . 2 RC M O D E L I N G O F B A S I C C R O S S - C O U P L E D D E C A P D E S I G N 3 4 4 . 3 T R A N S I E N T R E S P O N S E S I M U L A T I O N 3 6 4 . 4 E S D P E R F O R M A N C E S I M U L A T I O N 4 2 4 . 5 G A T E L E A K A G E S I M U L A T I O N 4 4 4 . 6 M O D I F I E D C R O S S - C O U P L E D D E C A P D E S I G N S 4 5 CHAPTER 5 A C T I V E D E C O U P L I N G C A P A C I T O R D E S I G N S 5 1 5 . 1 I N T R O D U C T I O N 5 1 5 . 2 S W I T C H E D D E C O U P L I N G C A P A C I T O R 5 2 5 . 3 S U N ' S V O L T A G E R E G U L A T O R 5 5 5 . 4 F U J I T S U ' S A C T I V E P O W E R S T A B I L I Z E R 5 9 5 . 5 L O W - P O W E R V O L T A G E R E G U L A T O R 6 3 5 . 6 S I M U L A T I O N S E T U P A N D R E S U L T S 6 7 CHAPTER 6 C O N C L U S I O N S A N D F U T U R E W O R K 7 4 6 . 1 S U M M A R Y 7 4 6 . 2 C O N T R I B U T I O N S I N T H I S T H E S I S 7 6 6 . 3 F U T U R E W O R K 7 6 R E F E R E N C E S 7 8 i v LIST OF TABLES T A B L E 4.1: COMPARISON ON E S D PERFORMANCE, TRANSIENT RESPONSE A N D G A T E L E A K A G E . ... 47 T A B L E 5.1: N O D E BIASING A N D SWING FOR S U N V O L T A G E REGULATOR [41] 56 T A B L E 5.2: N O D E BIASING A N D SWING FOR FUJITSU ACTIVE POWER STABILIZER [42] 61 T A B L E 5.3: N O D E BIASING A N D SWING FOR LOW-POWER V O L T A G E REGULATOR 64 v LIST OF FIGURES F IGURE 1.1: T W O COMPONENTS OF POWER SUPPLY NOISE [3] 2 F IGURE 1.2: U S E OF DECOUPLING CAPACITOR TO REDUCE POWER GRID NOISE [3] 4 F IGURE 1.3: DECOUPLING CAPACITOR IMPLEMENTED USING A N N M O S DEVICE 5 F IGURE 2.1: D E C A P MODELING AS A SERIES RC CIRCUIT 11 FIGURE 2.2: G A T E L E A K A G E CURRENT VERSUS GATE A R E A 13 FIGURE 2.3: G A T E L E A K A G E CURRENT DENSITY VERSUS OXIDE THICKNESS 14 FIGURE 2.4: C O M P L E T E E S D PROTECTION SCHEME 16 FIGURE 2.5: STANDARD C E L L N + P DECAP CONFIGURATION 17 F IGURE 2.6: S A M P L E L A Y O U T OF S T A N D A R D - C E L L N+P DECAP WITH N O FINGERS 18 FIGURE 2.7: S A M P L E L A Y O U T OF S T A N D A R D - C E L L N+P DECAP WITH TWO FINGERS 18 FIGURE 3.1: CROSS-COUPLED DECAP SCHEMATIC [11] 23 F IGURE 3.2: INTUITIVE UNDERSTANDING OF INPUT RESISTANCE IN CROSS-COUPLED DESIGN 24 F IGURE 3.3: BASIC G A T E D DECAP SCHEMATIC [12] A N D GATE L E A K A G E FLOW 25 F IGURE 3.4: S A M P L E C L O C K FOR T H E C T R L SIGNAL IN G A T E D DECAP 27 FIGURE 3.5: INSERTION OF S M A L L INVERTER IN G A T E D DECAP [12] 27 FIGURE 3.6: STACK OF THIN-OXIDE DECAPS VERSUS THICK-OXIDE DECAP 29 FIGURE 4.1: CROSS-COUPLED DECAP SCHEMATIC [11] A N D MODELING 35 F IGURE 4.2: SCHEMATIC FOR THE FIRST TRANSIENT SETUP 37 F IGURE 4.3: RC MODELING OF THE FIRST TRANSIENT SETUP 37 F IGURE 4.4: SIMPLIFICATION OF RC MODELING OF THE FIRST TRANSIENT SETUP 38 F IGURE 4.5: TRANSIENT RESPONSE FOR THE FIRST SETUP 40 FIGURE 4.6: SCHEMATIC FOR THE SECOND TRANSIENT SETUP 41 vi F IGURE 4.7: TRANSIENT RESPONSE FOR THE SECOND SETUP 42 F IGURE 4.8: S IMULATION SETUP FOR E S D ANALYSIS [10] 43 FIGURE 4.9: S A M P L E L A Y O U T OF M O D I (BASIC CIRCUIT WITHOUT FINGERING) 45 F IGURE 4.10: S A M P L E L A Y O U T OF MOD2 (REPLACE N M O S WITH P M O S ) 46 F IGURE 4.11: S A M P L E L A Y O U T OF MOD3 (REPLACE N M O S WITH P M O S , A N D USE S M A L L E S T N M O S ) 47 F IGURE 4.12: C O M P L E T E TRANSIENT RESPONSE FOR THE FIRST SETUP 48 F IGURE 4.13: C O M P L E T E TRANSIENT RESPONSE FOR THE SECOND SETUP 49 F IGURE 5.1: PRINCIPLE OF SWITCHED DECOUPLING CAPACITOR [40] 52 FIGURE 5.2: M O S IMPLEMENTED SWITCHED DECOUPLING CAPACITOR [40]-[42] 54 F IGURE 5.3: B L O C K D I A G R A M OF S U N V O L T A G E REGULATOR [41] 55 F IGURE 5.4: CIRCUIT IMPLEMENTATION OF S U N V O L T A G E REGULATOR [40] [41] 57 FIGURE 5.5: CIRCUIT IMPLEMENTATION OF FUJITSU ACTIVE POWER STABILIZER [42] 60 FIGURE 5.6: CIRCUIT IMPLEMENTATION OF LOW-POWER V O L T A G E REGULATOR 65 FIGURE 5.7: S IMULATION SETUP FOR ACTIVE DECAP CIRCUITS 69 F IGURE 5.8: S IMULATION RESULTS FOR N O DECAP INSERTED 70 FIGURE 5.9: SIMULATION RESULTS FOR STANDARD DECAPS INSERTED 70 F IGURE 5.10: S IMULATION RESULTS FOR S U N V R INSERTED 71 F IGURE 5.11: S IMULATION RESULTS FOR FUJITSU A P S INSERTED 71 FIGURE 5.12: S IMULATION RESULTS FOR LOW-POWER V R INSERTED 72 vii ACKNOWLEDGMENTS I would like to express my gratitude to my academic supervisor, Dr. Resve Saleh, whose expertise, understanding, patience, and support, added significantly to my graduate experience. I appreciate his profound and vast knowledge in many areas, both within and outside of the scope of my research. I would like to thank my industrial supervisor, Dr. Karim Arabi, for his technical advice, encouragement, and kindness. A very special thanks to my colleagues and friends in the SoC group, without whose assistance, I would not have accomplished this research. More specifically, I would like to thank Amit Kedia, Dipanjan Sengupta, Karim Allidina, Reza Molavi, Scott Chin, Roberto Rosales, Roozbeh Mehribadi, and Dr. Steve Wilton. I also thank Dr. Shahriar Mirabbasi who provided me with technical and personal support whenever I asked and became more of a mentor and friend, than a professor. I also acknowledge Asad Shayan and Mamta Bansal of PMC-Sierra Inc. for their suggestions and help in this study. I recognize that this research would not have been possible without the financial support from NSERC and PMC-Sierra Inc., and express my gratitude to them. I also thank C M C Microsystems for providing the C A D tools. Last, but not least, I would especially like to thank my family for both giving and encouraging me to seek for myself a demanding and meaningful education. In particular, I must acknowledge my fiancee, Liming, for her love, caring and patience through many years of my life. The appreciation extends beyond any words at my command. vi i i Chapter 1 Introduction 1.1 Motivation As integrated circuit (IC) technology scales, more and more transistors are being placed within a single chip, while the clock frequency continues to increase into the gigahertz range. The result is that large transient currents are drawn from the power supply rails in just a few hundred picoseconds [1] in modern custom and application-specific integrated circuit (ASIC) chips. Meanwhile, the supply voltage is scaled with technology to reduce overall power consumption, and as a consequence, the circuitry becomes more prone to power-supply noise. The management and regulation of the quality of the on-chip power supply is a major challenge [1]. The power grid, which provides V D D and Vss (or ground) signals throughout the chip, experiences fluctuations in value due to a variety of noise sources. If the supply voltage noise or variation is excessively large, it may lead to problems such as delay variation, timing unpredictability, or even improper functionality [2]. A commonly used metric, noise budget, is defined as the maximum allowable noise amplitude [3]. Typically, it is required to keep the power supply noise within a certain percentage (e.g., 10%) of the nominal supply voltage VDD- 1 Namely, 10% of VDD - Vss is typically the noise budget, a rule-of-thumb used in the industry [4]. Circuit designers must ensure that the chip operates correctly i f the maximum voltage difference between VDD and Vss is 10% or smaller than the nominal value. In today's advanced deep-submicron (DSM) technology, the power grid noise is due to two main issues: (1) As the power lines made of metal wires become thinner, the wire resistance R increases. When logic gates switch and a current / flow through the power lines to deliver charge to the gates, the voltage drop A V at the gates is A V i = I R. This type of power-supply noise is known as IR drop. (2) Due to package pin inductance and thin-interconnect inductance, the power lines experience inductance effect when the current flow changes with respect to time. This second source of voltage drop is given by A V 2 -L—. The two power supply noise dt components are illustrated in Figure 1.1, which depicts two inverters connected to an off-chip voltage supply through the on-chip power grid. V D D 0 Off-Chip AAAA rrrrs V V \ A I = V D D - A V /j Mr 4= On-Chip AAAA rrrr\ VNAA- 0 Figure 1.1: Two components of power supply noise [3]. Considering the two components together, the overall voltage drop A V at any point in the power grid is: 2 AV = IR + L— (1.1) dt To illustrate the IR drop, all the nodes in the power grid are initially charged to VDD with no activity in the circuit. As the second inverter starts to switch, the wire resistance along the VDD line creates voltage drops as current flows from the external voltage source towards the second inverter [3]. Similarly, the ground grid is subject to the same type of problem when the outputs of the buffers switch low, except that the voltage level of the ground line will increase. This is sometimes referred to as ground bounce. In practice, IR drop can be caused by simultaneous switching of clock buffers, bus drivers, memory decoder drivers, and so on, when there is high activity in the circuit. These simultaneous switching activities can happen anywhere on the chip. Thus, all regions in the chip are susceptible to IR drop. In a wire-bond (e.g., dual-inline) package, the supply voltage level remains relatively high at the periphery of chip where the voltage supply I/O pads are located, and drops noticeably at the centre of the chip. In contrast, in a flip-chip (or ball-grid array) package, the centre of the die has rather high voltage level, whereas the periphery of the die experiences larger IR drops. Considering the Ldi/dt term in Equation (1.1), the inductance L is another source of voltage drop in the power supply and is typically at 1 to 2 nH in a dual-inline package (DIP) or at roughly 0.1- 0.2nH in a ball-grid array (BGA) package [3]. In a traditional DIP package, this inductance arises from the bonding wire used to connect the chip I/O pads to the lead pins. On the other hand, in a modern ceramic B G A package, the inductance comes from the solder bumps that can be placed anywhere in the chip area [5]. Although B G A is a more expensive solution, it provides more I/O connection capability and less inductance value [3]. 3 In the past, compared to IR drop, the Ldi/dt term was not considered as a significant source of power supply noise, mainly because the chip clocks were not running at the gigahertz range. However, in today's chips, this inductance effect is a much more significant [6]. The value of L has not changed considerably over the years, while the value of di/dt has continued to increase due to faster and faster clock frequencies. A common technique for reducing power supply noise and keeping the noise within the noise budget is through the use of on-chip decoupling capacitors (decaps). Decaps are essentially capacitors that hold a reservoir of charge and are placed close to the power pads and near any large drivers. When large drivers switch, the decaps provide instantaneous current to the drivers to reduce IR drop and Ldi/dt effects, and hence keep the supply voltage relatively constant. As shown in Figure 1.2, the on-chip decap delivers current to charge up the load capacitance of the second inverter when it switches. The supply voltage level is relatively constant at the inverter tap point since the decap is nearby, so A V is minimal. V D D JrYYY\. 0 Off-Chip AAAA—rrrr\—v\AA H On-Chip Decap •AAAA—rmr\—v\AA Figure 1.2: Use of decoupling capacitor to reduce power grid noise [3]. This thesis focuses on the issues of power supply noise reduction through the use of decoupling capacitors. In a typical ASIC design, decaps can be placed in the open areas of the chip between 4 intellectual property (IP) blocks (called white-space decaps, or global decaps) and within the IP blocks composed of standard cells [7]. The thesis discusses both types of on-chip decaps, although off-chip decaps are commonly used as well [8]. 1.2 Decoupling Capacitors Issues at 90nm A standard decap is usually made from NMOS transistors in a CMOS process [3]. As shown in Figure 1.3, the gate of the NMOS transistor is connected to V D D , whereas source, drain and substrate of the transistor are tied to Vss- This approach is considered effective because the thin- oxide capacitance of the transistor gate provides a higher capacitance than any other oxide capacitance available in a standard CMOS fabrication process [4]. For this MOS decap, the first- order calculation of the capacitance is WLCox, where W is the transistor width, L is the transistor length, and Cox is the oxide capacitance per unit area. Accurate capacitance model needs to include the parasitic fringing and overlap capacitance of the transistor, and will be discussed in greater detail in Chapter 2. At the 90nm technology node, the oxide thickness of a transistor is reduced to roughly 2.0nm or less. The thin oxide causes two new problems: possible electrostatic discharge (ESD) induced oxide breakdown and gate tunneling leakage [9], [10]. Potential ESD oxide breakdown increases 5 V D D Decap the likelihood that an IC will be permanently damaged during an ESD event, and hence raises a reliability concern. Higher gate tunneling leakage increases the total static power consumption of the chip. As technology scales further down, with a thinner oxide, the result is an even higher ESD risk and more gate leakage. The standard decap design using NMOS transistors experiences these two problems and therefore becomes rather inappropriate for 90nm and below. While satisfying ESD reliability and gate leakage limitations, decap designs must also meet the transient performance requirements. Since a 90nm process (or below) usually provides the capability of running at gigahertz frequencies, the decap must respond in the order of a hundred picoseconds. A new cross-coupled standard-cell design approach [11] addresses the issue of ESD performance. The design provides a certain amount of ESD input protection to the decap, but does not offer any savings in gate leakage. Even worse, the new design experiences a much degraded transient response, making it somewhat unsuitable for high-speed chips. Modifications to the cross-coupled decap design that properly trades off ESD, transient response and gate leakage are needed. Another new global decap design approach, called gated decap, has been reported [12] to control gate leakage. Based on the approach of multi-threshold CMOS circuit, the gated decap is capable of saving a significant amount of leakage current while in power-saving mode. However, the design suffers from an oscillation problem. The lack of robustness of the design makes it somewhat less attractive for industrial use. 6 Most fabrication processes provide high-voltage thick-oxide I/O transistors. Those transistors have properties of excellent ESD reliability and almost-zero gate leakage. These desired properties make them good candidates for global decap implementation in 90nm or below. However, the effective capacitance for such thick-oxide decaps is much less than thin-oxide ones. Certain fabrication limitations also apply for thick-oxide devices. From a process perspective, the use of high-k gate dielectrics is an active field. Progress has been made to provide savings in gate leakage. Nevertheless, many process issues still exist and the high-k technology is far from mature enough for mass production. Metal-insulator-metal capacitors are available in many fabrication processes. It is useful to examine i f those capacitors are suitable for making global decaps since it is known that the leakage current for such a capacitor is low. For global decaps, to improve the area efficiency, the use of switched decoupling capacitors is an interesting alternative. Compared to the passive designs, the intent of the switched decaps is to boost the supply voltage to provide better power-grid noise reduction capability. Two existing designs from Sun and Fujitsu are worth investigating, and improving for low-power operation. Such a low-power voltage regulator would be suitable for general low-power applications in both ASIC and custom designs. 1.3 Research Objectives The research objectives of this work are as follows: 7 • Understand relationship between critical decap design issues, such as electrostatic discharge reliability, gate leakage and transient response. • Develop passive solutions to decap designs that properly trade off gate leakage, ESD and transient response, and provide designers with design flexibility for sub-90nm technologies for standard-cell decaps. • Explore active solutions to decap designs that provide better power-grid noise reduction capability than the passive approaches. Design a novel switched-decap voltage regulator that properly balances power dissipation and decap performance in white-space decaps. 1.4 Organization of the Thesis The remainder of this thesis is organized as follows. Chapter 2 provides the necessary background for decap modeling, gate tunneling leakage phenomenon, ESD reliability, and standard-cell placement of decoupling capacitors. Chapter 3 explores various decap design approaches that may be suitable for upcoming technologies. The circuit-level designs are discussed first, followed by the process-level efforts. The pros and cons of each approach are provided. Chapter 4 develops a set of new designs based on the cross-coupled decap. The modeling of the new designs is described to allow hand calculations and analyses to be carried out. The new designs are validated using a full set of simulations in transient, ESD and gate leakage measurements. Based on the simulation results, the proper layout of these designs is described. Chapter 5 analyzes the advantages and disadvantages of Sun's voltage regulator and Fujitsu's active power stabilizer to improve decap area efficiency. A novel low-power switched-decap 8 voltage regulator is designed to achieve good power-grid noise reduction performance while maintaining a low level of dc power consumption. Chapter 6 summarizes the results of the thesis and provides conclusions. Future research directions are provided. 9 Chapter 2 Background 2.1 Introduction The topics in this chapter provide the necessary background for the rest of the thesis. Moreover, some fundamental and practical decap design issues are highlighted in this chapter to motivate the topics in the remainder of the thesis. This chapter begins with modeling of standard NMOS decaps. An overview of design challenges and problems associated with decoupling capacitors in 90nm and below is provided. The overview includes gate tunneling leakage, electrostatic discharge phenomenon and protection, and standard-cell decap placement. The gate leakage is introduced from a physical point of view, and useful information from recent technologies is given. ESD reliability is presented and typical phenomena during an ESD event are discussed. Primary and local ESD protection schemes are briefly illustrated. Since ASIC designs typically utilize standard cells, the decap insertion and placement procedure within standard-cell blocks is briefly introduced. 10 2.2 Decoupling Capacitor Modeling A standard decap is usually implemented using an NMOS transistor with the gate connected to VDD and both source and drain connected to Vss, or a PMOS device with opposite connections. When implemented using MOS transistors, decaps experience parasitic channel resistance that imposes certain delay on the transient response of the decap [4] [13]. Therefore, a decap should be modeled as a series connection of effective resistance and effective capacitance [4], as shown in Figure 2.1. VDD VDD - Vss — Vss Figure 2.1: Decap modeling as a series RC circuit. For precise calculation, the effective capacitance at low frequencies can be written as: Ceff=CoxWL + 2COLW (2.1) where Cox is the oxide capacitance per unit area, COL is the overlap and fringing capacitance per unit width, and W and L are the width and length of the transistor, respectively. The decap's effective resistance at low frequencies is given by [4]: R ff = (2-2) eff 6MCoxW(Vcs-VT) where ju is the channel mobility, VGS (or VQD since source and drain are tied) is the voltage across the oxide, and V T is the threshold voltage. From Equation (2.2), Re/f\s proportional to the 11 channel length L. That is, for faster transient response, a decap design should use a small L to keep Reff small. Both Reff and Ce/f can be considered constant at low or moderate operating frequencies, but they are degraded a high frequencies [4] [14]. Since decaps are usually built using MOS transistors, the high-frequency behavior of MOS transistors needs to be investigated. Previous work in [4] has shown that both Re/f and Ceff will decay as operating frequency increases. That is, both Reg- and Cejf are functions of frequency, / Although it is desired to have a small Rej, the decreased Ceff results in reduced capability of the decap at high frequencies. The channel length L of the decap controls its frequency response. If L is small enough, the effective capacitance remains relatively constant at high frequencies. As a consequence, a fingering technique in decap layout is commonly used to maintain its frequency response [4]. Moreover, NMOS has better frequency response than PMOS [4]. Thus, the use of PMOS decaps should be limited, from the frequency response perspective. 2.3 Gate Tunneling Leakage A new design issue for decaps due to oxide thickness reduction is the gate tunneling current. The current is in the form of tunneling electrons or holes from substrate to gate or from gate to substrate through the gate oxide, depending on the voltage biasing conditions [15]. Two forms of gate tunneling exist: Fowler-Nordheim (FN) tunneling and direct tunneling. For normal operations on short-channel devices, FN tunneling is negligible, and direct tunneling is dominant [15]. In the case of direct tunneling, the gate leakage current in PMOS is much less than in 12 NMOS, and it has been shown experimentally that PMOS gate leakage is roughly 3 times smaller than NMOS gate leakage for same size transistors [16] [17]. The gate leakage simulations can be carried out by using BSIM4 SPICE models [18] [19]. Assuming a 90nm technology with 2.0nm oxide thickness and 1.0V power supply, the gate leakage current is shown in Figure 2.2. gate a r e a Figure 2.2: Gate leakage current versus gate area. The gate leakage current density J and the oxide thickness tox have an empirical relationship as follows, assuming the voltage across the oxide Vox is fixed [16]: j = 1 0 ^ - f i " o , ) (2.3) where A and B are experimental constants and are process dependent. Equation (2.3) implies that the gate leakage current is exponentially related to the oxide thickness. A typical J and tox relationship for a fixed Vox is illustrated in Figure 2.3. 13 J (A/um2) f gate leakage current density 3xl0H<--X 1x10 -7 oxide thickness Figure 2.3: Gate leakage current density versus oxide thickness. It is evident that from 90nm technology onward, the gate leakage from decaps will be significant [17]. The gate leakage contributes to the total static power consumption, and decaps usually occupy a large on-chip area. The use of PMOS devices exclusively is not a viable solution for high-frequency circuits since they have a poor frequency response relative to the NMOS devices In addition, the amount of gate leakage is also a strong function of the applied bias [20]. If the transistor has a Vox that is roughly equal to VDD, the leakage current density is largest. If the transistor has a Vox set to close or below V r , it leaks significantly less. Indeed, under such a condition, the gate leakage current is typically 3-6 orders of magnitude less, depending on the values of VDD and tox [20]. Thus, the gate leakage in the second condition can be roughly considered to be zero. In decaps, the gate is at VDD and the source and drain of a transistor are tied together. Therefore, decaps would experience the highest levels of leakage. [4]- 14 2.4 Electrostatic Discharge Reliability in Decap Design Another new consideration has arisen in the form of ESD protection due to the thin oxide in 90nm technology. ESD is the process of static discharge that can typically arise from human contact with any IC pin. Approximately 0.6uC of charge is carried on a body capacitance of lOOpF, generating a potential of 2kV or higher to discharge from the contacted IC pin to ground for a duration of more than 100ns [10]. Under such an event, the peak discharge current is in the ampere range, leading to permanent damage on certain transistors in the chip if not properly protected. The damage can be in one of two forms, or a combination of the two: one is thermal burnout in devices or interconnects, while the other is oxide breakdown of devices due to the high voltage across the oxide [10]. When running simulations for an ESD event, the maximum current density Jmax of devices and interconnects is measured to check for potential thermal damage. The oxide voltage also needs to be measured to compare with the oxide breakdown voltage of a device for a given fabrication process. The oxide breakdown voltage is almost linearly proportional to the oxide thickness [10]. For instance, assuming a 90nm process uses 2.0nm of oxide thickness, the corresponding oxide breakdown voltage is just below 5V. If the thickness is doubled, the oxide breakdown voltage is also doubled to around 10V [10]. An ESD event can be delivered between any two pins of an IC. To properly protect an IC from ESD damage, an ESD circuit must shunt ESD current between these two pins [10]. In the case of decaps within standard cells, the only two pins that the decaps have access to are the two local power rails, namely V D D and Vss- Primary and local (sometimes called secondary) protection elements are needed to protect the two rails by limiting the voltage difference between the two rails to a value below the oxide breakdown voltage. The primary element will shunt most of the 15 ESD current, whereas the local element serves to limit the voltage or current at the local circuit until the primary element is fully operational [10]. A primary element can be a thick oxide transistor, a silicon-controlled rectifier, an open-gate, grounded-gate or coupled-gate NMOS transistor, or a large diode [10]. A local protection element can be simply a diode formed by a grounded-gate NMOS transistor [10]. A typical ESD protection scheme is illustrated in Figure 2.4. In addition to the primary and local elements, a resistor Rin is required to limit the maximum current flow to the decap and to limit the voltage seen from the gate of the decap. For better ESD protection, this resistance is normally large and can be in the forms of polysilicon, diffusion, n-well, or even channel resistance [10]. The resistance is generally not implemented together with primary and local protection devices. Rather, it is usually inserted within standard cells where ESD damage is a concern. X Global V D D Local V D D 1 > Primary Primary element element Secondary $ element D f i W i X Local Vss Global V s s Figure 2.4: Complete ESD protection scheme. Previous decap designs (before 90nm technology) did not consider ESD performance mainly because: 1. The transistor's oxide thickness was large and the oxide breakdown voltage was high enough so that the transistor was likely to survive during an ESD event with adequate protection circuits. 2. Insertion of the large resistance R,n dramatically reduces the transient response of the 16 decap. However, starting from 90nm, the gate oxide is so thin that the designer cannot ignore the increased ESD risk. A large resistance is therefore recommended to be placed inside the decap cells to protect from potential ESD damage. As a consequence, this tradeoff between ESD performance and transient response becomes the main decap design challenge in 90nm. 2.5 Standard-Cell Decap Layout and Placement In white spaces, decaps are usually made of NMOS devices, as described in the early sections. However, within standard cells, it is more convenient to make decaps using both types of NMOS and PMOS to form a decap filler cell, as shown in Figure 2.5. This is because the n-well is already implemented and usually reserved for PMOS devices. Only the lower half-cell area is for NMOS devices [4]. One sample standard-cell decap layout is illustrated in Figure 2.6. In the figure, the NMOS decap occupies roughly the bottom half of the cell area, whereas the PMOS decap is located in the n- well. The capacitor areas are the polysilicon gates placed on top of the channel regions of the MOS transistors. For standard cells, the height of the cell is always fixed, and the designers can only adjust the cell width. Once the cell width is determined, the size of the decap and the 17 capacitance of the decap are established. Figure 2.6 implies a large decap cell (measured in cell width) with long channel transistors. The decaps laid out in this long-channel fashion have poor performance at high frequencies, as discussed in Section 2.3. Therefore, a fingering technique is commonly used to improve the frequency response. Figure 2.7 depicts the same decap cell but with two fingers. Figure 2.7: Sample layout of standard-cell N+P decap with two fingers. 18 To model this N+P decap configuration, the overall impedance of two parallel RC circuits is determined as (Reff n H —)H(R e f f ^ — ) , and simplified as SCeff_n S(~-eff_p Reff_nReff_p 1 H 1- higher order terms. Reff_n + Reff_p S(Ceff_n + Ceff_p) For first-order hand calculations, the higher-order terms are negligible. Thus, the overall effective capacitance is the sum of the two individual decoupling capacitances, and the overall effective resistance is the parallel combination of the two individual effective resistances. That is: c eff_overail , / / C e f f _ P + CeJT _ Reff_ "Reff_r D eff _ overall ~Reff_ JIReff.P Reff_n (2.4) (2.5) During the placement procedure, computer-aided design (CAD) tools place standard cells into rows. Because the height of each cell is always the same, when cells are placed adjacent to each other, the n-well region and the V D D and Vss lines are automatically aligned. The cells for placement are obtained from the standard-cell library, where all the cells are predefined in width and driving strength. Since the total width of the row is fixed and the individual cell widths are fixed, some empty spaces (typically small) between the cells are left after placing cells. Those empty spaces are good candidates for the placement of decap cells due to its convenience [4]. In fact, a set of decap cells with different cell widths is also implemented in the standard-cell library. A l l the cells in the library must be designed for a specific process and meet all the design rules. Routing is typically carried out right after placement. 19 Decap insertion is considered as a part of the complete design flow. In a typical ASIC design flow, once the standard-cell blocks are synthesized, placed and routed by C A D tools, the decap cells are naturally placed into the empty spaces. Generally, since the spaces are filled using a library of decap cells with various sizes, the decap placement is done without affecting the placement of other logic cells. After placement and routing, chip-level timing is analyzed and timing violations will be fixed by replacement and/or rerouting. Then, chip-level voltage-drop analysis is carried out by some C A D tools (e.g., Apache™ Redhawk™) such that the hot spots of severe voltage-drop areas are identified. If the voltage drop at the hot spots exceeds the noise budget, more decaps will be inserted into the violation regions and a modification of the placement of other logic cells may have to be done. The logic cell movement requires additional timing and routability analysis before moving on to next step. Then, the chip voltage drop is analyzed again for the remaining hot spots. These steps in the design flow are iterated until all the hot spots are eliminated and all the logic circuits pass timing analysis. Typically, it may take 1 or 2 (occasionally even more) iterations to eliminate all the hot spots [3]. In addition, the potential problem of electromigration is also checked alongside the IR drop analysis [3]. This generally used decap placement approach is not optimal simply because the empty cells may not be located near the high voltage-drop regions. After the hot spots are first identified, the remaining empty spaces near the hot spots may not be largely enough. Hence, the logic cells may have to be shifted, resulting in additional timing analysis. In order to improve the placement efficiency, researchers suggest a few approaches including: global decap placement between standard-cell blocks [21], decap placement using activity [7], standard-cell decap placement not 20 affecting relative placement of logic cells [2], and earlier-stage decap placement decision [4]. Since decaps are experiencing excessive gate leakage, decap placement methods considering leakage current are proposed in [9] and [22]. 21 Chapter 3 Existing Decoupling Capacitor Design Approaches 3.1 Introduction Before 90nm technology, the use of M O S transistors as decoupling capacitors appeared to be a straightforward solution to the decap design problem. However, many factors such as excessive gate leakage, increased E S D risk, and consideration for high-performance transient response come to play important roles from 90nm technology. The consequence is that the standard M O S decap design may no longer be appropriate for the use in 90nm or below. Researchers have provided some new design approaches to address the design issues for decaps. This chapter provides an overview that sequences through all the major existing methodologies and identifies their advantages and disadvantages. 3.2 Cross-Coupled Decap Knowing that the standard N+P decap design for standard cells may no longer be suitable for 90nm technology due to increased E S D risk, a new cross-coupled decap design has been proposed [11] to address this issue. In the new cross coupled design (Figure 3.1), the drain of the P M O S connects to the gate of the N M O S , whereas the drain of the N M O S is tied to the gate of the P M O S [11]. 22 J Figure 3.1: Cross-coupled decap schematic [11]. From the layout perspective, this cross-coupled circuit can be seen simply as a terminal-swapped version of the standard decap. In other words, the decap transistor areas need not to be modified, while only the metal wire connections are modified. Thus, this new design does not require additional area in layout, compared to the standard design. Both transistors in this design are still in the linear region. In the standard decap design, the gates of the transistors are directly connected to either V D D or Vss, depending on the transistor type. In this case, the gate of the NMOS device is connected to V D D through the channel resistance of the PMOS device. Similarly, the gate of the PMOS device is tied the channel resistance of the NMOS device and then connected to Vss- The added channel resistance to the gate provides the input resistance Rin for ESD protection, as previously mentioned in Section 2.4. The input resistance can help to limit the maximum current flow to the decap so that the voltage seen from the gate of the decap is also limited. Intuitively, the input resistance along with the decap can be thought as a low pass filter, as illustrated in Figure 3.2. When there is a sudden voltage jump at the power line (Vjn) during an 23 ESD event, the voltage at the gate of the decap (V g a te) does not increase instantaneously with the increase of V j n . Instead, the increase of V g ate is delayed due to the low-pass RC effect. This time delay in the voltage change at the transistor gate helps to protect the gate until the primary and secondary ESD devices are fully operational and shunt the ESD current away. Hence, it is desirable to have Rin as large as possible from the standpoint of ESD protection. Figure 3.2: Intuitive understanding of input resistance in cross-coupled design. By simply swapping the terminal connections, the cross-coupled design adds a considerably large Rjn to the gate of the decaps, without increasing the layout area. The tradeoff of this design is the reduced transient performance as a decap. Since both transistors are in the linear region, the two transistors are on and do not limit the gate leakage current flow compared to the standard design. Hence, no savings in gate leakage are achieved in this cross-coupled approach. This design nicely illustrates the concept of tradeoffs between transient response and ESD reliability when designing a decap. This cross-coupled circuit is also considered to be fairly passive and robust. In the next chapter, this circuit will be discussed in greater detail, and modifications will be provided to address different design tradeoffs including gate leakage. Two of the modification circuits can achieve higher gate leakage savings and provide comparable or better transient response, while another is better in terms of ESD reliability. 24 3.3 Gated Decap Chen et al. [12] has recently implemented a new decap structure that saves gate leakage. The structure is called gated decap, as shown in Figure 3.3. A control transistor is inserted between the standard NMOS-only decap and the Vss line. The source and drain of the decap are connected to the source of the control transistor, making the node a virtual ground (V_GND). The drain of the control transistor is tied to the real Vss- As shown in Figure 3.3, the substrate of the decap is still attached to Vss- There are two major components in gate leakage: leakage current from gate to channel (Igc), and from gate to substrate (Igb). The current Igc can be partitioned into two: leakage current from gate to source (Igcs) and from gate to drain (Igcd) [19]. The amount of gate leakage from gate to substrate Igb is roughly lOx smaller than the leakage from gate to channel Igc [15] [16]. Thus, the substrate of the decap does not need to be tied through the control transistor, and the leakage current Igb is neglected. V D D Igc + Igb Decap V S S Figure 3.3: Basic gated decap schematic [12] and gate leakage flow. There are two modes of operation of the circuit: active mode and power saving mode [12]. When in the active mode, the Ctrl signal of the control transistor is turned high. The gated decap 25 operates almost like the standard decap, except that there is a small channel resistance of the control transistor. The size of the control transistor needs to be large to have the channel resistance small since a large resistance will reduce the transient response of the decap. When in power saving mode, the Ctrl signal is turned low so that the control transistor operates in the subthreshold regime. The node V G N D can be considered a virtual ground (floating), where the voltage at V G N D can be determined by the series resistance of Re/f of the decap and the channel resistance of the control transistor. In this configuration, the gate leakage saving is projected to be 99% in a 70nm process [12]. The basic idea of the gated decap is from multi-threshold CMOS (MTCMOS). The control transistor comes from the concept of the sleep transistor in MTCMOS. As expected, the control transistor should have a high V r to keep the subthreshold leakage small. The largest challenge of this gated decap would be the proper selection of the Ctrl signal. At the top level, the Ctrl signal can be driven by the hardware/software interface. When there is no activity in the system, the operating system (software) will set up the signal to force the chip into power saving or standby mode. From the hardware architectural level, the Ctrl signal can be managed by some self- predictive architecture [12] [23]. At the circuit level, it is desired that the gated decap is self- maintained, and no external circuitry is required to control it on or off. In that case, it may need to have a special clock, as shown in Figure 3.4. Before the regular clock rises, the Ctrl signal can be set high to allow some setup time for the decap to fully setup. When the regular clock falls, the Ctrl signal can also fall simultaneously to save power. The time period when the Ctrl signal is low can be considered as the power saving period. 26 clock Ctrl signal setup time power saving Figure 3.4: Sample clock for the Ctrl signal in gated decap. Another substantial difficulty is oscillation. It was observed in [12] that the voltage levels at the local power lines oscillate when the gated decap is turned on or off. The reason is that sharp rises and falls in the Ctrl signal get passed through the decap and hence make the power lines noisy. The oscillation level is determined to be excessive: more than 10% of VDD of the simulation process [12]. Such large oscillation is certainly non-acceptable and some form of modification has to be taken. The solution of reducing excessive oscillation provided in [12] is to insert a small-size inverter, as shown in Figure 3.5. Sharp rising and falling edges in the Ctrl signal correspond to the concept of large slew rate. The insertion of the small inverter helps reduce the slew rate at Ctrl. V D D I j—I I—j Decap V G N D Ctrl , „ Cm [>o 1 [" Gate Small \ inverter ~_r V s s Figure 3.5: Insertion of small inverter in gated decap |12|. 27 The gated decap is a good attempt in solving the problems of excessive gate leakage for decap designs. Nonetheless, the design style is not conservative enough so that it experiences many issues such as oscillation. In other words, the robustness of this gated decap may not be good enough to implement in industrial designs. 3.4 Thick Oxide Decap Fabrication foundries usually provide high-voltage, thick-oxide MOS devices in a CMOS process. The thick-oxide devices are intended for the use in I/O interfaces and other places where a higher voltage supply is present. Typically, for a 90nm process, the nominal V D D is scaled to 1.0V, while the thick devices can still hold for 3.3V voltage level [24]. Similarly, for a 130nm process, with a nominal power supply of 1.2V, the high voltage for the thick-oxide transistors is 3.3V [25]. For thick-oxide devices in a 90nm process [24], the oxide is roughly 3x thicker than the thin- oxide devices, resulting in 3x higher oxide breakdown voltage. Moreover, because of the exponential relationship between tox and Jgalejeak given in Equation (2.3), the gate leakage of such thick-oxide devices in 90nm is almost zero, which is also consistent with SPICE simulations. Hence, the use of thick-oxide transistors can eliminate the concerns of ESD reliability and gate leakage completely. The largest disadvantage of thick-oxide devices is that the effective capacitance Ceff is reduced by roughly 1/3. Moreover, it is difficult to place thick-oxide devices within a standard-cell block. The thick-oxide decaps must be properly placed around the periphery of the block. The use of 28 thick decaps is only suggested in the open areas between blocks when both ESD risk and gate leakage need to be minimized and while there is also a high demand on transient response performance. Under such scenarios, the 3x area penalty may have to be paid. To complete the concept of thick-oxide decaps, there is a similar situation where a stack of thin- oxide decaps is used, as shown in Figure 3.6. Assuming a 90nm process has a 1.0V power supply and a threshold voltage V T of 0.3V, V T is roughly at V D D / 3 . Stacking three thin-oxide decaps in series results in the gate voltage difference Vox across each decap to be V D D / 3 . A S mentioned in Section 2.3, gate leakage is a function of biasing voltage. If Vox is in the subthreshold region (close to V T ) , the leakage current is 3-6 orders of magnitude less than the leakage current in strong inversion. Namely, biasing the decap in the subthreshold region will have negligible gate leakage. The disadvantage of this approach is also similar to that of thick-oxide devices: it serializes 3 decaps and has therefore a resulting equivalent capacitance of 1/3 of one decap. Thus, in order to provide certain amount of decoupling capacitance, much more areas (~9x) are needed in this fashion. V D D ~2Z~ Thin-oxide Decap - Vss Figure 3.6: Stack of thin-oxide decaps versus thick-oxide decap. Thick-oxide Decap 29 Showing the idea of the stacked thin-oxide decaps is only to further illustrate the concept of using thick-oxide devices. Stacking thin-oxide decaps does not have practical applications due to its large area requirement. 3.5 High-k Gate Dielectric The oxide capacitance Cox is a critical factor to many physical properties of MOS transistors. The drain current IDS of a transistor is proportional to Cox- A larger Cox results in a larger drain current and hence a faster transition or a shorter gate delay [3]. Also, the subthreshold leakage including drain-induced barrier lowering (DIBL) is related to Cox- A larger Cox corresponds to smaller subthreshold leakage and less D I B L effect [15]. As a consequence, each technology generation attempts to increase Cox by roughly 1.4x while reducing the channel length L to 0.7x of the previous technology's channel length. The result is that the product of CoxL has been maintained constant for over 25 years [3] as technology scales. The increase in Cox balances the tradeoff between the drain current and the subthreshold leakage current in each technology node. From Equation (2.3), the gate leakage density is inversely related to tox- A smaller tox leads to exponentially increasing gate leakage. From the gate leakage perspective, the oxide thickness tox should be kept large. However, the oxide capacitance per unit area, Cox, is determined by [3]: C o x = ^ (3.1) 'ox where eox is the permittivity of the oxide and is fixed for a given oxide material. Equation (3.1) suggests that if eox is kept unchanged, the increase in Cox will lead to certain decrease in tox and hence exponential growth in gate leakage. 30 Knowing that the gate leakage increase may be excessive for 90nm technology and below, in order to keep tox thick while increasing Cox, one can adjust the relative dielectric constant, k, where s o x =k-s0, and £ 0 is the vacuum permittivity. If a high permittivity (high-k) dielectric can be used instead of the normal SiC>2 oxide, the physical oxide thickness tox would no longer be limited by its electrical property Cox- This concept of using high-k dielectrics was first presented in [26], and researchers and process engineers have continued to pursue better high-k materials [27]. Most experts agree that high-k gate dielectrics will help to keep the gate leakage under control [27]. Commonly suggested high-k materials include HfG"2, ZrC»2, and A I 2 O 3 [27]-[29], whose permittivity ranges from 10 to 30, compared to 3.97 of SiC>2. [30] presents the materials of barium titanate (BTO) and barium strontium titanate (BSTO) that have permittivity ranged from 100 to 400, about the highest among the up-to-date research results. The application of high-k gate dielectrics is currently an active research area. Many challenges still remain [27]: thermal stability of the dielectrics, interfacial layer formation, effective oxide thickness control and environmental sensitivity, channel mobility degradation, high-k dielectric stability with poly-silicon gates, and possible use of metal gate instead of poly-silicon. Among all, the two most critical problems are: (1) High-k and polysilicon gates are incompatible due to Fermi level pinning at the interface between high-k and polysilicon, which causes high threshold voltages in transistors. (2) The high-k/polysilicon transistor structure exhibits a degradation of 31 channel mobility p. due to Coulombic scattering since high-k MOSFETs tend to have more oxide charge and interface traps [27] [29]. Until the majority of the above mentioned issues are solved, high-k dielectrics may not be applied to industrial designs. [31] predicts the availability of high-k technology in the year 2007. At least for now, for a typical 90nm process [24], the oxide material still uses regular Si02, which was also the case for the 130nm technology. 3.6 Metal-Insulator-Metal Capacitor Many fabrication processes support the implementation of metal-insulator-metal (MIM) capacitors. M I M capacitors can be integrated into both aluminum and copper interconnect backend of the line (BEOL) processes [32]. In an A l process, a M I M capacitor is usually composed of an A l bottom plate with Ti or TiN liners, a silicon dioxide Si02 or silicon nitride Si3N4 dielectric, and a titanium nitride TiN top plate [33] [34]. For Cu processes, various M I M integration schemes have been reported for the past few years by several research groups [34]. The materials of metal electrodes and dielectrics in use vary from case to case [34]. Typically, in a Cu process, a M I M capacitor is composed of a Cu (or Ta or TaN) bottom place, a plasma- etched chemical vapor deposition SiN dielectric, and a Ta top electrode [34] [35]. M I M capacitor designs typically utilize the top metal layer and the next lower metal plate as the two capacitor electrodes in order to minimize the parasitic coupling capacitance between the bottom of the M I M plate and substrate [33]. 32 M I M capacitors are popular in analog, mixed-signal, and RF IC designs, mainly because of high linearity, low series resistance, high capacitance density, high precision, and low parasitic capacitance [33]. The depletion-free, highly-conducting metal electrodes are suitable for high speed applications at low cost [34]. In addition, M I M capacitors usually have small leakage currents, mainly because the dielectric thickness is large (>50nm) [33]. The low effective resistance and low leakage make M I M capacitors good candidates for white-space decaps. When connecting to the M I M capacitors, the interconnects need to be kept short and wide so that the total resistance is maintained low [35]. 3.7 Summary This chapter described a number of design approaches for decaps in recent technologies. Starting from circuit level, cross-coupled decap, gated decap, and thick-oxide decaps are discussed in details. Process level efforts are also taken into account, where the use of high-k gate dielectrics and M I M capacitors is addressed. Moreover, researchers are implementing decaps in special MOS structures [36] and claiming good results in gate leakage savings. Another circuit design approach, called switched decap, is more complex and will be discussed separately in Chapter 5. As already mentioned, decap design with gate leakage consideration is still an active field because the problem of excessive gate leakage is fairly new. In order to make further improvements from the existing design approaches, the cross-coupled design is investigated in this research since it is commonly used in 90nm standard-cell libraries. 33 Chapter 4 Passive Decoupling Capacitor Designs 4.1 Introduction The objective of this chapter is to provide passive decoupling capacitor designs that properly tradeoff between their transient response, ESD performance, and gate leakage. The basic idea of the cross-coupled decap is to use a crossly coupled N+P decap pair to reduce ESD risk by adding series resistances to the gates. Continuing on from the discussion in Section 3.2, modeling of the cross-coupled decap is provided. Before any improvements can be made, detailed transient, ESD and gate leakage simulations have to be setup and carried out to compare against the standard decap. After the quantitative analysis of the advantages and limitations, three modifications based on the basic cross-coupled design are then proposed in [37] and [38]. One sample cell layout for each of the modified circuits is provided. From the simulation results, recommendations are made as to how to select the appropriate design for a given technology or a process. 4.2 RC Modeling of Basic Cross-Coupled Decap Design Knowing that the standard N+P decap design may no longer be suitable for 90nm technology due to the increased ESD risk, a cross-coupled decap design has been proposed [11] to address 34 the issue of ESD reliability. It reconnects the terminals of the two transistors: the drain of the PMOS connects to the gate of the NMOS, whereas the drain of the NMOS is tied to the gate of the PMOS [11]. V D D V D D •efF_overall eff_overall Vss Figure 4.1: Cross-coupled decap schematic [11| and modeling. The design can be modeled as a series connection of Reff and Ce/f, similar to the standard decap, as illustrated in Figure 4.1. The overall Q ^ i s roughly the same, while the overall Reff increases significantly. Both transistors are still in the linear region, but the channel resistance is modified. Specifically, Ceff_overall ~ ^eff _n ̂  ^ eff _p = ^eff _n + ^eff _p (4-1) Rejf_overall ~ (Reff _p + Ron_n) 1 1 iReff _n + Ron_p) *R IIR = R°"-"R°"-» (4.2) on_n on_p D , D V ' Kon_n+K0„_p where Ceff_n, Ceff_p, Rejf_n and Reffj> are the intrinsic effective capacitances and resistances, respectively, and RONJ> and RONN are the channel resistances of the two transistors. Since RONJ) and R0n_n are at least one order of magnitude larger, Re/f_p and Ref/_n can be neglected in the overall Re/f calculation. Here, R KR — (4.3) 35 where Req is the process-dependent square resistance (kf2/a). It is important to realize that Equation (4. l)-(4.3) are first-order, low-frequency approximations only. The real transistor channel resistance by nature is nonlinear and depends strongly on applied voltages, operating frequency, and geometry [3]. The only reason for providing these formulae is to give designers some insight into the design tradeoffs. This cross-coupled design improves the ESD performance of the decap by making the overall effective resistance larger without adding additional area. The tradeoff of the design is a reduced transient response. The larger Ref corresponds to a longer RC delay. In addition, this design provides no savings in gate leakage as compared to the standard design. To quantitatively measure ESD performance, transient response, and gate leakage, a number of simulations were carried out. The layouts were created in Virtuoso™ Layout Editor, verified by Calibre™ DRC checker, and then extracted by Calibre X R C parasitic extraction tool. The extracted data were simulated with HSPICE™ for different simulation setups. For fairness, the same cell area was used for all the designs. 4.3 Transient Response Simulation In order to carry out some simple but efficient transient simulations, the setup in Figure 4.2 was chosen to evaluate the time domain decap performance. The setup is a pessimistic situation where no power supply is present. Only the decap provides the needed current to charge up the load when the inverter switches. The node V * is initially charged to VDD, and the output is initialized to OV while the total switched capacitance Cm,j,ched is set to roughly 1/10 of the 36 decoupling capacitance and then fixed. Note that CSWi,ched includes the output parasitic capacitance of the inverter. The input of the inverter is initially set to V D D . At 3Ops, it starts to drop linearly from V D D to OV, reaches OV at 60ps, and then remains constant. Cross-Coupled Decap " D D Driver Standard Decap 'switched r 5 i " D D Driver 'switched Figure 4.2: Schematic for the first transient setup. To simplify this transient setup, the decap can still be treated as series of Reff_overaii and Ceff_overaii, as shown in Figure 4.3. The values of Reff overall and Ceff_overau would be different for cross- coupled and standard decaps. Reff overall " D D Ceffoverall S~ Driver c ^switched Figure 4.3: RC modeling of the first transient setup. It is possible to gain insight into the required Ceff_overaii value obtained from the final V * voltage. When the transient analysis runs for sufficient long time (>lns), the voltage level at V * stabilizes. Applying the charge-sharing equation, the final voltage V * can be derived as a function of Ceff_ovemii, as follows: Qbefore Qafter C V =c V*+C V* eff _overall DD *-eff _overallr switched' 37 v* = c iff ̂ overall \ ^eff_overall T ^switched J V, DD (4.4) If the decap has a large Ceff_ovemii, which is desired, the final V * value will be also large and close to the initially charged value V D D . One step of further simplification of this circuit can be used to understand the significance of R-eff_overaii of the decap. For this purpose, the Reff_0veraii and Ceff_overau are both assumed to be fixed. Also, the T to '0 ' ramp transition that the inverter switches is replaced by a pulse, meaning that there is no time delay from V D D to 0 when the inverter switches. Since the NMOS device in the inverter can be assumed off during the transition, it can be neglected from the model. The switching of the PMOS device in the inverter is simplified as a constant channel resistance of RP_channei- Therefore, the circuit can be modeled as shown in Figure 4.4. Switch closed att = 0 It. eff overall Ceff overall Rp channel C switched Figure 4.4: Simplification of RC modeling of the first transient setup. After applying Laplace transform to the circuit in Figure 4.4, the voltage at V * can be expressed in the s-domain as a simple voltage divider: Rp channel + 1 V*(s) = - sC switched V, DD Rp channel sC switched ^eff_overall + ^ eff ̂ overall 38 C R C — R C / eff _ overall } V ^ ~ c*'K m w e' switched eff _ overall eff _ overall D C + C DD (C -+- C V R + R ) rr at / \ f / T overall switched V overall switched / V eff overall P channel) / A c\ V*(s) = — + — ^ = (4.5) , eff overall switched S H — c c r R + R ~\ eff _overall switched \ eff _overall P_channel* Applying the inverse-Laplace transform, the time-domain voltage V * is (for t > 0) : V*(t) = VDD (^  Ceff-° veraJ! ) • u(t) + c +c eff _overall switched y r_channel switched eff _overall ejj _overall ^ <-ejr_m<™//+ Linched (^eff_overa/l + ^'swilched^i^eff overall + ^P channel^ Here, the final voltage V * is consistent with the simple charge-sharing calculation. The time C C constant associated with V * is (Reff overall +RP chame,){—eff-overa" 5W"ched ) ; the serial combination ^eff _overall ^switched of Reff_overaii and Rpchannei multiplied by the serial combination of Ceffoveraii and Csv>itched- The effective overall resistance Reff_overaii of the decap should be made small so that this time constant will also be small. HSPICE simulation for this setup is illustrated in Figure 4.5, where the response of the two designs is plotted. Although not shown, the hand calculation can generate curves that are close to the SPICE results, as expected. From the figure, the two designs have close effective capacitance because their final voltage levels at V * are close. On the other hand, the cross-coupled design experiences larger Reff_overaii, resulting in an undershoot and faster voltage drop as input switches. Clearly, the standard decap can provide much better transient response. 39 0 50 100 150 200 Time (ps) Figure 4.5: Transient response for the first setup. The simplified model indicates that the value of Reff_overaii determines if there is an undershoot in the transient response. Specifically, if RP_^,CmllcM>R^_mmllC^_overall, then the voltage at V * exponentially drops to its final value without an undershoot. Otherwise, if Rp_cHanneiCmUcte, < Reff _overal,Ceff ^era„ > the voltage at V * will drop below its final value first and then exponentially increase back to the final level, which is an undesired case. It is evident from the transient response perspective that the decap should be designed to have large Ceff_overau and Small Reff_overaU- Note that the above circuit simplification is only intended for giving designers some useful guidelines. The real situation involves many nonlinear factors such as varying Rp_channei when switching, and varying Reff_OVera/i and Ceff_overaii at high frequencies. However, for the purpose of 40 first-order calculations or estimations, the simplified model gives valuable insight into the design tradeoffs. Another simple setup was also used to determine the effective capacitance value and the RC delay of the decap. The setup is shown in Figure 4.6. The V D D node is connected to the nominal supply of IV (for 90nm), and Vss is tied to a common ground. When there is no activity, the current flow from V D D to Vss is solely due to gate leakage. At Ins, V D D starts to drop linearly from IV to 0.9V, reaches 0.9V at 2ns, and then remains constant. I decap Idecap V D D 0 . 9 V D D Cross-Coupled Decap i V D D 0 . 9 V D D Standard Decap r£b Figure 4.6: Schematic for the second transient setup. By definition, an ideal capacitor responds to a voltage change as a current source i f it is fully charged, as follows [39]: / = C — (47) decap decap ^ decap V * / If the voltage change is a ramp, the current provided by the ideal capacitor should be a pulse. In practice, due to the presence of the effective resistance associated with the decap designs, a certain amount of RC delay exists. A good transient response should have sharp rise and fall edges (at Ins and 2ns in this case), and also provide a large average current Iavg during the time period from Ins to 2ns. The sharpness of rise and fall is measured from the rise/fall slopes with a 41 unit of A/s. The average capacitance CAVG is calculated from IAVG from Equation (4.7). Figure 4.7 illustrates the curves for the two designs in transient analysis, and indicates that the standard decap is better in the transient response. The result in this plot is consistent with the result obtained from Figure 4.5 previously. 6 5 < 3 ¥ 3 t 3 U 2 ~ 1 0.0 1.0 Std. N+P decap Basic cross-coupled decap 2.0 3.0 4.0 5. Time (ns) Figure 4.7: Transient response for the second setup. 4.4 ESD Performance Simulation The ESD simulation requires an ESD generation model. Among all the existing models, the human body model (HBM) was adopted for simplicity. Following the standard MIL-STD-883x method 3015.7 [10], a human body can be simulated as a series of 1.5kQ resistance RHBM and lOOpF capacitance CHBM- The capacitor CHBM is initially charged to 2kV that needs to be discharged through some primary elements. The primary element is arbitrarily chosen to be an 42 ESD diode plus a gate-coupled NMOS device (GCNMOS) with an n-well resistor Rnweu (~15kQ) and an NMOS bootstrap capacitor Two identical primary elements are used to protect the circuit placed in between the H B M generation and the elements, as shown in Figure 4.8. For simplicity, no secondary element is used. Initially charged to 2 K V V DD R-HBM — 1-5KQ ^ ^ = C H B M = 1 0 0 P F Circuits C B ^3 G C N M O S C B :R, nwell G C N M O S :R, -nwell = V; ss H B M generation Primary element Primary element (Duplicate) Figure 4.8: Simulation setup for ESD analysis [10]. Since the primary elements are designed to handle large current flow, the maximum current density, Jmax, is assumed to be within the safe range and is not measured. H B M generation raises the voltage level at node V D D , and hence turns on the primary elements to discharge. For device protection from oxide breakdown, the voltage differences across gate and source ( V G S ) and across gate and drain ( V G D ) of the two transistors are simulated. The V G S and V G D voltages should to be kept as low as possible, given that the oxide breakdown voltage for a typical 90nm is below 5V. From simulation measurements, it was found that: • For standard decaps, V G D _ p = VGs_p = V G D _ n = VGs_n = 4.2V. • For cross-coupled case, V G D _ p = 4.0V, VGS_II = 3.2V, and V Q S _ P = VGD_II = 3.0V. 43 The cross-coupled design provides better ESD protection by making the overall effective resistance larger without adding additional area. However, the improved ESD performance is at the expense of transient response, as described earlier. 4.5 Gate Leakage Simulation The gate leakage levels can be obtained from the two transient setups in Section 4.3. In the first transient setup (Figure 4.2), before the inverter switches, the static current flow through the decaps can be treated purely as gate leakage. In the second transient analysis (Figure 4.6), before the node V D D starts to drop its voltage, the current flow through the decaps is solely gate leakage. When carrying out SPICE simulations, it is essential to use BSIM4 version to have gate leakage models built-in [14]. Earlier BSIM versions do not support gate leakage models [14]. The gate leakage in BSIM4 is partitioned into two parts: the tunneling current between gate and substrate (Igb) and the current between gate and channel (Igc) [19]. Since the current Igb is considerably smaller than Igc, Igb is set off by default [24]. To make sure both current components are set on for the best accuracy, two selectors, IGBMOD and IGCMOD, need to be set' 1' [19]. As discussed in the earlier sessions, the cross-coupled decap design does not provide any savings in gate leakage. HSPICE simulations show that the two designs have almost identical gate leakage: 53.8nA for the standard decap and 53.7nA for the cross-coupled design. 44 4.6 Modified Cross-Coupled Decap Designs Three modifications are made to address different goals of decap design: ESD performance, transient response, and gate leakage. It is difficult to simultaneously make improvements on all the three goals, but trying to balance them and to make tradeoffs is certainly feasible and indeed achievable. Each modification is compared to the basic cross-coupled design to show advantages and disadvantages. Again, the total cell area is fixed for all the designs. The first modification (Modi) attempts to improve ESD performance by making the channel lengths of the two resistors longer (Figure 4.9). The two fingers are combined into one. As a result, the overall Re/f is almost doubled, while the overall Cejf remains roughly the same. The disadvantage of this design is reduced transient response and slightly larger gate leakage since the gate area increases a little. Figure 4.9: Sample layout of Modi (basic circuit without fingering). The second modification (Mod2) attempts to reduce gate leakage while maintaining ESD performance and transient response at roughly the same level (Figure 4.10). One NMOS is replaced by a PMOS with the n-well expanded to accommodate the new PMOS. The effect of 45 this change is then increased Ronj) and Ceff_p. To match ESD performance, Ron_n needs to be reduced. One simple change to obtain a small Ronj7 is to reduce the channel length of the NMOS. By the same token, Ceff„ is also reduced. The result is comparable ESD performance and transient response i f carefully designed. Using the fact that the new same-area PMOS leaks 3 times less than the replaced NMOS, extra saving in gate leakage is realized. Figure 4.10: Sample layout of Mod2 (replace NMOS with PMOS). The third modification (Mod3) (Figure 4.11) follows the similar approach as of Mod2. It further increases the new PMOS area while reducing the NMOS area. Indeed, the minimum length NMOS is used to obtain the smallest possible Ron_„ so that it dominates and makes the overall Reff smaller. Since the overall Rejf is greatly decreased while the overall Ceff is somewhat higher, the transient response dramatically improves. The only downside is reduced ESD protection capability due to the reduced overall Reff. 46 Figure 4.11: Sample layout of Mod3 (replace NMOS with PMOS, and use smallest NMOS). Table 4.1: Comparison on ESD performance, transient response and gate leakage. ESD performance with 2 primary elements Transient response Gate leakage V G D _ p ( V ) V G S _ n (V) V G S P = V G D _ n (V) First setup Second setup Leakage current (nA) Voltage drop rate (V/ns) Rise slope (A/s) Avg. cap (IT) Std. Decap 4.2 4.2 4.2 -1.8 2.8e5 54.3 53.8 Cross- coupled 4.0 3.2 3.0 -5.4 8.2e4 33.1 53.7 Modi 3.8 2.9 2.8 -5.3 8.7e4 21.4 59.7 Mod2 4.0 3.7 3.4 -8.6 7.0e4 35.8 33.6 Mod3 4.1 3.9 3.8 -7.0 l.le5 47.5 31.8 Following the same simulation procedures outlined earlier, Table 4.1 summarizes the comparisons for all the designs on ESD performance, transient slope response, and gate leakage. The bold numbers indicate the best results in the comparison. The standard decap provides the best transient response. Modi provides the best ESD protection, while Mod3 provides the lowest 47 gate leakage. One can view Mod2 as a compromise between Modi and Mod3. The complete transient simulations for the first and second setups are also depicted in Figure 4.12 and Figure 4.13, respectively. 1.2 | : 1 Time (ps) Figure 4.12: Complete transient response for the first setup. 48 Std. N+P decap Time (ns) Figure 4.13: Complete transient response for the second setup. There is no single design that is optimal for all the possible specifications. The reason for having several design options is to provide designers with different solutions so that they can make suitable tradeoffs for a specific process at a specific technology node. For 90nm technology, the standard decap still seems to be acceptable in ESD reliability, assuming the power rails have protection elements. However, Mod3 is more suitable because it has better ESD performance and saves roughly 41% on gate leakage. The only tradeoff then is a slightly reduced transient response. As technology further scales, or as a different process increases the transistor speed, the oxide thickness will probably become thinner and the oxide breakdown voltage will occur. Under that scenario, the standard design or the Mod3 will no longer be appropriate. For improved ESD performance, Mod2 is recommended instead of the basic cross-coupled design. The reason is that Mod2 has similar ESD numbers and similar transient response compared to the basic cross-coupled design but saves approximately 40% on gate leakage. When technology 49 scales down to a point that the oxide thickness makes the ESD reliability a more serious concern, the use of Modi will be advised for the best ESD performance, although its transient response will be sacrificed significantly. The recommendations above are good for moderate or low frequency chips. If the targeting frequency is extremely high, even Mod3 may not be able to provide desired amount of current within an excessively small period of time. Under such a case, the use of thick-oxide decaps is suggested around the standard-cell blocks. As mentioned in Section 3.4, for 90nm technology, the oxide is 3x thicker than the thin oxide, resulting in almost zero gate leakage and 3x ESD breakdown voltage. The disadvantage is the effective capacitance reduced to 1/3. Hence, the area needed for a fixed capacitance is 3x for thick-oxide decaps. The thick-oxide decaps must be properly placed around the periphery of the block. The fabrication cost for using thick-oxide devices may also be slightly higher, although it may be needed for I/O and other features. As technology further scales to 45nm or below, the gate oxide will probably become ultra thin and will dramatically increase the ESD risk and the amount of gate leakage. The use of the cross- coupled design and its modifications in this chapter will be eventually limited. The anticipation at this stage would be the use of high-k gate dielectrics as the oxide materials so that the electrical thickness and the physical thickness can be differentiated to completely eliminate the concerns of ESD reliability and gate leakage. Other approaches would be to utilize M I M capacitors as decaps or some other innovative structures, as discussed earlier. In any case, solutions that properly balance gate leakage, ESD, transient response and area will be required. 50 Chapter 5 Active Decoupling Capacitor Designs 5.1 Introduction Passive decaps described previously have a small layout and are useful within the standard cells. However, for large global decaps (i.e., outside the block), other approaches can be used. This chapter investigates active decap design approaches at the circuit level that help reduce voltage variation on the global power grid. Specifically, the design of switched decoupling capacitors, as power grid voltage regulators or stabilizers, will be studied here. The switched decaps amplify the charge storage capacity of the basic decap while monitoring the power rail activity to provide dynamic control of the switching response. The switched decaps have better area efficiency, compared to the passive designs. The design complexity of switched decaps is much higher than those discussed in Chapter 3. As a consequence, these designs are separated out and analyzed in this chapter. There are two designs that use switched decaps: a voltage regulator (VR) from Sun™ [40] [41] and an active power stabilizer (APS) from Fujitsu™ [42]. The objective in this chapter is to evaluate the two designs to replace the global thick-oxide decaps with better voltage regulation capability. After a full understanding of the advantages and limitations of the two designs, a new 51 low-power and high-performance design is proposed. The new design has similar performance to the Sun VR, but requires a lower power level that is close to the Fujitsu APS. The significance of this work is that the switched decaps can potentially be used for all global decaps outside standard-cell arrays. It provides better power-grid noise reduction and lower power consumption, making the designs valuable for both ASIC and full custom designs. 5.2 Switched Decoupling Capacitor There exists the need for a more area efficient way of regulating the voltages on the power grid other than the standard decaps. Sun Microsystems and Fujitsu have proposed two designs to address this issue [40]-[42]. The fundamental idea of the two designs is to actively switch the decoupling capacitors to boost up the power grid voltage and provide more instantaneous current. The principle of operation of a switched capacitor is illustrated in Figure 5.1. In the standby state, two standard decaps, Cdecap, are positioned in parallel, resulting in an equivalent capacitance of 2Cdecap. The total charge accumulated at the capacitors is AQ = 2 C / C W A V', where A V is the voltage difference on the power grid, V D D - Vss- When current flows into a switching logic gate, the voltage difference A V between V D D and Vss will reduce as well. AQ = 2Cdecap-AV ^ 7 Vss Vss Figure 5.1: Principle of switched decoupling capacitor [40]. Vss 52 Some circuitry senses this voltage variation and switches the two parallel capacitors into a series connection. When the capacitors switch, the charge AQ cannot vary instantaneously, and thus remains at its initial value for a short while. The equivalent capacitance, however, shrinks to Cdecapll by stacking the two capacitors in series. As a result, the new A V turns out to be 4AV. In other words, the power grid voltages V D D and Vss are boosted up by four times (ideally). Similarly, when the power grid moves to a charging stage, the two capacitors are switched from series to parallel to make the voltage difference A V smaller. By switching either from series to parallel or from parallel to series, the switched capacitor circuit has the capability of regulating the voltage variations on the power grid. Ideally, A V can be increased or reduced by 4 times. However, this can never be achieved in reality because the power mesh and the decap circuitry non-idealities limit the excessive voltage variations on-chip. The switches in the circuit can be implemented using MOS transistors. One possible configuration is depicted in Figure 5.2 [40]-[42]. The two NMOS and two PMOS transistors operate as switches. The control signals at the gates of the transistors are aup, bup, adown, and bdown. When the capacitors are in parallel, both M n l and M p l are on while both Mn2 and Mp2 are off (i.e., in the subthreshold region). When the capacitors are in series, both M n l and M p l are off while both Mn2 and Mp2 are on. 53 V D D V D D C decap ={= C"lP M p l j ( 3 - M P2jff- adown —| [~ Mnl bdown I Cdecap < 7 Vss Vss Figure 5.2: MOS implemented switched decoupling capacitor [40]-[42]. Since the transistors operate as switches, their "on" resistance Ron are the device channel resistances. When the capacitors are in parallel, the "on" resistances (Ron) of M n l and M p l are connected to the decaps. When the capacitors are in series, the new Ron' is the parallel combination of "on" resistances of Mn2 and Mp2, as previously shown in Figure 5.1. To reduce ohmic losses, the "on" resistances need to be minimized by increasing the widths of the transistors. Specifically, the /?o n's should be kept in the range of a few ohms. Therefore, the widths W of the switch transistors are required to be in the range of 10,000A., where X, is a half of the minimum transistor length for a given technology [3]. However, with such large switches, the drivers generating the switching signals need to be strong enough, indicating the necessity of having a large sensing and switching circuitry that consumes a considerable amount of power and area. The decaps used in the circuit can be designed using either a thick or thin oxide, depending on the leakage and area tradeoff. As mentioned previously, the switched-decap designs are intended 54 to maximize the area efficiency, and do not directly provide for any gate leakage savings. However, some of the gate leakage saving techniques discussed in the previous chapters can be applied here to control the leakage power. 5.3 Sun's Voltage Regulator Sun's sensing and switching circuit is a voltage regulator (VR) that contains four main blocks: a reference voltage generator, a high-pass filter, a two-stage amplifier, and switched decoupling capacitors. The block diagram is illustrated in Figure 5.3 [41]. In the same figure, a user logic circuit block is shown to be placed close to the active decap and is considered the main noise source to the global power grid. g l o b a l VDD s t a n d b y @ Vss s t a n d b y @ VDD/2 g l o b a l Vss Reference Voltage High-Pass Two-Stage w/ feedback Filter Amplifier V. Switched Decap V Active Decap : Sun VR Figure 5.3: Block diagram of Sun voltage regulator [41]. User Logic Circuit 55 Three modes of operation can be identified: standby, discharging, and charging. If a voltage change A V on the power grid is sensed by the sensing circuitry, the voltage regulator will switch from the standby state to the discharging state to boost the voltage level back up. After the voltage difference at the power lines rises above the nominal value, the active decap will then switch into the charging phase. When the power-grid voltages are back to the roughly nominal values, the circuit changes to the standby mode. In the standby situation, both nodes bup and adown are positioned at V D D / 2 , whereas aup is at roughly V D D and bdown is at roughly Vss- In the discharging and charging phases, small input variations are amplified to a level where large swings at the output are observed. The large swings of amplified signals are used to switch the decoupling capacitors in either series or parallel. Table 5.1 lists the node biasing and swing values [41]. Standby state indicates how the nodes are biased in steady-state, while discharging and charging states specify the target voltage levels under discharging or charging situations, respectively. Table 5.1: Node biasing and swing for Sun voltage regulator [41]. aup adown bup bdown Standby ~ V D D V D D / 2 V D D / 2 ~ V S S Discharging -Vss ~ V S S ~ V D D ~ V D D Charging - V D D - V D D - V S S -Vss 56 The circuit-level schematics of the reference generator, high-pass filter and amplifier are shown in Figure 5.4 [40] [41]. The first portion comprises the reference voltage generator. The reference voltage is based on a simple voltage divider and is set to roughly V D D / 2 . The second portion is the i?C-based high-pass filter. The noisy V D D (or Vss) signal is fed to the filter. The output signal of the high-pass filter is centered at V D D / 2 and varies according to the noise passed in from the supplies. When the enable signal is high, the corresponding pass transistor behaves as a resistor in the kilo-ohm range. The third portion is the two-stage pseudo-cascode amplifier. The role of the amplifier is to generate the aup, bup, adown and bdown signals to drive the switched decap. t h r e s h o l d offset s k e w e d h i g h @ V D D Reference Voltage High-Pass I Two-Stage Filter Amplifier Figure 5.4: Circuit implementation of Sun voltage regulator [40] [41]. bdown bup 57 To set the outputs at desired voltage levels at the amplifier stage of the circuit, proper sizing of the transistors is required. Since the transistors in the first amplifier stage are on in standby, the node voltages are determined by the series resistance of the stacked transistors. Similarly, the second stage of the amplifier can be considered as a ratioed inverter. Thus, all the nodes that are skewed either high or low by the second stage of the amplifier can approach V D D or Vss, but not reach these values. Typically, the large swings at the aup, bup, adown and bdown signals will result in longer delay (switching time), but will save standby power consumption. Overall, the sizes of the transistors can be designed by considering the target voltage levels, the desired slew rate at the output, and the total power budget. The operation of the V R moves from standby to discharging to charging. The voltage regulator configuration implies that the decoupling capacitors are in shunt in both standby and charging states. The only situation that capacitors will be switched into series is when the power grid discharges due to logic gates switching in the user logic circuit. From simulation results, this shunt to series switch does not happen until the voltage variation exceeds a certain threshold, for example 60mV. In other words, the sensitivity of the sensing circuitry in this V R that would trigger a switch of the decaps from parallel to series is at about 60mV. One interesting feature of the V R is the feedback loop. Both adown and bup are fed back to the reference generator to enhance stability. Because these two nodes are biased at V D D / 2 , they require the voltage levels at the internal nodes to stabilize after a few ripples generated by the power grid noise. Intuitively, the gain of the amplifier stage must be large since there is a small 58 signal at the input and a large signal at the output. Such a high-gain system has the potential problem of oscillation in the presence of small power grid noise. The feedback ensures that the oscillation will not occur in the system. The main drawback of the V R is its power consumption. The switch transistors as a part of the switched decaps are normally large to produce small "on" resistances (Ron), as mentioned in the previous section. To drive those large switches, however, it is required that the second stage of the amplifier (ratioed inverter) is large enough. Since both adown and bup are biased at V D D / 2 in the standby state, both PMOS and NMOS transistors of the inverter are in saturation region, resulting in relatively high standby power. The high power requirement for the Sun voltage regulator limits its use to high-performance ICs only. For low power ASICs or even portable devices, such a design cannot be used without modifications. 5.4 Fujitsu's Active Power Stabilizer Based on Sun's voltage regulator, designers from Fujitsu developed an active power stabilizer (APS), as shown in Figure 5.5, to help reduce power grid voltage fluctuations [42]. The switched decaps are the same as before and are not included in the figure. Conceptually, the APS and the V R are similar. The small-signal portions of the two designs are almost identical. The basic structure includes a pair of switched decoupling capacitors, a reference generator, and a high- pass filter. However, the two designs differ in the amplification stage. 59 Vup @V D D/2 + 15mV Vmiddle Vmiddle (filtered) @ VDD/2 VW—T- VI ow | @V DD/2-15mV |>H>H>H>H>J^ aup bdown adown bup Reference High-Pass First/Second Stage Amplifier Third Stage Amplifier Voltage Filter (Current Mirror + Common Source) (CMOS Inverter Chain) Figure 5.5: Circuit implementation of Fujitsu active power stabilizer [42]. Similar to Sun's VR, the reference generation is provided by a simple voltage divider without the feedback characteristics. An identical high-pass filter is utilized to sense the Vss grid voltage variations and pass the signal to the amplification stage. The noise on the V D D grid is assumed to be a duplicate of the Vss grid noise and hence is not monitored. Unlike Sun's VR, Fujitsu's APS uses two differential pairs with current mirrors to produce the first-stage amplification. Each amplifier is capable of providing full swings at the output i f the two input nodes are properly biased [43]. However, the gain of such a stage is typically not large. The differential amplifier is followed by a common-source amplifier with a current-source load. The required voltage drop across the current-source load (or required V D S across the load transistor) degrades the maximum voltage swing at the output, V + and V" [43]. The third amplification stage is a chain of standard CMOS inverters that provides two valuable features: (1) the capability of regenerating logic values (either V D D or Vss) at the output by increasing the voltage swings, and (2) the capability 60 of driving large output loads without slew rate limitations. The inverter chain can be sized using the procedure of logical effort [42]. In the APS circuit in Figure 5.5, the nodes at which the switches are turned on or off to drive the decaps are biased differently from Sun's V R in the standby state. Referring back to Figure 5.2, the voltages for each node are listed in Table 5.2 [42]. It is evident that there is no equivalent decoupling capacitance at all in the standby mode since all the switches (Mnl , Mn2, M p l , and Mp2) of the switched decap are turned off. The other modes remain the same as in the case of the Sun VR. Table 5.2: Node biasing and swing for Fujitsu active power stabilizer [42]. aup adown bup bdown Standby V D D Vss V D D Vss Discharging V s s Vss V D D V D D Charging V D D V D D Vss Vss Compared to Sun's VR, Fujitsu's APS has the following advantages. Knowing that the APS occupies slightly less or comparable area compared to the VR, its power consumption is only about 1% of the Sun V R (details in Section 5.6). Such low power characteristics make it attractive for many ASIC designs. Also, it has better control on sensitivity. In the Fujitsu circuit shown in Figure 5.5, the sensitivity was ideally set to be 15mV (per rail). In practice, the circuit will switch only if more than 25mV of voltage variation is present in the power grid. This 61 triggering voltage can be easily adjusted by sizing the transistors differently in the reference voltage generator. On the other hand, the APS also experiences the disadvantages of longer delay time and the potential problem of self-oscillation. The delay time occurs due to insertion of the CMOS inverter chains. For a 0.13um simulation process, the delay can be 300ps or more. For the purpose of regulating voltage variations, such a long delay is not appropriate since the instantaneous voltage drop on the power grid requires an immediate circuit response to boost the voltage back up. The switching response that happens after a long delay is not particularly useful. The other problem is possible self-oscillation. Specifically, lacking a feedback loop, the presence of a switching delay and high sensitivity level may cause oscillations if the gain of the first two stages of the amplifier (the differential pair and the common-source amplifier) is inadvertently large. In addition, there are some minor disadvantages of the APS. If the power grid noise is less than the sensitivity level of the APS, the circuit stays in the standby mode and both decaps are disconnected from the power grid. This configuration is undesirable. Also, the two biasing voltages, Vbiasl and Vbias2, need to be generated by additional reference circuitry. Although not included in the figure, this reference circuitry requires additional area and power consumption. 62 5.5 Low-Power Voltage Regulator After investigating the voltage regulator and the active power stabilizer, it is clear that each one has its own advantages and drawbacks. There still exists a need for developing a new design that has better noise reduction performance than the APS but also requires much less power than the VR. The motivation for the new design is to properly balance performance and power. More specifically, the goal of the new circuit is to try to match the performance of the VR, while trying to control the power dissipation similar to the APS. To understand the new design, first consider Figure 5.2 again. Shown in Table 5.1, in standby condition, both adown and bup are biased at roughly V D D / 2 in Sun's VR. To turn off the two corresponding switches (Mnl and Mpl) , adown needs to be lowered to below V T and bup needs to be raised to above V D D - V T . The delay is basically the average time it takes to shift the two voltage levels, and this delay runs counter to the capability of rapid noise regulation. The new circuit attempts to increase the voltage swing at adown and bup. That is, in the standby mode, adown is biased roughly at V D D , whereas bup is set at about Vss- This reduces the dc current of the amplifier. When switched from standby to discharging state, adown must now fall from - V D D to V T , while bup has to rise from -Vss to V D D - V T . In order to have a large output swing, a common-source amplifier with triode load is chosen to be the second stage of the amplifier. In order to shorten the delay time for large signal transitions, large transistors in the amplifier are necessary. The detailed node biasing and transition details are listed in Table 5.3. 63 Table 5.3: Node biasing and swing for low-power voltage regulator. aup adown bup bdown Standby ~ V D D ~ V D D ~ V S S -Vss Discharging ~ V S S -Vss ~ V D D ~ V D D Charging ~ V D D ~ V D D -Vss -Vss The next step is to design the first amplification stage. Since the main purpose of the second stage is to provide driving capability while the output swing is considered for low power, the first stage needs to provide a high gain. One simple implementation is to use a push-pull amplifier. The push-pull nature of a pseudo-inverter-like amplifier provides a high gain i f biased properly [44]. In addition, the push-pull amplifier also has a high output swing. Typically, a cascode amplifier can provide higher gain due to its high output impedance [43]. Thus, combining the cascode and the push-pull amplifier together, a pseudo-cascode amplifier is used for the first amplification stage. One concern of such a pseudo-cascode amplifier would be its limited input swing. However, the input of the first-stage amplifier is fed from the high-pass filter and can only vary in the range of lOOmV. If the reference generator is well-designed and biases the input of the amplifier at the margin of its high-gain region, the limited input swing of the amplifier is not a problem. Another concern for the amplifier would be variations in the gain under process and temperature variations. In order to have the reference voltages track the high gain region, a reference generator that has a similar structure to the pseudo-cascode amplifier is used. 64 Reference High-Pass Pseudo-Cascode Common-Source Voltage Filter Amplifier Amplifier Figure 5.6: Circuit implementation of low-power voltage regulator. The complete circuit diagram of the low-power voltage regulator is illustrated in Figure 5.6. The reference generators and the high-pass fdters come from the designs of Sun and Fujitsu. The pseudo-cascode amplifier is the first amplifier stage, whereas the second stage is a common- source (CS) amplifier with triode load. In the top CS stage, the NMOS device is significantly larger than the PMOS device, while the PMOS device is larger than the NMOS device in the bottom CS circuit. Considering the top-half circuit in the standby mode, the output of the pseudo-cascode stage is biased below V T . Thus, the NMOS device in the CS amplifier is in subthreshold, whereas the 65 PMOS device in the CS is in saturation. This results in the output nodes of the CS close to V D D - When the power grid starts discharging, the output of the pseudo-cascode stage rises. Assuming that the voltage drop in the power grid is A V and the gain of the pseudo-cascode amplifier is A, the gate voltage at the NMOS device in the CS rises by AAV, which will bring it into saturation if the gain A is large enough. Once the NMOS device is in saturation, the PMOS device in the CS will be forced into the linear (or triode) region. Since both transistors are on, the output is ratioed and is fairly close to V D D because the size of the NMOS device is much larger than the size of the PMOS device. A l l the above discussion applies in a complementary way to the bottom-half circuit. In the standby mode, the NMOS device in the top CS and the PMOS device in the bottom CS experience a comparatively large amount of subthreshold leakage because their sizes are large. This subthreshold leakage, however, is still much less than the on current in the last amplification stage in Sun's VR. Moreover, since the transistor sizes are large enough to provide driving capability, the delay time for the low-power V R does not increase significantly compared to Sun's VR. Hence, the performance of regulating power grid noise is not reduced noticeably for the new design. From simulation results, the power dissipation for the low-power V R is approximately at 10% of the Sun VR. However, its power is still larger than the Fujitsu APS. The new design removes the feedback connection since the standby voltage levels at the output of the amplifier and at the reference generator are no longer identical. That is, in the standby state, the output of the amplifier is at either ~ V D D or ~Vss, whereas the reference voltage is set to 66 be roughly V D D / 2 to bias the pseudo-cascode in its maximum gain region. Therefore, the output signal cannot be fed back to the reference in this new design. Losing the feedback characteristics reduces the stability of the circuit. If the gain of the pseudo-cascode is too large, the output nodes will start oscillating. Designers need to size the amplifier properly to have a suitable gain to avoid this potential problem. Or, if a longer delay is tolerable, the biasing voltage from the reference generator can be shifted slightly away from the high gain region of the amplifier to avoid oscillation. 5.6 Simulation Setup and Results The simulations for all the designs were carried out using HSPICE under BSIM 3v2 transistor models in a 0.13um technology. Since BSIM 3v2 does not support thin-oxide gate leakage simulation, all the power calculations exclude the tunneling leakage, which makes the results slightly optimistic. However, compared to the power dissipation level in the circuit designs, the tunneling leakage is only a small portion of the total power. If thick-oxide decaps are used, the gate leakage can be neglected. Although the simulations were carried out in a 0.13um process, it can be easily adapted for a 90nm process or below since the design concept remains the same. In realistic designs, two elements contribute to the voltage variation A V on the power grid: power-grid resistance R and packaging inductance L. The simulation needs to consider both IR drop and Ldi/dt effect. To estimate the power mesh resistance R, a Layer-8 metal-sheet resistance is considered for a 0.13um, 8-layer copper process, assuming metals 7-8 have a thickness of 0.8um and a width of 20um. The sheet resistance Rsq is roughly 1.7uf2-cm / 0.8um = 20mn/a. 67 Suppose the mesh length is lOOum, one can have the mesh resistance Rmesh equal to 2 0 m f l / D ( ^ ) = 0 . i a 20U7M The inductance L is due to packaging via and bumps (solder balls) from either dual-inline or ball- grid-array (BGA) packaging method. A typical number for the packaging inductance from a B G A packaging option is 0.2nH on both V D D and Vss lines [3]. This 0.2nH applies to all the simulations for consistency. The simulation setup is shown in Figure 5.7. The simplified user logic circuit consists of a switching inverter and a load capacitance Cload connected at the output of the inverter. Since only one inverter is used for simplicity, the size of the inverter is large with the PMOS device at 10,000X / 2k and the NMOS device at 5,000X / 2k. A pair of switched decoupling capacitors has an equivalent capacitance value of O.lnF with 0.05nF on each. The load capacitance Cload is set to be O.lnF since designers typically set Cdecap to be at least 2 to 10 times larger than the Cload to keep the power grid noise within 10% noise budget [4]. A periodic ramp signal Vtest driving the inverter gate comes from an ideal voltage source, and its switching frequency is set to be 400MHz, a typical value for modern ASICs. 68 ide; i eal V D D (1.2V) j r r r r \ 0.2 nH AW r 0.2 nH O.lfi u u u u — * W - idealVss (OV) global V D D (noisy) Decap Circuit Vtest —I JUL E Cload global Vss (noisy) User Logic Circuit Figure 5.7: Simulation setup for active decap circuits. The decap circuit in the setup can be standard decaps, Sun VR, Fujitsu APS, or the low-power VR. Initially, no decap circuit will be used as a reference point for the other circuits. It is important to understand that this simulation setup is somewhat contrived. However, realistic values of packaging inductance, power-mesh resistance and parasitic decoupling capacitance have been used wherever possible. The results are shown in Figures 5.8 to 5.12. The plots illustrate transient analyses for a time interval of 15ns. The upper lines represent the voltage values at the global V D D , whereas the lower lines are the voltage numbers at the global Vss- Quantitatively, the performance of the designs can be determined by measuring the voltage variations on either of the noisy V D D or Vss power line. A smaller voltage variation on the power grid indicates that the circuit has better performance. 69 Time (ns) Figure 5.8: Simulation results for no decap inserted. 1.4 Time (ns) Figure 5.9: Simulation results for standard decaps inserted. 70 Time (ns) Figure 5.11: Simulation results for Fujitsu APS inserted. 71 The simulation results of Figures 5.8 to 5.12 are consistent with the expected performance of the circuits described in the earlier sections. The standard decap helps to reduce noise to a certain extent, but the active decap designs provide better performance. From another perspective, assuming the area is fixed, the active designs have better area/noise efficiency. However, the noise reduction performance of each circuit improves at the expense of increasing power consumption. A standard decoupling capacitor ideally consumes zero power i f not considering gate leakage, but its noise regulation performance is the lowest. Sun's voltage regulator, on the other hand, is the most effective design in terms of power-grid noise reduction, but it requires the most dc power dissipation. Fujitsu's active power stabilizer and the low-power voltage regulator lie somewhere in between the two extremes. 72 The detailed dc power numbers are as follows: The Sun V R draws roughly 25mA of dc current in standby, which corresponds to about 30mW of power for 0.13um technology. The Fujitsu APS consumes 250uA of dc power, only about 1% of the Sun VR. The power dissipation for the low-power V R is approximately 2.6mA (3.0mW), 90% less than the Sun VR. The simulation uses a 400 MHz clock to switch a large buffer connected to the power grid. Under this situation, the regulation performance between the low-power V R and Sun V R is reasonably close. It is shown that for operating at a few hundred megahertz range, the low-power version performs well in a 0.13um simulation process. 73 Chapter 6 Conclusions and Future Work 6.1 Summary As technology scales further into the deep submicron regime, with increasing clock frequency and decreasing supply voltage, maintaining the quality of power supply becomes a critical issue. On-chip power supply noise, due to IR drop and Ldi/dt effects, has a great impact on delay variation, and may even cause improper functionality. Power supply noise can be reduced by placing decoupling capacitors close to power pads and large drivers throughout the power distribution system. Decaps provide instantaneous current to the switching drivers and keep the power supply within certain noise budgets. Traditionally, a standard decap is made from an NMOS transistor outside the standard-cell blocks, or a pair of NMOS and PMOS transistors within the blocks. However, starting from 90nm technology, the oxide thickness of MOS transistors is reduced to approximately 2.0nm or less, resulting in increased ESD risk and gate leakage. Standard decap designs, therefore, may no longer be appropriate for 90nm and below because they suffer greatly from these two problems. 74 In this thesis, the goal was to provide practical solutions to decap design for present-day and upcoming technologies. The thesis began with an overview of decap modeling, gate leakage phenomenon, ESD occurrence, and basic decap layout knowledge. Some essential decap design issues were highlighted through the background discussion to motivate the topics in the rest of the thesis. Next, a number of design approaches for decaps in recent technologies were described along with their advantages and disadvantages. The approaches from circuit level, including cross- coupled decap, gated decap, and thick-oxide decaps were discussed first. The use of high-k gate dielectrics and M I M capacitors were also described. In order to make further improvement from the existing design approaches, the cross-coupled decap design was chosen because of its use in existing libraries. In the basic cross-coupled design, the tradeoff between ESD reliability and transient response is a key issue. The objective was to achieve gate leakage savings while keeping a reasonable tradeoff between ESD and transient response. This thesis proposed three modifications of the basic cross-coupled design. Among the three, Mod2 is designed to replace the cross-coupled design for reduced leakage; Modi has the best ESD performance; Mod3 provides better transient response and the least gate leakage. Finally, the designs of active decoupling capacitors for power-grid noise reduction were investigated. The switched decaps amplify the charge storage capacity of the basic decap while monitoring the power rail activity to provide dynamic control of the switching response. The 75 switched decaps have better area efficiency and better noise reduction performance than the passive decaps. It was observed that the Sun Voltage Regulator (VR) performs well but dissipates excessive power, whereas the Fujitsu Active Power Stabilizer (APS) saves power but experiences excessively long delays. A new low-power switched-decap voltage regulator was proposed to make design tradeoffs between power and performance. The low-power V R adopts novel amplification circuitry to control its power consumption while providing a reasonable swing at the output. Its noise reduction performance is acceptable and close to the Sun V R when operated in moderate frequencies. 6.2 Contributions in this Thesis The following summarizes the major contributions in this thesis: • Developed practical decap layouts that properly tradeoff between transient response performance, ESD reliability, and gate leakage; • Designed a low-power voltage regulator using switched decaps that provides adequate power-noise reduction performance while consuming relatively low standby power. 6.3 Future Work A number of issues regarding decoupling capacitors will have to be addressed in the near future. First, knowing that the thin-oxide decaps leak a significant amount of current in 90nm and below, it is important to place only the necessary amount of decaps in a certain design to avoid overdesign. The use of thick-oxide decaps may not solve the issue completely because the effective capacitance is much less for thick-oxide devices and the total free area for decaps is limited. Also, the active decaps provide better noise reduction performance but at a cost of 76 increased standby power requirement, compared to the passive decaps. Therefore, to determine the optimal number of thick-oxide, thin-oxide and active decaps to be placed into a design remains a challenge. Another issue would be the placement of decaps. The proper placement and use of active decaps versus passive decaps is still under investigation. The presence of power-grid noise is indeed a two-dimensional problem. The noise is related to logic block, clock tree and power mesh distribution throughout the chip. Hence, the optimal placement of decaps must consider the placement of other functional blocks. 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