INVESTIGATION OF HIGH PERFORMANCE SINGLE-PHASE SOLUTIONS FOR AC-DC POWER FACTOR CORRECTED BOOST CONVERTERS by Fariborz Musavi M.A.Sc., Concordia University, 2001 B.Sc., Iran University of Science and Technology, 1994 A THESIS SUBMITTED IN PARTIAL FULFILLMENT OF THE REQUIREMENTS FOR THE DEGREE OF DOCTOR OF PHILOSOPHY in The Faculty of Graduate Studies (Electrical and Computer Engineering) THE UNIVERSITY OF BRITISH COLUMBIA (Vancouver) May 2011 © Fariborz Musavi, 2011 Abstract Plug-in Hybrid Electric Vehicles (PHEVs) and Electric Vehicles (EVs) are an emerging trend in automotive circles, and consumer’s interest is growing rapidly. With the development of PHEVs, battery chargers for automotive applications are becoming a large market for the power supply industries. The improvement of overall charger efficiency is critical for the emergence and acceptance of these vehicular technologies, as the charger efficiency increases, the charge time and utility cost decreases. Additionally, to meet the efficiency and power factor requirements and regulatory standards for the AC supply mains, power factor correction is essential. Due to limited space in vehicle and increasing power consumption, chargers are required to deliver more power with smaller volume. As a key component of a charger system, the frontend AC-DC converter must achieve high efficiency and high power density. In this dissertation, several conventional plug in hybrid electric vehicle charger front end AC-DC converter topologies are investigated and a new bridgeless interleaved and a phase shifted semi-bridgeless power factor corrected converter are proposed to improve the efficiency and performance, which is critical to minimize the charger size, charging time, and the amount and cost of electricity drawn from the utility. A detailed analytical model for these topologies is developed, enabling the calculation of power losses and efficiency. Experimental and simulation results of several prototype boost converter converting universal AC input voltage to 400 V DC at 3.4 kW are given to verify the proof of concept, and analytical work reported in this thesis. The results show a power factor greater than 0.99 from 750 W to 3.4 kW, THD less than 5% from half load to full load and a peak efficiency of 98.94 % at 265 V input and 1200 W load. ii Preface This thesis contains four chapters that present results that have been published or submitted for consideration in the form of IEEE refereed papers and scientific journals of which I am the lead author. The initial project overview was proposed by me. For the scientific journal submissions and papers, I performed all the research, data analyses, and interpretation of the results, and prepared the final manuscripts. Co-authors provided advice on methodology and made editorial comments as required. The complete citations for these papers and the chapters in which they appear are provided as follows: Chapter 2 is based on: [1] F. Musavi ; W. Eberle ; W.G. Dunford, "A High-Performance Single-Phase AC-DC Power Factor Corrected Boost Converter for plug in Hybrid Electric Vehicle Battery Chargers," Published in IEEE Energy Conversion Congress and Exposition, Atlanta, Georgia, 2010. [2] F. Musavi ; W. Eberle ; W.G. Dunford, "Efficiency Evaluation of Single-Phase Solutions for AC-DC PFC Boost Converters for Plug-in-Hybrid Electric Vehicle Battery Chargers," Published in IEEE Vehicle Power and Propulsion Conference, Lille, France, 2010. [3] F. Musavi ; W. Eberle ; W.G. Dunford, "A High-Performance Single-Phase Bridgeless Interleaved PFC Converter for Plug-in Hybrid Electric Vehicle Battery Chargers," IEEE Transactions on Industry Applications, in press, 2010-IPCC-417.R1 (T-IA). Chapter 3 is based on: [4] F. Musavi ; W. Eberle ; W.G. Dunford, "A Phase Shifted Semi-Bridgeless Boost Power Factor Corrected Converter for Plug in Hybrid Electric Vehicle Battery Chargers," Published in IEEE Applied Power Electronics Conference and Exposition, APEC, Fort Worth, TX, 2011. iii Chapter 4 is based on: [5] F. Musavi ; M. Edington ; W. Eberle ; W.G. Dunford, "Effect of Ripple Steering Technique on Control Loop Stability of A CCM PFC Boost Converter," Published in IEEE Energy Conversion Congress and Exposition, Phoenix, AZ, 2011. Part of Chapter 1, Chapter 2 and Chapter 3 are based on: [6] F. Musavi ; M. Edington ; W. Eberle ; W.G. Dunford, "Energy Efficiency in Plug in Hybrid Electric Vehicle Chargers: Evaluation and Comparison of Front End AC-DC Topologies," Published in IEEE Energy Conversion Congress and Exposition, Phoenix, AZ, 2011. [7] D. Gautam ; F. Musavi ; M. Edington ; W. Eberle ; W.G. Dunford, " An Automotive OnBoard 3.3 kW Battery Charger for PHEV Application," Published in IEEE Vehicle Power and Propulsion Conference, Chicago, IL, 2011. iv Table of Contents Abstract................................................................................................................................................... ii Preface ................................................................................................................................................... iii Table of Contents .................................................................................................................................... v List of Tables ....................................................................................................................................... viii List of Figures ........................................................................................................................................ ix List of Symbols .................................................................................................................................... xiv Acknowledgements ............................................................................................................................. xvii Dedication .......................................................................................................................................... xviii CHAPTER 1. Introduction .................................................................................................................. 1 1.1. Introduction ............................................................................................................................. 1 1.2. General Background................................................................................................................. 3 1.3. Literature Review..................................................................................................................... 6 1.3.1. Conventional PFC Boost Converter .................................................................................. 8 1.3.2. Bridgeless PFC Boost Converter ..................................................................................... 11 1.3.2.1. Positive "HALF Cycle" Operation .............................................................................. 12 1.3.2.2. Negative "HALF Cycle" Operation ............................................................................. 12 1.3.2.1. PSIM Simulation of Bridgeless Boost Converter ......................................................... 14 1.3.3. Semi-Bridgeless PFC boost Converter ............................................................................ 15 1.3.4. Interleaved PFC Boost Converter .................................................................................... 16 1.3.4.1. PSIM Simulation of Interleaved Boost Converter ........................................................ 18 1.3.5. Ripple Steering Technique in PFC Applications.............................................................. 19 1.3.6. Average Switch Model ................................................................................................... 21 1.4. Motivation and Objectives of the Thesis ................................................................................. 22 1.5. Outline of the Thesis .............................................................................................................. 23 CHAPTER 2. Bridgeless Interleaved PFC Boost Converter ............................................................... 24 2.1. Introduction ........................................................................................................................... 24 2.2. Circuit Description and Steady State Analysis ........................................................................ 24 2.2.1. Positive Half Cycle Operation ........................................................................................ 25 2.2.2. Negative Half Cycle Operation ....................................................................................... 25 2.2.3. Detailed Positive Half Cycle Operation and Analysis for D > 0.5 .................................... 26 v 2.2.4. Detailed Positive Half Cycle Operation and Analysis for D < 0.5 .................................... 30 2.3. Loss Modeling ....................................................................................................................... 35 2.4. Math Modeling Verification ................................................................................................... 41 2.5. Simulation Results ................................................................................................................. 45 2.6. Logic Implementation ............................................................................................................ 47 2.7. Experimental Results.............................................................................................................. 49 2.8. Conclusion ............................................................................................................................. 57 CHAPTER 3. Phase-Shifted Semi Bridgeless PFC Converter ............................................................ 59 3.1. Introduction ........................................................................................................................... 59 3.2. Circuit Description and Steady State Analysis ........................................................................ 59 3.2.1. Positive Half Cycle Operation ........................................................................................ 60 3.2.2. Negative Half Cycle Operation ....................................................................................... 61 3.2.1. Detailed Positive Half Cycle Operation and Analysis for D > 0.5 .................................... 61 3.2.1. Detailed Positive Half Cycle Operation and Analysis for D < 0.5 .................................... 64 3.3. Loss Modeling ....................................................................................................................... 66 3.4. Simulation Results ................................................................................................................. 68 3.5. Logic Implementation ............................................................................................................ 70 3.6. Experimental Results.............................................................................................................. 71 3.7. Conclusion ............................................................................................................................. 82 CHAPTER 4. The Ripple Steering Technique and Converter Modeling ............................................ 83 4.1. Introduction ........................................................................................................................... 83 4.2. Average Switch Model ........................................................................................................... 83 4.2.1. PWM Switch Model of Conventional Boost Converter ................................................... 83 4.2.1. Feedback Compensation Design of Conventional Boost Converter ................................. 84 4.3. Ripple Steering Techniques in PFC Application ..................................................................... 88 4.3.1. PWM Switch Model of Boost Converter with Coupled Inductor ..................................... 89 4.3.1. Feedback Compensation Design of Boost Converter with Coupled Inductor ................... 92 4.4. Simulation Results ................................................................................................................. 93 4.5. Experimental Results.............................................................................................................. 95 4.6. Conclusion ........................................................................................................................... 102 CHAPTER 5. 5.1. Conclusions and Future Work ................................................................................... 103 Conclusions ......................................................................................................................... 103 vi 5.1.1. Bridgeless Interleaved Boost PFC Converter................................................................. 103 5.1.2. Phase Shifted Semi-Bridgeless Boost PFC Converter .................................................... 103 5.1.3. New Loss Modeling for PFC Boost Converters ............................................................. 104 5.1.1. New Average Modeling for PFC Boost Converters with Coupled Inductors .................. 104 5.2. Future Work ......................................................................................................................... 105 5.2.1. LLC Resonant Converter for DC/DC Stage................................................................... 105 5.2.2. Resonant PFC Converter .............................................................................................. 105 5.2.3. Level 3 Chargers .......................................................................................................... 105 5.2.4. Wireless Chargers......................................................................................................... 106 Bibliography ....................................................................................................................................... 107 Appendix ............................................................................................................................................ 112 vii List of Tables Table 2-1: Summary of component RMS current for conventional boost and bridgeless boost topologies ...................................................................................................................................................... 37 Table 2-2: Summary of component RMS current for interleaved and bridgeless interleaved boost topologies ...................................................................................................................................... 37 Table 2-3: Summary of component average current for conventional boost and bridgeless boost topologies ...................................................................................................................................... 38 Table 2-4: Summary of component average current for interleaved and bridgeless interleaved boost topologies ...................................................................................................................................... 38 Table 2-5: Component / devices used in prototype unit .......................................................................... 52 Table 3-1: Component / devices used in prototype unit .......................................................................... 71 viii List of Figures Figure 1-1: Simplified system block diagram of a universal two-stage battery charger ............................. 3 Figure 1-2: Passive power factor correction AC main voltage and current waveforms .............................. 5 Figure 1-3: Conventional PFC boost converter......................................................................................... 8 Figure 1-4: Simplified block diagram of the boost PFC circuit ................................................................. 9 Figure 1-5: Inductor current and the duty cycle in a typical PFC boost converter in CCM ...................... 10 Figure 1-6: Top: Transistor current - Bottom: Diode current in a typical PFC boost converter in CCM... 10 Figure 1-7: Bridgeless PFC boost topology ............................................................................................ 11 Figure 1-8: Bridgeless PFC converter operation ..................................................................................... 13 Figure 1-9: PSIM simulation circuit for bridgeless PFC boost converter: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz .............................................................................................................. 14 Figure 1-10: Simulation waveforms for bridgeless PFC boost converter including output voltage, input voltage and input current: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz ....................... 15 Figure 1-11: Bridgeless PFC boost topology .......................................................................................... 16 Figure 1-12: Interleaved PFC boost topology ......................................................................................... 17 Figure 1-13: PSIM simulation circuit for interleaved PFC boost converter: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz .............................................................................................................. 18 Figure 1-14: Simulation waveforms for interleaved PFC boost converter including output voltage, input voltage and input current: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz ....................... 19 Figure 1-15: Smoothing transformer in a coupled filter .......................................................................... 20 Figure 1-16: Modified PFC boost converter with coupled inductors ....................................................... 20 Figure 1-17: One active switch and one passive switch .......................................................................... 21 Figure 1-18: PWM-switch ..................................................................................................................... 21 Figure 1-19: Averaged model of PWM-switch ....................................................................................... 21 Figure 2-1: Proposed bridgeless interleaved (BLIL) PFC boost converter ............................................... 25 Figure 2-2: Interval 1: Q1 and Q2 are “ON”, and body diode of Q4 conducting ..................................... 26 Figure 2-3: Interval 2 and 4: Q1, Q2, Q3 and Q4 are “ON” .................................................................... 26 Figure 2-4: Interval 3: Q3 and Q4 are “ON”, and body diode of Q2 conducting ..................................... 27 Figure 2-5: BLIL PFC boost converter steady-state waveforms at D > 0.5.............................................. 28 Figure 2-6: Interval 1 and 3: Body diodes of Q2 and Q4 conducting ...................................................... 31 Figure 2-7: Interval 2: Q1 and Q2 are “ON”, and body diode of Q4 conducting ..................................... 31 Figure 2-8: Interval 4: Q3 and Q4 are “ON”, and body diode of Q2 conducting ..................................... 31 Figure 2-9: BLIL PFC boost converter steady-state waveforms at D < 0.5.............................................. 33 ix Figure 2-10: RMS ripple current through output capacitors vs input voltage at Po = 3400 W, fsw = 70 kHz ............................................................................................................................................... 39 Figure 2-11: Comparison of the estimated loss distribution in the semiconductors at 70kHz switching frequency, 240V input, 3400W load at 400V ................................................................................. 40 Figure 2-12: Simulation and math modeling of inductor RMS current for an interleaved PFC converter . 42 Figure 2-13: Simulation and math modeling of MOSFET RMS current for an interleaved PFC converter ...................................................................................................................................................... 42 Figure 2-14: Simulation and math modeling of boost diode average current for an interleaved PFC converter ....................................................................................................................................... 43 Figure 2-15: Simulation and math modeling of inductor RMS current for a bridgeless PFC converter .... 43 Figure 2-16: Simulation and math modeling of MOSFET RMS current for a bridgeless PFC converter .. 44 Figure 2-17: Simulation and math modeling of boost diode average current for a bridgeless PFC converter ...................................................................................................................................................... 44 Figure 2-18: Simulation and math modeling of MOSFET body diode average current for a bridgeless PFC ...................................................................................................................................................... 45 Figure 2-19: PSIM simulation circuit for the proposed BLIL PFC boost converter: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz ............................................................................................ 46 Figure 2-20: Simulation waveforms for the proposed BLIL PFC boost converter including output voltage, input voltage and input current: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz .............. 47 Figure 2-21: Logic implementation of BLIL PFC boost converter using UCC28070 controller .............. 48 Figure 2-22: Breadboard prototype of BLIL PFC boost converter .......................................................... 49 Figure 2-23: Breadboard proposed BLIL PFC experimental waveforms; ................................................ 50 Figure 2-24: Gating signal, boost MOSFET and diode current for D > 0.5 ............................................. 50 Figure 2-25: Gating signal, boost MOSFET and diode current for D < 0.5 ............................................. 51 Figure 2-26: Load transient response from FL to NL (3400W to 0W) .................................................... 51 Figure 2-27: Load transient response from NL to FL (0W to 3400 W).................................................... 52 Figure 2-28: Input current harmonics at full load for Vin = 120 V at Po = 1700 W and 240 V at Po = 3400 W .................................................................................................................................................. 53 Figure 2-29: RMS Total harmonics distortion vs. output power at Vin = 120V and Vin = 240V ............. 53 Figure 2-30: Power factor vs. output power at Vin = 120V and Vin = 240V ........................................... 54 Figure 2-31: Efficiency vs. output power for the proposed bridgeless interleaved PFC boost converter .. 54 Figure 2-32: Efficiency vs. output power for different topologies at Vin = 240V .................................... 55 Figure 2-33: Loss reduction as a function of output power at Vin = 240V, Vo = 400 V and 70 kHz for the proposed BLIL converter compared to the benchmark interleaved boost converter ......................... 56 x Figure 2-34: Loss reduction as a function of output power at Vin = 120V, Vo = 400 V and 70 kHz for the proposed BLIL converter compared to the benchmark interleaved boost converter ......................... 57 Figure 3-1: Phase shifted semi-bridgeless PFC boost topology ............................................................... 60 Figure 3-2: Interval 1and 3: Q1 and Q2 are ON...................................................................................... 61 Figure 3-3: Interval 2: Q1 ON, body diode of Q2 conducting ................................................................. 62 Figure 3-4: Interval 4: Q1 OFF and Q2 ON............................................................................................ 62 Figure 3-5: Phase shifted semi-bridgeless boost converter steady-state waveforms at D > 0.5................. 63 Figure 3-6: Interval 1and 3: Q1 and Q2 are OFF, body diode of Q2 conducting ..................................... 64 Figure 3-7: Interval 2: Q1 ON, body diode of Q2 conducting ................................................................. 64 Figure 3-8: Interval 4: Q1 OFF and Q2 ON............................................................................................ 64 Figure 3-9: Phase shifted semi-bridgeless boost converter steady-state waveforms at D < 0.5................. 65 Figure 3-10: Comparison of the estimated loss distribution in the semiconductors at 70kHz switching frequency, 240V input, 3300W load at 400V ................................................................................. 67 Figure 3-11: PSIM simulation circuit for the phase shifted semi-bridgeless PFC boost converter: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz ......................................................................... 68 Figure 3-12: Simulation waveforms for the proposed phase shifted semi-bridgeless PFC boost converter: Output voltage, input voltage and input current: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz ............................................................................................................................................... 69 Figure 3-13: Logic implementation of phase shifted semi-bridgeless PFC boost converter using UCC28070 controller ..................................................................................................................... 70 Figure 3-14: Capacitor bank of 820 uF................................................................................................... 72 Figure 3-15: Control board of PFC boost converter ................................................................................ 72 Figure 3-16: IMS power board attached to a heatsink with the PFC inductors ........................................ 73 Figure 3-17: Efficiency as a function of output power at Vin = 240V, Vo=400V and 70kHz switching .. 74 Figure 3-18: Loss reduction as a function of output power at Vin = 240V, Vo=400V and 70kHz for the proposed phase shifted semi-bridgeless converter compared to the benchmark interleaved boost converter ....................................................................................................................................... 74 Figure 3-19: Efficiency as a function of output power at Vin = 120V, Vo=400V and 70kHz .................. 75 Figure 3-20: Loss reduction as a function of output power at Vin = 120V, Vo=400V and 70kHz for the proposed phase shifted semi-bridgeless converter compared to the benchmark interleaved boost converter ....................................................................................................................................... 75 Figure 3-21: THD as a function of output power at Vin = 120 V and 240V, Vo=400V and 70kHz ......... 76 Figure 3-22: Power factor as a function of output power at Vin = 120 V and 240V, Vo=400V and 70kHz ...................................................................................................................................................... 76 xi Figure 3-23: Harmonics orders at Vin = 120 V and 240V, compared against EN61000-3-2 standard for Vin = 120 V at Po = 1700 W and 240 V at Po = 3400 W ................................................................ 77 Figure 3-24: Input current, input voltage and output voltage. ................................................................. 78 Figure 3-25: Input current, inductor current, input voltage and output voltage. ....................................... 79 Figure 3-26: Inductor current, input current and sensed MOSFET current. ............................................. 79 Figure 3-27: Gating signal, inductor and sensed MOSFET current for D < 0.5 ....................................... 80 Figure 3-28: Gating signal, inductor and sensed MOSFET current for D > 0.5 ....................................... 80 Figure 3-29: Load transient response from NL to FL (0W to 3400 W).................................................... 81 Figure 3-30: Load transient response from FL to NL (3400 W to 0 W)................................................... 81 Figure 4-1: Conventional boost converter with PWM switch.................................................................. 84 Figure 4-2: Current loop plant and compensator type II Bode plots ........................................................ 85 Figure 4-3: Type II compensator network .............................................................................................. 85 Figure 4-4: Open loop Bode plot for current loop ................................................................................... 86 Figure 4-5: Voltage loop plant and compensator type II Bode plots ........................................................ 87 Figure 4-6: Open loop Bode plot for voltage loop .................................................................................. 88 Figure 4-7: Modified PFC boost converter with coupled inductors ......................................................... 88 Figure 4-8: Equivalent circuit of coupled inductors ................................................................................ 89 Figure 4-9: Equivalent circuit of coupled inductors ................................................................................ 89 Figure 4-10: Modified boost converter with PWM switch ...................................................................... 90 Figure 4-11: Current loop plant Bode plots ............................................................................................ 91 Figure 4-12: Current loop plant and compensator Bode plots for boos converter with coupled inductor .. 92 Figure 4-13: Open loop plant Bode plot for boost converter with coupled inductor ................................. 93 Figure 4-14: PSIM simulation circuit for ripple steering technique applied to PFC boost converter ........ 94 Figure 4-15: Inductor Current - no filtering technique (Top) – ripple steering technique (Middle), series capacitor current (Bottom): Vin = 240 V, Vo = 400 V, Po = 1600 W, fsw = 70 kHz ....................... 94 Figure 4-16: Coupled inductors used in experimental circuit - Top left: DC inductor, Top right: AC inductor, Bottom: Coupled inductor ............................................................................................... 95 Figure 4-17: Inductor current Idc ripple at 120 V input and 800 W output - Experimental ...................... 96 Figure 4-18: Inductor current Idc ripple at 120 V input and 800 W output - Simulation .......................... 96 Figure 4-19: Inductor current Iac at 120 V input and 800 W output - Experimental ................................ 97 Figure 4-20: Inductor current Iac at 120 V input and 800 W output - Simulation .................................... 97 Figure 4-21: Peak inductor current Iac ripple at 120 V input and 800 W output - Experimental .............. 98 Figure 4-22: Peak inductor current Iac ripple at 120 V input and 800 W output - Simulation .................. 98 Figure 4-23: Inductor current Idc ripple at 240 V input and 1600 W output - Experimental .................... 99 xii Figure 4-24: Inductor current Idc ripple at 240 V input and 1600 W output - Simulation ........................ 99 Figure 4-25: Inductor current Iac at 240 V input and 1600 W output - Experimental ............................ 100 Figure 4-26: Inductor current Iac at 240 V input and 1600 W output - Simulation ................................ 100 Figure 4-27: Peak inductor current Iac ripple at 240 V input and 1600 W output - Experimental .......... 101 Figure 4-28: Peak inductor current Iac ripple at 240 V input and 1600 W output - Simulation .............. 101 xiii List of Symbols CO Output Capacitor dB/dec Decibel per Decade δD Boost Diode Duty Cycle δQ Boost Main Switch Duty Cycle Irms Total Line Current I1rms Fundamental Component Kd Distortion Factor Kθ Displacement Factor Rs Current Sense Resistor TS One Switching Period θ1 Angle between Current Fundamental and Sinusoidal Line Voltage VS Maximum Input Voltage Vref Reference Voltage Abbreviations AC Alternating Current ADC Analog to Digital Converter ASIC Application Specific Integrated Circuits BJT Bipolar Junction Transistor BL Bridgeless BLIL Bridgeless Interleaved CCM Continuous Conduction Mode CRM Critical Conduction Mode D Duty Cycle xiv DC Direct Current DCM Discontinuous Conduction Mode DSP Digital Signal Processors FL Full Load FPGA Field Programmable Gate Array IC Integrated Circuits IGBT Insulated Gate Bipolar Transistor IL Interleaved IMS Insulated Metal Substrate EMI Electrical Magnetic Interference EV Electric Vehicle HS High Side LLC Inductor Inductor Capacitor M Mutual Inductance MOSFET Metal Oxide Silicon Field Effect Transistor NL No Load PID Proportional Integral Derivative PCB Printed Circuit Board PF Power Factor PFC Power Factor Correction PHEV Plug in Hybrid Electric Vehicle PSFB Phase Shift Full-Bridge PWM Pulse Width Modulation RLC Resistance Inductor Capacitor RMS Root Mean Square SR Synchronous Rectifier xv THD Total Harmonics Distortion ZCS Zero Current Switching ZVS Zero Voltage Switching Prefixes for SI Units p Pico (10-12) n Nano (10-9) µ Micro (10-6) m Milli (10-3) k Kilo (103) M Mega (106) G Giga (109) SI Units A Amperes C Coulombs F Farads H Henries Hz Hertz s seconds V Volts W Watts ° Degrees Ω Ohms xvi Acknowledgements I would like to thank my supervisors, Dr. William G. Dunford and Dr. Wilson Eberle for their guidance, encouragement and continuous support through the course of this work. Their knowledge, research attitude and ways of thinking are greatly appreciated. Financial and technical support in the form of project funding, lab equipment and tuition fee reimbursement from Delta-q Technologies Corp. is greatly acknowledged and appreciated. I would like to thank all of my past and present colleagues from Delta-q Technologies. In particular, I would like to thank Ken Fielding, President & CEO, Rob Cameron, former CTO and VP of Engineering, Art Gau, Senior Design Engineer and Deepak Gautam Power Electronics Engineer at Delta-Q Technologies Corp. for all their support and valuable discussions on several topics. It has been a great pleasure to work in the UBC Electric Power and Energy Systems Group. I would like to acknowledge the group administrative and management staff. I would like to give my special thanks to all my family; their love and support make my life more colorful and meaningful. xvii Dedication To my family xviii CHAPTER 1. Introduction 1.1. Introduction As the demand for energy drastically increased in the 20th Century, fossil fuels became the main source of energy due to convenience and cost. Over the years, however, the price of oil and problems caused by pollution, have increased considerably, putting pressure on governments and industries to invest on other solutions to replace fossil fuels. Consequently, interest in other means of transportation, such as Plug in Hybrid Electric Vehicles (PHEV) and Electric Vehicles (EV), has increased again. EV and PHEV technology has existed since the early 1900s. However, the high cost and low energy density of available energy storage systems, primarily batteries, along with the very low cost of oil, had limited the interest in EV and PHEV. Recent innovations in lithium-ion batteries, the higher price of gas, and the air pollution associated with fossil fuels have significantly impacted the alternative transportation industry. As the adoption rate of these vehicles increases, the stress on the utility grid is projected to increase significantly at times of peak demand [1]. Therefore, efficient and high power factor charging is critical in order to minimize the utility load stress, and reduce the charging time. In addition, a high power factor is needed to limit the input current harmonics drawn by these chargers and to meet regulatory standards, such as IEC 61000-3-2 [2]. A PHEV is a hybrid vehicle with a storage system that can be recharged by connecting a plug to an external electric power source. The charging AC outlet inevitably needs an on-board AC/DC charger with a power factor correction. An on-board 3.5 kW charger could charge a depleted battery pack in PHEVs to 95% charge in about four hours from a 240 V supply [3]. 1 Chargers are also classified by the level of power they can provide to the battery pack [4]: • Level 1: Common household circuit, rated to 120 volts AC and 15 amperes. These chargers use the standard three-prong household connection, and they are usually considered portable equipment. • Level 2: Permanently wired electric vehicle supply equipment used especially for electric vehicle charging; rated up to 240 volts AC, up to 60 amps, and up to 14.4 kilowatts. • Level 3: Permanently wired electric vehicle supply equipment used especially for electric vehicle charging; rated greater than 14.4 kW. Fast chargers are rated as Level 3, but not all Level 3 chargers are fast chargers. This designation depends on the size of the battery pack to be charged and how much time is required to charge the battery pack. A charger can be considered a fast charger if it can charge an average electric vehicle battery pack in 30 minutes or less. The front-end AC-DC converter is a key component of the charger system. The purpose of this document is to illustrate how this research will be conducted on the high-performance single-phase solutions for AC-DC power factor corrected converters for PHEV battery chargers. A variety of circuit topologies and control methods have been developed for the PFC application [5-7]. The single-phase active PFC techniques can be divided into two categories: the single-stage approach and the two-stage approach. The single-stage approach is suitable for low power applications. In addition, due to large low frequency ripple in the output current, only lead acid batteries are chargeable. Therefore, the two-stage approach is the proper candidate for PHEV battery chargers [8], where the power rating is relatively high, and lithium-ion batteries 2 are used as the main energy storage system. The front end PFC section is then followed by a DC/DC section to complete the charger system. Figure 1-1 illustrates a simplified block diagram of a universal input two-stage battery charger used for PHEVs. Figure 1-1: Simplified system block diagram of a universal two-stage battery charger The PFC stage rectifies the input AC voltage and transfers it into a regulated intermediate DC link bus. At the same time, power factor correction function is achieved. The following DC/DC stage then converts the DC bus voltage into a regulated output DC voltage for charging batteries, which is required to meet the regulation and transient requirements. 1.2. General Background According to the requirements of input current harmonics [9] and output voltage regulation, a front-end converter is normally implemented by a power factor correction (PFC) stage. Conventionally, most of the power conversion equipment employs either a diode rectifier or a thyristor rectifier with a bulk capacitor to converter AC voltage to DC voltage before processing it [10]. Such rectifiers produce input current with rich harmonic content, which pollute the power system and the utility lines. Power quality is becoming a major concern for many electrical users. The simplest form of PFC is passive (Passive PFC). A passive PFC uses a filter at the AC input to correct poor power factor. The passive PFC circuitry uses only passive components - an 3 inductor and some capacitors. Although pleasantly simple and robust, a passive PFC rarely achieves low Total Harmonic Distortion (THD). Also, because the circuit operates at the low line power frequency of 50Hz or 60Hz, the passive elements are normally bulky and heavy. Figure 12 shows input voltage and current for a passive PFC, and the harmonic spectrum of input current. The input power factor (PF) is defined as the ratio of the real power over apparent power as: 1-1 Assuming an ideal sinusoidal input voltage source, the power factor can be expressed as the product of two factors, the distortion factor and the displacement factor, as given: 1-2 The distortion factor Kd is the ratio of the fundamental RMS current to the total RMS current. Input Current Input Voltage a) Input voltage and Input current 4 st 1 Fundamental rd 3 Harmonic 5th Harmonic b) Harmonic Spectrum of Input Current Figure 1-2: Passive power factor correction AC main voltage and current waveforms The displacement factor, Kθ, is the cosine of the displacement angle between the fundamental input current and the input voltage fundamental RMS current. !"# !"# $%& 1-3 1-4 where I1rms is the fundamental component of the line current, Irms is the total line current, and θ1 is the phase shift of the current fundamental relative to the sinusoidal line voltage. The distortion factor is close to unity, even for waveforms with noticeable distortion; therefore, it is not a very convenient measure of distortion for practical use. The distortion factor is uniquely related to another figure of merit; the total harmonic distortion (THD): '() * !"# + , !"# + !"# + *&-./0+ & 1-5 1-6 5 Kd is regulated by IEC 1000-3-2 [2] for lower power levels and by IEEE Std 519-1992 [11] for higher power levels, where Kθ is regulated by utility companies [12]. Significant reduction of current harmonics in single-phase circuits can only be achieved by using rectifiers based on pulse width modulated (PWM) switching converters. These converters can be designed to emulate a resistive load and, therefore, produce very little distortion of the current. By using PWM or other modulation techniques, these converters draw a nearly sinusoidal current from the ac line in phase with the line voltage. As a result, the rectifier operates with very low current harmonic distortion and very high, practically unity power factor. This technique is commonly known as power factor correction (PFC). As a result of this research, the existing PFC technology based on the boost converter topology with averagecurrent-mode control was significantly improved. The proposed improvements allowed an extended range of operating conditions and additional functionality. 1.3. Literature Review All basic power converter topologies, such as boost, buck [13], buck-boost, and their variations, can be used to realize active PFC techniques [14, 15]. At lower power ratings, MOSFETs [16] are the switching power devices of choice because of their low conduction losses and high switching speed. For medium and high-power applications, IGBTs can be used in PWM-controlled converters with switching frequency of up to 30 kHz. There are many integrated circuits (ICs) [17] on the market that incorporate control functions for PFC converters and facilitate compact and cost-effective designs. Digital signal processors (DSPs) [18-21] and microcontrollers [22] have been successfully used to control PFC converters. Microcontrollers and DSPs can be used to realize traditional [23, 24] proportional-integral derivative (PID) control laws as well as non-traditional control principles, such as sliding mode control [25], fuzzy logic 6 [26], and neural networks [27]. One of the drawbacks of using microcontrollers and DSPs is the significant effort that goes into software development. The boost topology is by far the most popular topology in PFC applications. The boost PFC converter draws a continuous current from the line and, therefore, does not require much filtering, which is usually accomplished by an input filter capacitor. Other topologies such as buck, buck-boost, and flyback draw pulsed current and need a much better input filter. Unlike the buck topology, the boost converter easily accommodates the input voltage range, from zero to the line peak voltage. The boost converter can operate in continuous conduction mode (CCM) [28], discontinuous conduction mode (DCM), or critical conduction mode (CRM). These names refer to the continuity of the inductor current within the switching cycle [29]. The boost converter operating in DCM and CRM modes is usually easier to control, but it has higher peak-to-peak current ripple, which causes higher RMS value of the inductor current, higher magnetic and conduction losses, and higher switching noise, which leads to increased filtering requirements. Therefore, these modes are restricted to relatively low power levels, while the CCM is used at medium and high power levels. While the discontinuous conduction mode (DCM) converters such as boost and flyback converters are well suited for low power applications, continuous conduction mode (CCM) boost converters with average current mode [30], peak current mode [31] or hysteresis control [32] are commonly chosen for many medium and high power applications. The output voltage of the boost PFC converter should be always higher than the peak line voltage. For universal line application (85 V-265 V), the output voltage is usually set around 400 VDC [33]. 7 The boost circuit-based PFC topology operated in CCM is employed in this study as the main candidate for front end single-phase solutions for AC-DC power factor corrected converters used in PHEV battery chargers. 1.3.1. Conventional PFC Boost Converter The conventional boost topology is the most popular topology for PFC applications. It uses a dedicated diode bridge to rectify the AC input voltage to DC, which is then followed by the boost section, as shown in Figure 1-3. Figure 1-3: Conventional PFC boost converter The simplified block diagram of the boost PFC circuit is shown in Figure 1-4 [34]. This circuit has two control loops: One is the fast acting internal current loop. It defines the input current shape to be sinusoidal and forces it in phase with the input voltage. The other is the external slow voltage loop which regulates the output dc voltage. The voltage loop should not react to the 120Hz rectified mains variations, so its bandwidth is between 10 to 20 Hz. The current loop usually has a bandwidth frequency of less than one tenth of the switching frequency. 8 AC Source & Rectifier Power Stage LB D1 D2 Output Filter, Load & Voltage Divider DB Rvi RL QB Vin Co D3 D4 Rvd Rs Current Sensing PWM Drive Multiplier Full wave Rectified AC Input Sensing Voltage Error Output Voltage Sensing Vref Figure 1-4: Simplified block diagram of the boost PFC circuit The principle operation of boost PFC is as follows: The rectified sinusoidal input voltage goes to a multiplier circuit, providing a current reference to the multiplier and a feedforward signal proportional to the RMS value of the line voltage. The filtered dc output voltage of the boost PFC is compared to a reference voltage, Vref and amplified. The error amplifier senses the variations between the output voltage and the fixed dc reference voltage. The error signal is then applied to the multiplier. The multiplier's output follows the shape of the input ac voltage, with an average value inversely proportional to the RMS value of the ac input voltage. This signal is compared to the current signal sensed by Rs in a pulse-width modulation (PWM) circuit. The inductor current waveform follows the shape of the rectified ac line voltage. The gate drive signal controls the inductor current amplitude and maintains a constant output voltage. As can be seen in Figure 1-5 and Figure 1-6, the inductor current follows the shape of the input voltage. The transistor and diode current waveforms in a typical boost converter in 9 continuous conduction mode of operation are pulsed-width modulated, with both the duty cycle and the peak amplitude varying with the ac input voltage. Boost Converter Inductor Current Boost Converter Duty Cycle Figure 1-5: Inductor current and the duty cycle in a typical PFC boost converter in CCM Boost Transistor Current Boost Diode Current Figure 1-6: Top: Transistor current - Bottom: Diode current in a typical PFC boost converter in CCM The inductor ripple current is directly seen at the converter’s input and will require filtering to meet EMI specifications [35-46]. The diode output current is discontinuous and needs 10 to be filtered out by the output capacitor Co. In this topology, the output capacitor ripple current is very high and its value is the difference between diode current and the dc output current [47]. In practical applications as the power level increases, the diode bridge losses become significant, so dealing with heat dissipation in a limited surface area is important, particularly from an efficiency point of view. Therefore, the conventional PFC boost is limited to a low to a medium power range (e.g. less than 1000 W). 1.3.2. Bridgeless PFC Boost Converter The bridgeless PFC (BL PFC) boost converter avoids the need for the rectifier input bridge yet maintains the classic boost topology, as shown in Figure 1-7 [48-57]. As it is demonstrated in the following, BL PFC does not have any advantages over conventional PFC in terms of passive components sizing. However, it eliminates the input bridge rectifiers, so it is an attractive solution for higher power applications, where the dissipating power in a small area is becoming extremely difficult and overall efficiency is very important. Figure 1-7: Bridgeless PFC boost topology The intrinsic body diode connected between the drain and source of the Power MOSFET switches has an important role in this topology. The circuit shown from a functional point of view is similar to the common boost converter. In the traditional topology current flows through 11 two of the bridge diodes in series. In the bridgeless PFC configuration, current flows through only one diode with the Power MOSFET providing the return path. To analyze the circuit operation, it is necessary to separate it into two sections. The first section operates as the boost stage and the second section operates as the return path for the AC input signal. 1.3.2.1. Positive "HALF Cycle" Operation When the AC input voltage goes positive, the gate of Q1 is driven high and current flows from the input through the inductors, storing energy in L1 and L2 as shown in Figure 1-8-a). When Q1 turns off, energy in the inductors is released as current flows through D1, through the load and returns through the body diode of Q2 back to the input mains. During the-off time, the current through the inductors (which during this time discharges its energy) flows into the boost diode D1 and close the circuit through the load. 1.3.2.2. Negative "HALF Cycle" Operation During the negative half cycle circuit operation is mirrored. Q2 turns on, current flows through the inductor, storing energy, as shown in Figure 1-8-b). When Q2 turns off, energy is released as current flows through D2, through the load, and back to the mains through the body diode of Q1. a) Positive half cycle 12 b) Negative half cycle Figure 1-8: Bridgeless PFC converter operation Note that the two Power MOSFETs are driven synchronously. It doesn't matter whether the sections are performing as an active boost or as a path for the current to return. In either case there is a benefit of lower power dissipation when current flows through the Power MOSFETs (through intrinsic body diodes) during the return phase. As can be noted, in bridgeless topology a new loss has been introduced in the intrinsic body diodes of MOSFETs and it might add extra power dissipation in the MOSFETs, but since input bridge rectifiers were eliminated, there is some efficiency gain in overall performance of the bridgeless topology. For a conventional PFC, the current sense is easy to monitor by simply inserting a shunt sensing resistor at the return path of the inductor current. However, for a bridgeless PFC, current path does not share the same ground at each half-line cycle. A sensing-power MOSFET and diode current are needed, which makes the bridgeless PFC’s current sensing complicated and difficult to monitor [52, 58]. In addition, since the AC line is floating compared to the PFC stage ground, simple circuitry cannot sense input voltage. Normally a low-frequency transformer or optical coupler is required to perform input voltage sensing. 13 1.3.2.1. PSIM Simulation of Bridgeless Boost Converter PSIM simulation software was used to verify the steady state waveforms of each component. Figure 1-9 shows the PSIM simulation circuits of the bridgeless boost PFC converter. As it can be seen, the power stage section of converter consists of two boost inductors, Ld1 and Ld2, two fast boost diodes, Db1 and Db2, two MOSFETs, Q1 and Q2 and their body diodes. Also it consists of one current loop and one voltage loop. The sensed input voltage is multiplied by the compensated output voltage, and then generates a control signal to be compared with the switching carrier waveforms to generate the gating signals for main MOSFETs. Figure 1-9: PSIM simulation circuit for bridgeless PFC boost converter: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz 14 Figure 1-10 shows the PSIM simulation results of a bridgeless PFC boost converter. The input current is in phase with input voltage, and it has close to unity power factor. Also the output voltage is regulated at around 400V, with a 120 Hz low frequency ripple. The converter is operating at 70 kHz switching frequency, 240 V input voltage and 3.4 kW output power. Figure 1-10: Simulation waveforms for bridgeless PFC boost converter including output voltage, input voltage and input current: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz 1.3.3. Semi-Bridgeless PFC boost Converter The main disadvantage of bridgeless boost converter is the EMI noise issues [54, 59]. For a bridgeless PFC, the output voltage ground is always floating relative to the AC line input. Thus, all parasitic capacitance including MOSFET drain to earth and the output terminals to the earth ground contribute to common mode noise. This large dv/dt at each phase's switching node leads to an increased common mode noise that is difficult to filter. At the same time, the 15 switching node Q2 and D2 are directly connected to input line terminal, which leads to high dv/dt common mode noise. The semi-bridgeless PFC boost converter is proposed [60] to address the EMI noise issues for BL PFC Converters by adding two slow diodes, Da and Db to the input line as shown in Figure 1-11. The added diodes have no effect on the efficiency of converter since they are in parallel with another semiconductor when they conduct. As a matter of fact, they will reduce the current stress on the min MOSFET in BL PFC configuration. Figure 1-11: Bridgeless PFC boost topology 1.3.4. Interleaved PFC Boost Converter The interleaved PFC boost converter (IL PFC) illustrated in Figure 1-12 is simply two boost converters in parallel operating 180° out of phase. The input current is the sum of the two inductor currents IL1 and IL2. Because the inductor’s ripple currents are out of phase, they tend to cancel each other and reduce the input ripple current caused by the boost inductors [61-67]. The maximum input inductor ripple current cancellation occurs at 50% duty cycle. The output capacitor current is the sum of the two boost diode currents less the dc output current. Interleaving reduces the output capacitor ripple current as a function of the duty cycle [64]. As the duty cycle approaches 0%, 50%, and 100% duty cycle, the sum of the two diode currents approaches dc. At these points, the output capacitor only has to filter the inductor ripple current. 16 In order to design the IL PFC converter, it should be treated as two conventional boost PFC converters with half of power rating. Therefore, all equations for the inductor, transistor, and diode in conventional PFC are valid here, since the stresses are unchanged except the ripple current through output capacitors. The input bridge diode has the same power rating as the conventional PFC boost converter. But the capacitor will get the most benefit of interleaving through reduced current ripple. Figure 1-12: Interleaved PFC boost topology As it can be noted, the input rectifier current is exactly the same as input rectifier current in a conventional boost PFC converter, but the inductor current is exactly half, and boost diode and boost MOSFET have less stress, due to the fact that they have to deliver half of the power as in a conventional boost PFC converter. The capacitors will get the most benefits, due to interleaving of each converter. In addition, the interleaved boost converter takes advantage of paralleling semiconductors, and by having them switched out of phase, it doubles the effective switching frequency and introduces smaller input current ripples, so the input EMI filters will be smaller [68, 69]. But it still has the problem of heat management for the input rectifier diode bridges, and suffers from lower efficiency at light load, and low line conditions. 17 1.3.4.1. PSIM Simulation of Interleaved Boost Converter PSIM simulation software was used to verify steady state waveforms of each component. Figure 1-13 shows the PSIM simulation circuits of the bridgeless boost PFC converter. Figure 1-13: PSIM simulation circuit for interleaved PFC boost converter: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz As it can be seen, the power stage section of the converter consists of two boost inductors, Ld1 and Ld2, two fast boost diodes, Db1 and Db2, two MOSFETs, MOS1 and MOS2 and their 18 body diodes. Also it consists of two current loops and one voltage loop. The sensed input voltage is multiplied by the compensated output voltage, and then generates a control signal to be compared with the switching carrier waveforms to generate the gating signals for main MOSFETs. Figure 1-14 shows the PSIM simulation results of an interleaved PFC boost converter. The input current is in phase with input voltage, and it has close to unity power factor. Also the output voltage is regulated at around 400V, with a 120 Hz low frequency ripple. The converter is operating at 70 kHz switching frequency, 240 V input voltage and 3.4 kW output power. Figure 1-14: Simulation waveforms for interleaved PFC boost converter including output voltage, input voltage and input current: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz 1.3.5. Ripple Steering Technique in PFC Applications Coupled magnetics filter techniques, known also as ripple steering techniques [70], have existed for many years, and have been applied to different topologies in industries [71, 72]. As shown in Figure 1-15, this technique replaces a series smoothing inductor with a pair of coupled 19 inductors and a blocking capacitor. Recent attention is given to the application of ripple steering to PFC boost converters [73, 74]. Figure 1-16 shows the modified PFC boost converter with a coupled inductor. Ripple-steering technique has several advantages in a PFC boost converter. Since it eliminates most of the differential-mode conducted noise, it enables the reduction in EMI filter size and complexity, especially in its differential filtering section (Cx capacitors and differential mode inductors). Reducing Cx capacitors to a minimum has an additional benefit for applications with tight specifications on standby consumption. Cx capacitors cause a considerable reactive current to flow through the filter, which is a source of additional and unwanted loss. Furthermore, the discharge resistor that must be placed in parallel to Cx for safety can be higher. As a result, both losses will be minimized. Although the control strategy of the PFC stage is similar to that of a conventional boost converter, but their power stage transfer functions are different. No modeling has been done to verify the effect of added coupled filter to the power stage transfer functions, and thus the design of the feedback loop compensator. Figure 1-15: Smoothing transformer in a coupled filter Figure 1-16: Modified PFC boost converter with coupled inductors 20 Replacing the inductor in a conventional boost converter shown in Figure 1-3 with the coupled inductor and a blocking capacitor shown in Figure 1-15 will result to the modified boost converter with coupled magnetic shown in Figure 1-16. 1.3.6. Average Switch Model Models for the PWM-switch were first introduced in [75, 76]. It was then adopted for different converters in [77-81]. For the average switch model of a converter, the active and passive switches shown in Figure 1-17 are replaced with the PWM-switch equivalent circuits shown in Figure 1-18 and Figure 1-19. Figure 1-17: One active switch and one passive switch Figure 1-18: PWM-switch Figure 1-19: Averaged model of PWM-switch 21 Using this model and applying that to modified boost converter with coupled inductors, any of the transfer functions between the output variables (output voltage and inductor current) and the input variables (input voltage and duty ration) can be derived. 1.4. Motivation and Objectives of the Thesis The objective of this thesis is to investigate and conduct research on the highperformance single-phase solutions for AC-DC power factor corrected converters for plug-in hybrid electric vehicle battery chargers. A variety of circuit topologies and control methods have been developed for PFC applications. The two-stage approach is the proper candidate for PHEV battery chargers, where the power rating is relatively high, and lithium-ion batteries are used as the main energy storage system. The improvement of overall charger efficiency is critical for the emergence and acceptance of these vehicular technologies, as the charger efficiency increases, the charge time and utility cost decreases. Additionally, to meet the efficiency and power factor requirements and regulatory standards for the AC supply mains, power factor correction is essential. Due to limited space in vehicle and increasing power consumption, chargers are required to deliver more power with smaller volume. As a key component of a charger system, the frontend AC/DC converter must achieve high efficiency and power density. In this dissertation, several conventional plug in hybrid electric vehicle charger front end AC-DC converter topologies are investigated and a new bridgeless interleaved and a phase shifted semi-bridgeless power factor corrected converter are proposed to improve the efficiency and performance, which is critical to minimize the charger size, charging time and the amount and cost of electricity drawn from the utility. 22 1.5. Outline of the Thesis This thesis consists of five chapters. Chapter 1 introduces the subject of AC-DC power factor corrected boost converters with particular focus on its application for plug-in hybrid electric vehicles. This chapter establishes motivation and sets objectives for the research contributions presented in Chapters 2-5. A literature review on existing topologies - conventional boost converter, bridgeless boost converter and interleaved boost converter - modes of operation, control techniques and PSIM simulation of existing topologies are presented in Chapter 1. Chapter 2 proposes a new bridgeless interleaved PFC boost converter for PHEV battery charger applications. The circuit description and steady state analysis, semiconductor loss analysis, design procedure, logic implementation, PSIM simulation and experimental results are presented. Chapter 3 introduces a new phase shifted semi-bridgeless PFC boost converter for PHEV battery charger applications. The circuit description and steady state analysis, semiconductor loss analysis, design procedure, logic implementation, PSIM simulation and experimental results are presented. A new average switch modeling for ripple steering technique applied to PFC boost converters and compensation network design based on derived voltage and current power stage transfer functions are presented in Chapter 4. Simulation and experimental results are presented to verify the model. Chapter 5 summarizes the contributions of the research presented in this thesis and gives recommendations for future work. 23 CHAPTER 2. Bridgeless Interleaved PFC Boost Converter 2.1. Introduction In this chapter, a new bridgeless interleaved power factor corrected converter for plug in hybrid electric vehicle charger front end AC-DC converter topology is investigated and proposed to improve the efficiency and performance, which is critical to minimize the charger size, charging time, and the amount and the cost of electricity drawn from the utility. A detailed analytical model for this topology is developed, enabling the calculation of power losses and efficiency. Experimental and simulation results of a prototype boost converter converting universal AC input voltage to 400 V DC at 3.4 kW are given to verify the proof of concept, and analytical work reported in this chapter. The results show a power factor of greater than 0.99 from 750 W to 3.4 kW, THD less than 5% from half load to full load, and a peak efficiency of 98.94 % at 265 V input, and 1.2 kW output power. 2.2. Circuit Description and Steady State Analysis The bridgeless interleaved (BLIL) PFC converter [82] shown in Figure 2-1 is proposed to address the problems discussed in Chapter 1 for the conventional boost, bridgeless boost, and interleaved boost topologies. This converter introduces two more MOSFETs and two more fast diodes in place of four slow diodes used in the input bridge of the benchmark interleaved boost PFC converter. To analyze the circuit operation, the input line cycle has been separated into the positive and negative half cycles, as explained in sub-sections that follow. In addition, the detailed circuit operation depends on the duty cycle, therefore positive half cycle operation analysis is provided for D > 0.5 and D < 0.5. 24 Figure 2-1: Proposed bridgeless interleaved (BLIL) PFC boost converter 2.2.1. Positive Half Cycle Operation Referring to Figure 2-1, during the positive half cycle, when the AC input voltage is positive, Q1/Q2 turn on and current flows through L1 and Q1 and continues through Q2 and then L2, returning to the line while storing energy in L1 and L2. When Q1/Q2 turn off, energy stored in L1 and L2 is released as current flows through D1, through the load and returns through the body diode of Q2 back to the input mains. With interleaving, the same mode happens for Q3/Q4, but with a 180 degree phase delay. The operation for this mode is Q3/Q4 on storing energy in L3/L4 through the path L3-Q3-Q4-L4 back to the input. When Q3/Q4 turn off, energy is released through D3 to the load and returning through the body diode of Q4 back to the input mains. 2.2.2. Negative Half Cycle Operation Referring to Figure 2-1, during the negative half cycle, when the AC input voltage is negative, Q1/Q2 turn on and current flows through L2 and Q2 and continues through Q1 and then L1, returning to the line while storing energy in L2 and L1. When Q1/Q2 turn off, energy stored in L2 and L1 is released as current flows through D2, through the load and returns through the body diode of Q1 back to the input mains. 25 With interleaving, the same mode happens for Q3/Q4, but with a 180 degree phase delay. The operation for this mode is Q3/Q4 on storing energy in L3/L4 through the path L4-Q4-Q3-L3 back to the input. 2.2.3. Detailed Positive Half Cycle Operation and Analysis for D > 0.5 The detailed operation of the proposed BLIL PFC converter depends on the duty cycle. During any half cycle, the converter duty cycle is either greater than 0.5 (when the input voltage is smaller than half of output voltage) or smaller than 0.5 (when the input voltage is greater than half of output voltage). Figure 2-2: Interval 1: Q1 and Q2 are “ON”, and body diode of Q4 conducting Figure 2-3: Interval 2 and 4: Q1, Q2, Q3 and Q4 are “ON” 26 Figure 2-4: Interval 3: Q3 and Q4 are “ON”, and body diode of Q2 conducting Figures 2-2 to 2-4 show the three unique operating interval circuits of the proposed converter for duty cycles greater than 0.5 during positive half cycle operation. Waveforms of the proposed converter during these conditions are shown in Figure 2-5. Since the switching frequency of proposed converter is much higher than the frequency of input line voltage, the input voltage v2 is considered constant during one switching period T4 . The input voltage is given by: 56 √2 9: $;< % 2-1 In a positive half cycle of the input voltage, the duty ratio of the proposed converter determines the following voltage relation: = >? & &,0 2-2 The intervals of operation are explained as follows. In addition, the ripple current components are derived, enabling calculation of the input ripple current, which provides design guidance to meet the required input current ripple standard. Interval 1 [t0-t1]: At t0, Q1/ Q2 are ON, and Q3/Q4 are off, as shown in Figure 2-2. During this interval, the current in series inductances L1 and L2 increases linearly and stores the energy in these inductors. The ripple currents in Q1 and Q2 are the same as the current in series 27 inductances L1 and L2, where the ripple current is given by: Figure 2-5: BLIL PFC boost converter steady-state waveforms at D > 0.5 28 ∆;A& & A -A+ 56 1 − ) ': 2-3 The current in series inductances L3 and L4 decreases linearly and transfers the energy to the load through D3, Co and body diode of Q4. The ripple current in series inductances L3 and L4 is given by: ∆;AD & AE -AF 9 − 56 1 − ) ': 2-4 The input ripple current is the sum of currents in L1/L2 and L3/L4: ∆G6 & A -A+ 9 1 − ) ': 2-5 Interval 2 [t1-t2]: At t1, Q3/Q4 are turned on, while Q1/Q2 remain on, as shown in Figure 2-3. During this interval, the current in the four inductors each increase linearly, storing energy in these inductors. The ripple currents in Q1 and Q2 are the same as the ripple current in series inductances L1 and L2 as given by: ∆;A& & A -A+ 56 ) − H ': & 2-6 Similarly, the ripple currents in Q3 and Q4 are the same as the ripple current in series inductances L3 and L4: ∆;AD & AE -AF 56 ) − H ': & 2-7 The input ripple current is the sum of currents in L1/L2 and L3/L4: ∆G6 H A -A+ 56 ) − H ': & 2-8 Interval 3 [t2-t3]: At t2, Q1/Q2 are turned off, while Q3/ Q4 remain on, as shown in Figure 2-4. During this interval, the current in series inductances L3 and L4 increases linearly and stores the energy in these inductors. The ripple currents in Q3 and Q4 are the same as the ripple current in series inductances L3 and L4: 29 ∆;AD & AE -AF 56 1 − ) ': 2-9 The current in L1 and L2 decreases linearly and transfers the energy to the load through D1, Co and body diode of Q2. The ripple current in series inductances L1 and L2 is given by: ∆;A& & A -A+ 9 − 56 1 − ) ': 2-10 The input ripple current is the sum of currents in L1/L2 and L3/L4: ∆G6 & A -A+ 9 1 − ) ': 2-11 Interval 4 [t3-t4]: At t3, Q3/Q4 remain on, while Q1/Q2 are turned on, as shown in Figure 2-3. During this interval, the currents in the four inductors each increase linearly, storing energy in these inductors. The ripple currents in Q1 and Q2 are the same as the ripple currents in L1/L2: ∆;A& & A -A+ 56 ) − H ': & 2-12 Similarly, the ripple currents Q3 and Q4 are the same as the ripple current in series inductances L3 and L4: ∆;AD & AE -AF 56 ) − H ': & 2-13 The input ripple current is the sum of currents in L1/L2 and L3/L4: ∆G6 H A -A+ 56 ) − H ': & 2-14 2.2.4. Detailed Positive Half Cycle Operation and Analysis for D < 0.5 Figure 2-6 to Figure 2-8 show the operating interval circuits of the proposed converter for duty cycles smaller than 0.5 during the positive half cycle. The waveforms of the proposed converter during these conditions are shown in Figure 2-9. The intervals of operation are explained as follows: 30 L1 D1 D2 D3 D4 L3 Vin L2 Co L4 Q2 Q1 Vg1 Q3 L O A D Q4 Vg2 Figure 2-6: Interval 1 and 3: Body diodes of Q2 and Q4 conducting Figure 2-7: Interval 2: Q1 and Q2 are “ON”, and body diode of Q4 conducting Figure 2-8: Interval 4: Q3 and Q4 are “ON”, and body diode of Q2 conducting 31 Interval 1 [t0-t1]: At t0, Q1 and Q2 turn off, while Q3 and Q4 remain off, as shown in Figure 2-6. During this interval, the current in series inductances L1 and L2 decreases linearly and transfers the energy to the load through D1, Co and body diode of Q2. The ripple current in series inductances L1 and L2 is: ∆;A& 9 − 56 & A -A+ & H − ) ': 2-15 In addition, the current in the series inductances L3 and L4 also decreases linearly, transferring the energy to the load through D3, Co and body diode of Q4. The ripple current in series inductances L3 and L4 is: ∆;AD 9 − 56 & AE -AF & H − ) ': 2-16 The input current is the sum of currents in L1/L2 and L3/L4: ∆G6 9 − 56 H A -A+ & H − ) ': 2-17 Interval 2 [t1-t2]: At t1, Q1/Q2 turn on, while Q3/Q4 remain off, as shown in Figure 2-7. During this interval, the current in series inductances L1 and L2 increases linearly, storing energy in these inductors. The ripple currents in Q1 and Q2 are the same as the current in series inductances L1 and L2, where the ripple current is given by: ∆;A& & A -A+ 56 )': 2-18 The current in series inductances L3 and L4 decreases linearly and transfers the energy to the load through D3, Co and body diode of Q4. The ripple current in L3 and L4 is: ∆;AD & AE -AF 9 − 56 )': 2-19 The input ripple current is the sum of the currents in L1/L2 and L3/L4: ∆G6 & A -A+ 9 )': 2-20 32 Vg1 Gating Signals t Vg2 t ∆Iin Input Current t IL1 IL3 Inductor Current t IQ3 IQ1 FET Current t ID3 Boost Diode Current ID1 t IQd3 Body Diode Current t IQd1 t0 t1 t2 (1-D)TS t3 t4 t5 t DTS Figure 2-9: BLIL PFC boost converter steady-state waveforms at D < 0.5 33 Interval 3 [t2-t3]: At t2, Q1/Q2 are turned off, while Q3/Q4 remain off, as shown in Figure 2-6. During this interval, the current in series inductances L1 and L2 decreases linearly and transfers the energy to the load through D1, Co and body diode of Q2. The ripple current in series inductances L1 and L2 is given by: ∆;A& & A -A+ 9 − 56 & H − ) ': 2-21 Similarly, the current in the series inductances L3 and L4 also decreases linearly, transferring the energy to the load through D3, Co and body diode of Q4. The ripple current in series inductances L3 and L4 is: ∆;AD & AE -AF 9 − 56 & − ) ': 2-22 & − ) ': 2-23 H The input current is the sum of currents in L1/L2 and L3/L4: ∆G6 H A -A+ 9 − 56 H Interval 4 [t3-t4]: At t3, Q3/Q4 are turned on, while Q1/Q2 remain off, as shown in Figure 2-8. During this interval, the current in series inductances L3 and L4 increases linearly and stores the energy in these inductors. The ripple currents in Q3 and Q4 are the same as the current in series inductances L3 and L4, where the ripple current is given by: ∆;AD 56 )': & AE -AF 2-24 The current in series inductances L1 and L2 decreases linearly and transfers the energy to the load through D2, Co and body diode of Q4. The ripple current in L1 and L2 is: ∆;A& & A -A+ 9 − 56 )': 2-25 The input ripple current is the sum of currents in L1/L2 and L3/L4: ∆G6 & A -A+ 9 )': 2-26 34 The operation of converter during the negative input voltage half cycle is similar to the operation of converter during the positive input voltage half cycle. 2.3. Loss Modeling In order to properly select the power stage components of a converter and calculate the associated power losses, it is necessary to determine the RMS and the average values of their currents [83]. In a typical boost converter, the MOSFET and diode current waveforms are pulsed-width modulated, with both the duty cycle and peak amplitude varying with the AC input. Without an effective mathematical method for computing these RMS and average values, the proper design and selection of power stage components can be flawed. The following assumptions were made in order to analyze the converters and to derive the stress equations: a) These calculations are based on CCM operation of the PFC boost converter. b) Assuming unity power factor, the line current is in phase and shape with the input line voltage – a sinusoidal waveform. c) The PFC output voltage is DC with no voltage ripple. In a typical boost converter, the converter MOSFET duty cycle is given by: IJ % 1− |?L | = 1− MN |:6 | = 2-27 Therefore IJ,O: * & QT R1 − MN P P |:6 | H = S U% 2-28 Assuming the inductor current is a sinusoidal waveform: ;A % GV |$;< % | 2-29 The instantaneous MOSFET current and its RMS current can be derived respectively: ;J % GV |$;< % |. IJ,O: 2-30 35 * & QT RGV |$;< % | 1 − MN GJ,O: P P H S U% |:6 | = 2-31 The inductor current ripple is assumed to be half of peak inductor current: ∆G & MN H H 2-32 The high frequency ripple components of inductor current is assumed to be a triangular waveform with a fixed duty cycle, so the RMS current in each inductor is defined by: GA,O: * & MN H √H H + & H√D ∆G H Y ?L Z√D MN 2-33 The boost diode duty cycle is given by: I0 % MN |:6 | 1 − IJ % = 2-34 Therefore the instantaneous boost diode current and its RMS current can be derived respectively: GV |$;< % | ;0 % G0,O: MN |:6 | = * QT RGV |$;< % | P & P 2-35 MN |:6 | = H S U% 2-36 The output capacitor current has high frequency and low frequency components. The low frequency component is simply calculated by: G[,O: A\ = √H √H = H = 2-37 The high frequency RMS ripple current component is: G[,O: /\ ?L = + *]P = − =+ &] MN ?L 2-38 The same method was used to derive RMS current in different topologies. Table 2-1 shows a summary of component RMS current stress for conventional boost converter and bridgeless boost converter. Table 2-2 provides the same summary for interleaved boost converter and bridgeless interleaved boost converter. 36 Table 2-1: Summary of component RMS current for conventional boost and bridgeless boost topologies Topology Input Bridge Diode Boost Fast Diode Boost Transistor Intrinsic Bridgeless PFC 6 9V Not Applicable 97 6 ^ 48 9V Boost Inductor Boost Transistor Conventional PFC 6 3 ^ 6 2 9 H 3e 39V + 49H − 649V 9 ^ e √69V 9 97 6 ^ 48 9V 6 H 3e 39V + 49H − 649V 9 ^ e √69V 9 √3 6 2 9 Not Applicable Diode √2 2 9 Output Capacitor Ripple (LF) √2 2 9 √2 H *36 − 2H 29 Output Capacitor Ripple (HF) √3 6 2 9 √2 H *36 − 2H 29 Table 2-2: Summary of component RMS current for interleaved and bridgeless interleaved boost topologies Topology Boost Inductor Boost Fast Diode Boost Transistor Intrinsic Diode Output Capacitor Ripple (LF) Output Capacitor Ripple (HF) Bridgeless Interleaved PFC 5 6 4√3 9V 3 6 ^ 2 2. 9 √3 6 4 9 6 9V Input Bridge Diode Boost Transistor Interleaved PFC 5 6 4√3 9V 6 H 3e 39V + 49H − 649V 9 ^ e 2√69V 9 Not Applicable √2 2 9 6 16 9 H ^ − H 9 6e 9V 6 Not Applicable 6 H 3e 39V + 49H − 649V 9 ^ e 2√69V 9 √3 6 4 9 √2 2 9 6 16 9 H ^ − H 9 6e 9V 6 37 Table 2-3 shows a summary of component Average current stress for conventional boost converter and bridgeless boost converter. Table 2-4 provides the same summary for interleaved boost converter and bridgeless interleaved boost converter. Table 2-3: Summary of component average current for conventional boost and bridgeless boost topologies Topology Conventional PFC 4 6 e 9V Boost Inductor 2 6 e 9V Input Bridge Diode Diode Not Applicable 1 6 2 9 Not Applicable 1 6 2 9 6 Boost Transistor Intrinsic 0 6 9 Boost Fast Diode Boost Transistor Bridgeless PFC 4 1 1 − e 9V 9 1 6 2 9 Table 2-4: Summary of component average current for interleaved and bridgeless interleaved boost topologies Topology Interleaved PFC 2 6 e 9V Boost Inductor 2 6 e 9V Input Bridge Diode Boost Fast Diode Boost Transistor Boost Transistor Intrinsic Diode 6 Bridgeless Interleaved PFC 0 Not Applicable 6 2. 9 1 6 4 9 Not Applicable 1 6 4 9 4 1 11 − e 9V 2 9 1 6 4 9 38 As can be noted, in both the bridgeless topology and bridgeless interleaved topology, a new loss has been introduced in the intrinsic body diodes of the MOSFETs, but since input bridge rectifiers were eliminated, there is some efficiency gain in overall performance of these topologies. The intrinsic body diode of MOSFETs conduct when the boost transistors are off and its value is the same as the current in the boost diodes, when they conduct and transfer energy to the output capacitors. Also the low frequency RMS ripple current through output capacitors is constant and interleaving has no effect on it. But the high frequency ripple current will be reduced significantly, as it is shown in Figure 2-10. Also it is noted that as the input voltage increases, the high frequency ripple reduces. Figure 2-10: RMS ripple current through output capacitors vs input voltage at Po = 3400 W, fsw = 70 kHz Figure 2-11 shows the loss distribution of the semiconductors in the four topologies investigated in this thesis at Vin = 240 V, Po = 3400 W, Vo = 400 V and fsw = 70 kHz. The regular diode losses are only conduction losses in bridge rectifier diodes. Because of low reverse 39 recovery characteristics of SiC diodes, it is the selected component for boost diodes; therefore it is assumed that there is only conduction loss in fast diodes. Switching losses, conduction losses, gate charge losses and ½ CV2 losses are all included in MOSFET losses. It should be noted that the inductor loss analysis is not included in this study. 39.6 W Bridgeless Boost 50 7.8 W 0W 0W 7.8 W 16.6 W 8.3 W 0W 0W 10 8.3 W 12.9 W 12.7 W 12.9 W 11.3 W 20 19.1 W Bridgeless Interleaved Boost 27.6 W 30 27.6 W Power Losses (W) Interleaved Boost 40 48.7 W Conventional Boost 35.7 W 48.7 W 60 Total Losses Devices / Total Losses Intrinsic Body Diodes FETs Fast Diodes Regular Diodes 0 Figure 2-11: Comparison of the estimated loss distribution in the semiconductors at 70kHz switching frequency, 240V input, 3400W load at 400V The regular diodes in the input bridge rectifiers have the largest share of losses among the topologies with the input bridge rectifier. The bridgeless topologies eliminate this large loss component (~27.6 W). However, the tradeoff is that the MOSFET losses are higher and the intrinsic body diodes of the MOSFETs conduct, producing new losses (~7.85 W). The fast diodes in the bridgeless interleaved PFC have slightly lower power losses, since the boost diode average current is lower in these topologies. Overall, the MOSFETs are under more stress in bridgeless topologies, but the total semiconductor losses for the proposed bridgeless interleaved 40 boost are 37% lower than the benchmark conventional boost, 10% lower than the bridgeless boost and 37% lower than the benchmark interleaved boost. Since the bridge rectifier losses are so large, it was expected that the bridgeless interleaved boost converter would have the least power losses among all four introduced topologies. Also it was noted that the losses in the input bridge rectifiers were 56% of total losses in the conventional PFC converter and in the benchmark interleaved PFC converter. Therefore eliminating the input bridges in PFC converters is justified despite the fact that new losses are introduced. 2.4. Math Modeling Verification In order to verify the accuracy of converters modeled in section 2.3, the math modeling results were compared with PSIM simulations for two different topologies. The RMS inductor current, the RMS MOSFET current, the average boost diode current and the average MOSFET body diode current for an interleaved boost PFC topology and a bridgeless boost PFC topology were examined. Figure 2-12 to Figure 2-14 show the PSIM simulation and Math modeled values of boost inductor RMS current, MOSFET RMS current and boost diode average current for an interleaved boost PFC converter at 240 V input and 120 V input voltages. Figure 2-15 to Figure 2-18 show the PSIM simulation and Math modeled values of boost inductor RMS current, MOSFET RMS current, boost diode average current and MOSFET body diode average current for a bridgeless boost PFC converter at 240 V input and 120 V input voltages. 41 9 8 Inductor RMS Current (A) 7 6 5 4 3 2 Math Modelling - Vin = 240 V PSIM Simulation - Vin = 240 V Math Modelling - Vin = 120 V PSIM Simulation - Vin = 120 V 1 4000 3500 3000 Output Power (W) 2500 2000 1500 1000 500 0 0 Figure 2-12: Simulation and math modeling of inductor RMS current for an interleaved PFC converter 7 5 4 3 2 4000 3500 2000 1500 1000 500 0 0 3000 Math Modelling - Vin = 240 V PSIM Simulation - Vin = 240 V Math Modelling - Vin = 120 V PSIM Simulation - Vin = 120 V 1 2500 MOSFET RMS Current (A) 6 Output Power (W) Figure 2-13: Simulation and math modeling of MOSFET RMS current for an interleaved PFC converter 42 5 4 3.5 3 2.5 2 1.5 4000 2000 1500 1000 500 0 0 3500 0.5 3000 Math Modelling - Vin = 240 V PSIM Simulation - Vin = 240 V Math Modelling - Vin = 120 V PSIM Simulation - Vin = 120 V 1 2500 Boost Diode Average Current (A) 4.5 Output Power (W) Figure 2-14: Simulation and math modeling of boost diode average current for an interleaved PFC converter 18 16 Inductor RMS Current (A) 14 12 10 8 6 4 Math Modelling - Vin = 240 V PSIM Simulation - Vin = 240 V Math Modelling - Vin = 120 V PSIM Simulation - Vin = 120 V 2 4000 3500 3000 2500 2000 1500 1000 500 0 0 Output Power (W) Figure 2-15: Simulation and math modeling of inductor RMS current for a bridgeless PFC converter 43 10 9 7 6 5 4 3 4000 2000 1500 1000 500 0 0 3500 1 3000 Math Modelling - Vin = 240 V PSIM Simulation - Vin = 240 V Math Modelling - Vin = 120 V PSIM Simulation - Vin = 120 V 2 2500 MOSFET RMS Current (A) 8 Output Power (W) Figure 2-16: Simulation and math modeling of MOSFET RMS current for a bridgeless PFC converter 5 4 3.5 3 2.5 2 1.5 4000 2000 1500 1000 500 0 0 3500 0.5 3000 Math Modelling - Vin = 240 V PSIM Simulation - Vin = 240 V Math Modelling - Vin = 120 V PSIM Simulation - Vin = 120 V 1 2500 Boost Diode Average Current (A) 4.5 Output Power (W) Figure 2-17: Simulation and math modeling of boost diode average current for a bridgeless PFC converter 44 9 7 6 5 4 3 2 4000 2000 1500 1000 500 0 0 3500 1 3000 Math Modelling - Vin = 240 V PSIM Simulation - Vin = 240 V Math Modelling - Vin = 120 V PSIM Simulation - Vin = 120 V 2500 MOSFET Body Diode Average Current (A) 8 Output Power (W) Figure 2-18: Simulation and math modeling of MOSFET body diode average current for a bridgeless PFC 2.5. Simulation Results PSIM simulation software was used to verify steady state waveforms of each component. Figure 2-19 shows the PSIM simulation circuits of the proposed BLIL PFC converter. As it can be seen, the power stage section of converter consists of four boost inductors, Ld1 to Ld4, four fast boost diodes, Db1 to Db4, four switches, Q1 to Q4 and their body diodes Dq1 to Dq4. Also it consists of two current loops and one voltage loop. The sensed input voltage is multiplied by the compensated output voltage, and then generates a control signal to be compared with the switching carrier waveforms to generate the gating signals for the main MOSFETs. 45 Figure 2-19: PSIM simulation circuit for the proposed BLIL PFC boost converter: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz Figure 2-20 shows the PSIM simulation results of a BLIL PFC boost converter. The input current is in phase with input voltage, and it has close to unity power factor. Also the output voltage is regulated at around 400V, with a 120 Hz low frequency ripple. The converter is operating at 70 kHz switching frequency, 240 V input voltage and 3.4 kW output power. 46 Figure 2-20: Simulation waveforms for the proposed BLIL PFC boost converter including output voltage, input voltage and input current: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz 2.6. Logic Implementation A standard two-phase interleaved CCM PFC controller from Texas Instrument, UCC28070 was used to implement the logic circuitry of the prototype unit. The key features of this controller are as following: • Interleaved average current mode PWM control • Advanced current synthesizer for superior efficiency, accurate current sensing, and high power factor • Highly linear multiplier output with internal voltage feed-forward correction for near unity power factor • Programmable switching frequency (30 kHz to 300 kHz) • Selectable frequency dithering for reduced EMI 47 • Phase management for high-efficiency light-load operation A simplified logic implementation of this controller applied to bridgeless interleaved PFC boost converter is given in Figure 2-21. Figure 2-21: Logic implementation of BLIL PFC boost converter using UCC28070 controller 48 2.7. Experimental Results An experimental prototype, illustrated in Figure 2-22, was built to verify the operation of the proposed converter. Figure 2-23 shows the input voltage, input current and PFC bus voltage of the converter under the following test conditions: Vin = 240 V, Iin = 15 A, Po = 3400 W, Vo = 400 V, fsw = 70 kHz. The input current is in line and phase with the input voltage, and its shape is close to a sinusoidal waveform. Table 2-3 shows the semiconductors and power components used in the 3.4 kW CCM experimental prototype, a bridgeless interleaved PFC converter. Figure 2-22: Breadboard prototype of BLIL PFC boost converter 49 Output Voltage Ch1= Vo 100V/div. Input Voltage Ch2 = Vin 100V/div. Input Current Ch3 = Iin 10A/div. Figure 2-23: Breadboard proposed BLIL PFC experimental waveforms; Test condition: Po = 3400W, Vin = 240V, Iin = 15A. MOSFET Current Ch4 = IQ1 2A/div. Diode Current Ch3 = ID1 2A/div. Gating Signal Ch1 = Vg 10V/div. Figure 2-24: Gating signal, boost MOSFET and diode current for D > 0.5 50 Diode Current Ch3 = ID1 5A/div. MOSFET Current Ch4 = IQ1 5A/div. Gating Signal Ch1= Vg 10V/div. Figure 2-25: Gating signal, boost MOSFET and diode current for D < 0.5 Output Voltage Ch1= Vo 100V/div. Input Current Ch4 = Iin 10A/div. Input Voltage Ch2 = Vin 100V/div. Output Current Ch3 = Iin 5A/div. Figure 2-26: Load transient response from FL to NL (3400W to 0W) 51 Output Voltage Ch1= Vo 100V/div. Input Voltage Ch2 = Vin 100V/div. Input Current Ch4 = Iin 10A/div. Output Current Ch3 = Iin 5A/div. Figure 2-27: Load transient response from NL to FL (0W to 3400 W) Table 2-5: Component / devices used in prototype unit PFC Bridgeless Interleaved Benchmark Interleaved PFC Topology Device Part # / Value # of devices Regular Diode 25ETS08S 4 Fast Diode IDB06S60C 2 MOSFET IPB60R099CP 2 Inductor Kool mu 77071 core 60 turns / 400 µH 2 Capacitor EKXJ451ELL820 / 82 µF 10 Fast Diode CSD10060 4 MOSFET IPP60R099CP 4 Inductor Capacitor Kool mu 77071 core 60 turns / 400 µH EKXJ451ELL820 / 82 µF 4 10 52 In order to verify the quality of the input current, its harmonics, up to 39th harmonic, are given and compared with the IEC 61000-3-2 standard. 2.5 Harmonic Current (A) 2 EN 61000-3-2 Class D Limits (A) Amplitude (A) Vin = 120 V 1.5 Amplitude (A) Vin = 240 V 1 0.5 39 37 35 33 31 29 27 25 23 21 19 17 15 13 11 9 7 5 3 0 Harmonics Order Figure 2-28: Input current harmonics at full load for Vin = 120 V at Po = 1700 W and 240 V at Po = 3400 W 45 40 35 Vin=240 30 THD (%) Vin=120 25 20 15 10 5 3500 3000 2500 2000 1500 1000 500 0 0 Output Power (W) Figure 2-29: RMS Total harmonics distortion vs. output power at Vin = 120V and Vin = 240V 53 1.02 1 0.98 Power factor 0.96 0.94 0.92 Vin=240 0.9 Vin=120 0.88 0.86 3500 3000 2500 2000 1500 1000 500 0 0.84 Output Power (W) Figure 2-30: Power factor vs. output power at Vin = 120V and Vin = 240V 100 99 97 96 Vin = 90 V 95 Vin = 120 V Vin = 220 V 94 Vin = 240 V Vin = 265 V 5000 4500 4000 Output Power (W) 3500 3000 2500 2000 1500 1000 500 93 0 Efficiency (%) 98 Figure 2-31: Efficiency vs. output power for the proposed bridgeless interleaved PFC boost converter 54 100 99 Efficiency (%) 98 97 96 Bridgeless Interleaved PFC Converter Vin = 240 V 95 Benchmark Interleaved PFC Converter Vin = 240 V 4500 4000 3500 3000 2500 2000 1500 1000 500 0 94 Output Power (W) Figure 2-32: Efficiency vs. output power for different topologies at Vin = 240V Figure 2-24 shows the gating signal, boost MOSFET current, and boost diode current for duty cycles greater than 50%. Figure 2-25 shows the same voltage and current for duty cycles smaller than 50%. Figure 2-26 and Figure 2-27 show the output voltage transient response to a change to the load form FL to NL and vice versa. As can be noted, the output voltage regulates right away. Figure 2-28 shows the input current harmonics versus harmonic numbers at full load for 120 V and 240 V input voltages. It is clearly shown that the generated harmonics are well below IEC 61000-3-2 standard for the input line harmonics which is required for PHEV chargers. In Figure 2-29, the input current total harmonics distortions are given at full load and for 120 V and 240 V input voltages. It can be noted that mains current THD are smaller than 5% from 50% load to full load and it is compliant to IEC 61000-3-2. Another parameter to show the quality of the input current is the power factor. In Figure 2-30, the converter power factor is shown over the entire load range for different input voltages. As it can be seen, the power factor is greater than 0.99 from 50% load to full load. 55 The efficiency of converter versus output power for different input voltages is provided in Figure 2-31. With the proposed bridgeless interleaved PFC converter a peak efficiency of 98.94% was reached at 265 V input and 1.2 kW output power. High efficiency over the entire load range is achieved in this topology, enabling fewer problems with heat dissipation and cooling systems. Furthermore, a higher efficiency means more power is available from the mains feed to charge the batteries, reducing charging time and electricity costs. Figure 2-32 illustrates the measured experimental efficiency of benchmark interleaved boost PFC and BLIL boost PFC topology discussed in this chapter for 240 V input voltage. The proposed BLIL PFC converter has the highest efficiency over the entire load range. 70 60 Loss Reduction (%) 50 40 30 20 10 3500 3000 2500 2000 1500 1000 500 0 0 Output Power (W) Figure 2-33: Loss reduction as a function of output power at Vin = 240V, Vo = 400 V and 70 kHz for the proposed BLIL converter compared to the benchmark interleaved boost converter 56 Loss reduction curves as a function of output power for BLIL PFC against benchmark IL PFC are provided in Figure 2-33 and Figure 2-34 for 240 V and 120 V. 60 Loss Reduction (%) 50 40 30 20 10 1800 1600 1400 1200 1000 800 600 400 200 0 0 Output Power (W) Figure 2-34: Loss reduction as a function of output power at Vin = 120V, Vo = 400 V and 70 kHz for the proposed BLIL converter compared to the benchmark interleaved boost converter 2.8. Conclusion A high performance AC-DC boost converter topology has been presented in this chapter for the front-end AC-DC converter in PHEV battery chargers. The proposed converter topology has been analyzed and performance characteristics presented. Also an analytical model for the four different topologies was developed, enabling the calculation of power losses and efficiencies. A prototype converter was built to verify this proof-of-concept. The theoretical waveforms were compared with the results taken from prototype unit. Also some key experimental waveforms 57 were given. Finally, input current harmonics at each harmonic order were compared more explicitly with the IEC 61000-3-2 standard limits. It can be noted that mains current THD are smaller than 5% from 50% load to full load and the converter is compliant with the IEC 6100-32 standard. The converter power factor was also provided for full power range and different input voltages. It can be seen that the power factor is greater than 0.99 from 50% load to full load. The proposed converter achieves a peak efficiency of 98.94 % at 265 V input and 1.2 kW output power. The converter topology shows a high input power factor, high efficiency over the entire load range, and excellent input current harmonics. It is a potential option for single phase PFC in higher power battery charging applications. 58 CHAPTER 3. Phase-Shifted Semi Bridgeless PFC Converter 3.1. Introduction In this chapter, a phase shifted semi-bridgeless boost power factor corrected converter is proposed for plug in hybrid electric vehicle battery chargers. The converter features high efficiency at light loads and low lines, which is critical to minimize the charger size, charging time, and the amount and cost of electricity drawn from the utility; the component count, which reduces the charger cost; and reduced EMI. The converter is ideally suited for automotive level 1 residential charging applications. A detailed converter description and steady state operation analysis of this converter is presented. Experimental results of a prototype boost converter, converting universal AC input voltage to 400 V DC at 3.4 kW are given and the results are compared to a benchmark interleaved boost converter to verify the proof of concept, and analytical work reported. The results show a power factor greater than 0.99 from 750 W to 3.4 kW, THD less than 5% from half load to full load, a peak efficiency of 98.6 % at 240 V input, and 1000 W load. 3.2. Circuit Description and Steady State Analysis The phase shifted semi-bridgeless topology [84], shown in Figure 3-1, is proposed as another solution to address the problems outlined in chapter 1 for the conventional boost, bridgeless boost and interleaved boost topologies. The proposed topology has high efficiency at light loads and low lines, which is important to minimize the charger size, charging time, and the amount and cost of electricity drawn from the utility; the component count, which reduces the charger cost; and reduced EMI. The converter is ideally suited for automotive level 1 residential charging applications in North America where the typical supply is limited to 120V and 1.44kVA. 59 Figure 3-1: Phase shifted semi-bridgeless PFC boost topology The proposed topology introduces two more slow diodes (Da and Db) to the bridgeless configuration to link the ground of the PFC to the input line. However, the current does not always return through these diodes, so their associated conduction losses are low. This occurs since the inductors exhibit low impedance at the line frequency, a large portion of the current flows through the MOSFET intrinsic body diodes. Also the gating signals for MOSFETs are 180° out of phase. To analyze the circuit operation, the input line cycle has been separated into the positive and negative half-cycles, as explained in the sub-sections that follow. In addition, the detailed circuit operation depends on the duty cycle. Positive half-cycle operation analysis is provided for D > 0.5 and D < 0.5. 3.2.1. Positive Half Cycle Operation Referring to Figure 3-1, during the positive half-cycle, when the AC input voltage is positive, Q1 turns on and current flows through L1 and Q1 and continues through Q2 and then L2, returning to the line while storing energy in L1 and L2. When Q1 turns off, energy stored in L1 and L2 is released as current flows through D1, through the load and returns through the body diode of Q2/partially through Db back to the input. 60 3.2.2. Negative Half Cycle Operation Referring to Figure 3-1, during the negative half-cycle, when the AC input voltage is negative, Q2 turns on and current flows through L2 and Q2 and continues through Q1 and then L1, returning to the line while storing energy in L2 and L1. When Q2 turns off, energy stored in L2 and L1 is released as current flows through D2, through the load and returns split between the body diode of Q1 and Da back to the input. 3.2.1. Detailed Positive Half Cycle Operation and Analysis for D > 0.5 The detailed operation of the proposed converter depends on the duty cycle. During any half-cycle, the converter duty cycle is either greater than 0.5 (when the input voltage is smaller than half of output voltage) or smaller than 0.5 (when the input voltage is greater than half of output voltage). The three unique operating interval circuits of the proposed converter are provided in Figure 3-2 to Figure 3-4 for duty cycles larger than 0.5 during the positive half-cycle. Waveforms of the proposed converter during positive half-cycle operation with D>0.5 are shown in Figure 3-5. The intervals of operation are explained as follows. In addition, the ripple current components are derived, enabling calculation of the input ripple current, which provides design guidance to meet the required input current ripple standard. L1 D1 D2 Vin CO L2 Q1 Db L O A D Q2 Da Vg1 Vg2 Figure 3-2: Interval 1and 3: Q1 and Q2 are ON 61 Figure 3-3: Interval 2: Q1 ON, body diode of Q2 conducting Figure 3-4: Interval 4: Q1 OFF and Q2 ON Interval 1 [t0-t1]: At t0, Q1/ Q2 are on, as shown in Figure 3-2. During this interval, the current in series inductances L1 and L2 increases linearly and stores the energy in these inductors. The energy stored in Co provides energy to the load. The ripple currents in Q1 and Q2 are the same as the current in series inductances L1 and L2, where the ripple current is given by: ∆G6 & A -A+ 56 ) − H ': & 3-1 Interval 2 [t1-t2]: At t1, Q1 is on and Q2 is off, as shown in Figure 3-3. During this interval, the current in series inductances L1 and L2 continues to increase linearly and store the energy in these inductors. The energy stored in Co provides the load energy. The ripple currents in Q1 and body diode of Q2 are the same as the current in series inductances L1 and L2, where the ripple current is given by: 62 ∆G6 & A -A+ 56 1 − ) ': 3-2 Interval 3 [t2-t3]: At t2, Q1/Q2 are on again, and interval 1 is repeated, as shown in Figure 3-2. During this interval, the current in series inductances L1 and L2 increases linearly and stores the energy in these inductors. The ripple currents in Q1 and Q2 are the same as the ripple current in series inductances L1 and L2, as shown in equation (3-1). Interval 4 [t3-t4]: At t3, Q1 is off and Q2 is on, as shown in Figure 3-4. During this interval, the energy stored in L1 and L2 is released to the output through L1, D1, Q2 and L2. The ripple currents in D1 and Q2 are the same as the ripple currents in L1 and L2: ∆G6 & A -A+ 56 − 9 1 − ) ': 3-3 Figure 3-5: Phase shifted semi-bridgeless boost converter steady-state waveforms at D > 0.5 63 3.2.1. Detailed Positive Half Cycle Operation and Analysis for D < 0.5 The three unique operating interval circuits of the proposed converter are given in Figure 3-6 to Figure 3-8 for duty cycles smaller than 0.5 during the positive half-cycle. The waveforms of the proposed converter during these conditions are shown in Figure 3-9. The intervals of operation are explained as follows. Figure 3-6: Interval 1and 3: Q1 and Q2 are OFF, body diode of Q2 conducting Figure 3-7: Interval 2: Q1 ON, body diode of Q2 conducting Figure 3-8: Interval 4: Q1 OFF and Q2 ON 64 Figure 3-9: Phase shifted semi-bridgeless boost converter steady-state waveforms at D < 0.5 Interval 1 [t0-t1]: At t0, Q1/ Q2 are off, as shown in Figure 3-6. During this interval, the energy stored in L1 and L2 are released to the output through L1, D1, body diode of Q2 and L2. The ripple currents in D1 and body diode of Q2 are the same as the ripple currents in L1 and L2: ∆G6 & A -A+ 56 − 9 & H − ) ': 3-4 Interval 2 [t1-t2]: At t1, Q1 is on and Q2 is off, as shown in Figure 3-7. During this interval, the current in series inductances L1 and L2 continues to increase linearly and store the energy in these inductors. The energy stored in Co provides energy to the load. The ripple 65 currents in Q1 and the body diode of Q2 are the same as the current in series inductances L1 and L2, where the ripple current is given by: ∆G6 & A -A+ 56 )': 3-5 Interval 3 [t2-t3]: At t2, Q1/Q2 are off again, and interval 1 is repeated, as shown in Figure 3-6. During this interval, the current in series inductances L1 and L2 increases linearly and stores the energy in these inductors. The ripple currents in D1 and body diode of Q2 are the same as the ripple current in series inductances L1 and L2, as shown in equation (3-4). Interval 4 [t3-t4]: At t3, Q1 is off and Q2 is on, as shown in Figure 3-8. During this interval, the energy stored in L1 and L2 is released to the output through L1, D1, Q2 and L2. The ripple currents in D1 and Q2 are the same as the ripple currents in L1 and L2: ∆G6 & A -A+ 56 − 9 )': 3-6 The operation of converter during the negative input voltage half-cycle is similar to the operation of converter during the positive input voltage half-cycle. 3.3. Loss Modeling The estimated loss distribution of the semiconductors is provided in Figure 3-10 at 70 kHz switching frequency, 240V input and 3300W load for benchmark conventional boost and benchmark interleaved boost converters and the proposed phase shifted semi-bridgeless boost converter. The currents in regular diodes Da and Db were assumed to be split with the current going through intrinsic body diodes for phase shifted semi-bridgeless topology. The regular diodes input bridge rectifiers have the largest share of losses among the topologies with the input bridge rectifier. The phase shifted semi-bridgeless topology nearly eliminates this large loss component (~27.6W). However, the tradeoff is that the MOSFET losses are higher and the 66 intrinsic body diodes of MOSFETs conduct, producing new losses (~7.8W). The fast diodes in the benchmark conventional and benchmark interleaved PFC have slightly lower power losses, since the boost RMS current is higher in these topologies. 48.7 W 48.7 W 39.6 W 60 Conventional Boost 50 7.9 W 0W 0W 0W 10 19.1 W 20 8.3 W 8.3 W 30 Phase Shifted Semi-Bridgeless Boost 12.9 W 12.9 W 12.7 W 40 27.6 W 27.6 W Power Losses (W) Interleaved Boost Total Losses Devices / Total Losses Intrinsic Body Diodes FETs Fast Diodes Regular Diodes 0 Figure 3-10: Comparison of the estimated loss distribution in the semiconductors at 70kHz switching frequency, 240V input, 3300W load at 400V Overall the MOSFETs are under slightly more stress in phase shifted semi-bridgeless topology, but the total loss for the proposed phase shifted semi-bridgeless boost are 17% lower than the benchmark conventional boost and 7% lower than the benchmark interleaved boost . Since the benchmark converter bridge rectifier losses are large, it is expected that phase shifted semibridgeless boost converter should have the lowest losses among the topologies investigated. Additionally, it is noted that the losses in the input bridge rectifiers are 63% of total losses in the conventional PFC converter and 71% of total losses in the benchmark interleaved PFC converter. 67 Therefore, eliminating the input bridge in PFC converters is justified despite that the introduction of new losses. 3.4. Simulation Results PSIM simulation software was used to verify steady state waveforms of each component. Figure 3-11: PSIM simulation circuit for the phase shifted semi-bridgeless PFC boost converter: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz 68 Figure 3-11 shows the PSIM simulation circuits of the proposed BLIL PFC converter. As it can be seen, the power stage section of converter consists of two boost inductors, Ld1 and Ld2, two fast boost diodes, Db1 and Db2, two switches, Q1 and Q2 and their body diodes Dq1 and Dq2. Also it consists of two current loops and one voltage loop. The sensed input voltage is multiplied by the compensated output voltage, and then generates a control signal to be compared with the switching carrier waveforms to generate the gating signals for main MOSFETs. Figure 3-12 shows the PSIM simulation results of a phase shifted semi-bridgeless PFC boost converter. The input current is in phase with the input voltage, and it has close to unity power factor. Also the output voltage is regulated at around 400V, with a 120 Hz low frequency ripple. The converter is operating at 70 kHz switching frequency, 240 V input voltage and 3.4 kW output power. Figure 3-12: Simulation waveforms for the proposed phase shifted semi-bridgeless PFC boost converter: Output voltage, input voltage and input current: Vin = 240V, Vo = 400V, Po = 3400W and fsw = 70 kHz 69 3.5. Logic Implementation A standard two-phase interleaved CCM PFC controller from Texas Instrument, UCC28070 was used to implement the logic circuitry of the prototype unit. A simplified logic implementation of this controller applied to semi-bridgeless phase shifted PFC boost converter is given in Figure 3-13. Figure 3-13: Logic implementation of phase shifted semi-bridgeless PFC boost converter using UCC28070 controller 70 3.6. Experimental Results Prototypes of a phase shifted bridgeless boost converter and a benchmark interleaved boost converter were built to verify the proof-of-concept and analytical work presented in this chapter and to benchmark the proposed converter. The devices used in experimental prototypes are provided in Table 3-1. Table 3-1: Component / devices used in prototype unit Benchmark Interleaved PFC Phase Shifted Semi-bridgeless PFC Topology Components Used in Prototype Unit Head Device Part # / Value # of Devices Regular Diode 25ETS08S 2 Fast Diode IDB06S60C 2 MOSFET IPB60R099CP 2 Kool mu 77071 core 2 Inductors 60 turns / 400 µH Capacitors EKXJ451ELL820 / 82 µF 10 Regular Diode 25ETS08S 4 Fast Diode IDB06S60C 2 MOSFET IPB60R099CP 2 Kool mu 77071 core 2 Inductors 60 turns / 400 µH Capacitors EKXJ451ELL820 / 82 µF 10 Pictures of the proposed phase shifted bridgeless boost prototype are provided in Figure 3-14 to Figure 3-16. It consists of a capacitor bank of 820 µF, a control board and an IMS power board attached to a heatsink with the PFC inductors. 71 Figure 3-14: Capacitor bank of 820 uF Figure 3-15: Control board of PFC boost converter 72 Figure 3-16: IMS power board attached to a heatsink with the PFC inductors The experimental efficiency of the phase shifted bridgeless boost converter and benchmark interleaved boost converter is provided in Figure 3-17 for 240V input and Figure 319 for 120V input at 70 kHz switching frequency and 400 V output. Loss reduction curves as a function of output power are provided in Figure 3-18 and Figure 3-20 for 240V and 120V input, respectively. 73 100 99 Efficiency (%) 98 97 Interleaved PFC Converter 96 Phase Shifted Semi-Bridgeless PFC Converter 95 3500 3000 2500 2000 1500 1000 500 0 94 Output Power (W) Figure 3-17: Efficiency as a function of output power at Vin = 240V, Vo=400V and 70kHz switching 70 Loss Reduction (%) 60 50 40 30 20 10 3500 3000 2500 2000 1500 1000 500 0 0 Output Power (W) Figure 3-18: Loss reduction as a function of output power at Vin = 240V, Vo=400V and 70kHz for the proposed phase shifted semi-bridgeless converter compared to the benchmark interleaved boost converter 74 98 97 Efficiency (%) 96 95 94 Interleaved PFC Converter 93 Phase Shifted Semi-Bridgeless PFC Converter 92 1800 1600 1400 1200 1000 800 600 400 200 0 91 Output Power (W) Figure 3-19: Efficiency as a function of output power at Vin = 120V, Vo=400V and 70kHz 70 Loss Reduction (%) 60 50 40 30 20 10 1800 1600 1400 1200 1000 800 600 400 200 0 0 Output Power (W) Figure 3-20: Loss reduction as a function of output power at Vin = 120V, Vo=400V and 70kHz for the proposed phase shifted semi-bridgeless converter compared to the benchmark interleaved boost converter From the results, it is noted that proposed semi-bridgeless PFC converter achieves a peak efficiency of 98.6% at 1 kW output power. Additionally, the light load efficiency of the 75 proposed converter is significantly better than that of the benchmark interleaved PFC due to the absence of the input bridge rectifier. However, as the load increases, the efficiency drops due to additional heat dissipation in the intrinsic body diodes of the MOSFETs. 45 40 35 Vin=240 THD (%) 30 Vin=120 25 20 15 10 5 3500 3000 2500 2000 1500 1000 0 500 0 Output Power (W) Figure 3-21: THD as a function of output power at Vin = 120 V and 240V, Vo=400V and 70kHz 1.02 1 Power factor 0.98 0.96 0.94 0.92 Vin=240 0.9 Vin=120 0.88 0.86 3500 3000 2500 2000 1500 1000 500 0 0.84 Output Power (W) Figure 3-22: Power factor as a function of output power at Vin = 120 V and 240V, Vo=400V and 70kHz 76 2.5 Amplitude (A) 2 1.5 EN 61000-3-2 Class D Limits (A) Amplitude (A) Vin = 120 V Amplitude (A) Vin = 240 V 1 0.5 3 5 7 9 11 13 15 17 19 21 23 25 27 29 31 33 35 37 39 0 Harmonics Order Figure 3-23: Harmonics orders at Vin = 120 V and 240V, compared against EN61000-3-2 standard for Vin = 120 V at Po = 1700 W and 240 V at Po = 3400 W In order to verify the quality of the input current, the input current THD is shown in Figure 3-21. The power factor and harmonic orders are given and compared with EN 61000-3-2 standard in Figure 3-22 and Figure 3-23. It is noted that mains current THD is less than 5% from 50% load to full load and it is compliant to IEC 61000-3-2 (Figure 3-21 and Figure 3-23). The converter power factor is shown over entire load range for 120 and 240V input in Figure 3-22. The power factor is greater than 0.99 from 50% load to full load. Experimental waveforms from the proposed converter prototype are provided in Figure 3-24 through Figure 3-30. The input current, input voltage and output voltage are given in Figure 324. As it can be seen, the input current is in phase with the input voltage and has a sinusoidal shape. Additionally, there is a low frequency ripple on output voltage, which is inversely proportional to the value of PFC bus output capacitors. 77 In Figure 3-25, the inductor current is provided in addition to the above mentioned waveforms from Figure 3-24. It is noted that during the positive half-cycle, the inductor current is the same as the input current. However, during the negative half-cycle, the input current is partially flowing through slow diodes, Da and Db. In Figure 3-26 the inductor current, input current and current sensed in the MOSFET through a current transformer are given. The gating signals, sensed MOSFET current and the inductor current are provided for duty cycles less than 0.5, Figure 3-27, and greater than 0.5, Figure 3- 28. These waveforms match the theoretical models. Figure 3-29 and Figure 3-30 show the output voltage transient response to a change to the load form FL to NL and vice versa. As can be noted, the output voltage regulates right away. Output Voltage Ch1= Vo 100V/div. Input Current Ch4 = Iin 10A/div. Input Voltage Ch2 = Vin 100V/div. Figure 3-24: Input current, input voltage and output voltage. 78 Output Voltage Ch1= Vo 100V/div. Inductor Current Ch3 = IL1 10A/div. Input Current Ch4 = Iin 10A/div. Input Voltage Ch2 = Vin 100V/div. Figure 3-25: Input current, inductor current, input voltage and output voltage. Sensed MOSFET Current Ch1= IQ1 2V/div. Input Current Ch4 = Iin 10A/div. Inductor Current Ch3 = IL1/IDb 10A/div. Figure 3-26: Inductor current, input current and sensed MOSFET current. 79 Inductor Current Ch3 = IL1/IDb 10A/div. Sensed MOSFET Current Ch2= IQ1 2V/div. Gating Signal Ch1= Vg 10V/div. Figure 3-27: Gating signal, inductor and sensed MOSFET current for D < 0.5 Inductor Current Ch3 = IL1/IDb 10A/div. Sensed MOSFET Current Ch2= IQ1 2V/div. Gating Signal Ch1= Vg 10V/div. Figure 3-28: Gating signal, inductor and sensed MOSFET current for D > 0.5 80 Figure 3-29: Load transient response from NL to FL (0W to 3400 W) Ch1= Vo 100V/div. Ch2= Iin 10A/div. Ch4= Vo 100V/div Figure 3-30: Load transient response from FL to NL (3400 W to 0 W) Ch1= Vo 100V/div. Ch2= Iin 10A/div. Ch4= Vo 100V/div 81 3.7. Conclusion A new high performance phase shifted semi-bridgeless AC-DC Boost converter topology has been presented in this chapter for the front-end AC-DC converter in PHEV battery chargers. The proposed converter features high efficiency at light loads and low lines, which is critical to minimize the charger size, charging time and the amount and cost of electricity drawn from the utility; the component count, which reduces the charger cost; and reduced EMI. The converter is ideally suited for automotive level 1 residential charging applications in North America where the typical supply is limited to 120 V and 1.44 kVA. An analysis and performance characteristics are presented. A breadboard converter circuit has been built to verify the proof-of-concept. The theoretical waveforms were compared with the results taken from prototype unit. Additionally, key experimental waveforms were provided and input current harmonics at each harmonic order were compared more explicitly with the IEC 61000-3-2 standard limits. Experimental results demonstrate that the mains current THD is smaller than 5% from 50% load to full load and the converter is compliant with the IEC 61000-3-2 standard. The converter power factor was also provided for full power range at 120 and 240 V input. The power factor is greater than 0.99 from 50% load to full load. The proposed converter achieves a peak efficiency of 98.6 % at 240 V input and 1 kW output power. 82 CHAPTER 4. The Ripple Steering Technique and Converter Modeling 4.1. Introduction In this chapter, an average switch model approach to the power stage modeling, feedback compensation, and dynamic analysis of PFC boost converters with coupled magnetic filter is presented. The model is expressed by the derivation of power stage transfer functions for conventional boost converter, and then followed by the power stage transfer functions for PFC boost converter with coupled magnetic filters. Experimental and simulation results of a prototype boost converter converting universal AC input voltage to 400 V DC at 1.8 kW are given to verify the proof of concept, and analytical work reported in this chapter. The experimental result demonstrates that the model can correctly predict the steady-state and large signal dynamic behavior of a CCM PFC boost converter with coupled magnetic filter. 4.2. Average Switch Model Using the model discussed in section 1.3.6 applying that to modified boost converter with coupled inductors, any of the transfer functions between the output variables (output voltage and inductor current) and the input variables (input voltage and duty ration) can be derived. 4.2.1. PWM Switch Model of Conventional Boost Converter Applying the PWM switch model results in the equivalent circuit shown in Figure 4-1 for the conventional boost converter: 83 Figure 4-1: Conventional boost converter with PWM switch By application of superposition and other circuit theory, the power stage transfer functions can be derived: ;A $ 5 $ >? . h .[= :-& h .Ai .[= : +-Ai :-h &,0 + 4-1 >? .h &,0 h .Ai .[= : +-Ai :-h &,0 + 4-2 The low voltage loop cut off frequency of around 10 Hz is well below the LC resonant frequency (around 125 Hz), so the voltage loop transfer function can be approximated by: 5 $ >? ɳ.∆kl .h .[= : 4-3 4.2.1. Feedback Compensation Design of Conventional Boost Converter The compensation is selected so that the open-loop transfer function verifies the following criteria: 1- A high gain at low frequency in order to compensate the steady state error. 2- A gain slope maintained at -20 dB/dec around the crossover frequency in order to ensure enough phase margin and, therefore, closed-loop stability. 3- A very small gain at high frequency in order to reduce the influence of the switching harmonics and overall noise. 84 A. Current Loop Compensation Figure 4-2 shows the current loop plant and a type II compensator Bode plots. The current loop power stage has a cross over frequency at 10 KHz, so a type II compensator was chosen with the following pole and zero, as shown in Figure 4-3: 75 0 50 − 30 25 − 60 0 − 90 − 25 − 120 Plant Transfer Function Phase and Magnitude − 50 − 75 10 Phase (Degree) Magnitude (dB) Controller Type II Phase and Magnitude − 150 100 3 1×10 1×10 4 1×10 5 6 1×10 − 180 7 1×10 Frequency (Hz) Figure 4-2: Current loop plant and compensator type II Bode plots LB = 400 µH, Co = 1000 µF, RL = 88 Ω, R2 = 11.5 kΩ, C2 = 4.7 nF and C3 = 230 pF. C3 R2 C2 R1 Figure 4-3: Type II compensator network 85 The compensator has a pole at zero frequency, and another pole at m zero at mp & HP+ [+ & HP+ [+ nE n+ o nE and one . The values for compensator are: R2 = 11.5 KΩ, C2 = 4.7 nF and C3 = 230 pF. The open loop bode plot is given in Figure 4-4. As it can be seen, at cross over frequency, the open loop has a phase margin of 65° which is a stable design. − 90 100 80 Magnitude (dB) 40 − 126 20 0 − 20 − 40 − 60 − 80 − 100 3 1×10 − 144 Compensated Current Loop Plant (Open Loop) Phase and Magnitude 4 1×10 Phase (Degree) − 108 60 − 162 5 1×10 − 180 6 1×10 Frequency (Hz) Figure 4-4: Open loop Bode plot for current loop LB = 400 µH, Co = 1000 µF, RL = 88 Ω, R2 = 11.5 kΩ, C2 = 4.7 nF and C3 = 230 pF. B. Voltage Loop Compensation The voltage loop transfer function and compensator network Bode plot is given in Figure 4-5. The compensator for the voltage is a type II compensator as well. In the compensator network, voltage error amplifier gain is adjusted with compensation components to attenuate the twice- 86 line frequency ripple on the output capacitor to obtain a desired reduction of 3rd harmonic THD. C3 sets the reduction level, R2 sets the phase margin to 45 degrees at cross over frequency, and C2 sets the beginning of the phase boost. The desired cross over frequency for voltage loop is around 10 Hz, where the line frequency varies from 50 Hz to 60 Hz. The values for compensator are: R2 = 75 KΩ, C2 = 1 µF and C3 = 100 nF. Figure 4-6 shows the open loop Bode plot of compensated voltage loop. As it can be seen, it has a high gain at low frequencies, and at cross over frequency, the open loop has a phase margin of 55° which is a stable design. 100 0 80 − 36 60 Plant Transfer Function Phase and Magnitude 20 − 72 0 Controller Type II Phase and Magnitude − 20 − 108 Phase (Degree) Magnitude (dB) 40 − 40 − 60 − 144 − 80 − 100 0.1 1 10 100 − 180 3 1×10 Frequency (Hz) Figure 4-5: Voltage loop plant and compensator type II Bode plots LB = 400 µH, Co = 1000 µF, RL = 88 Ω, R2 = 75 kΩ, C2 = 1 µF and C3 = 100 nF. 87 100 − 45 80 − 72 60 − 99 20 Compensated Voltage Loop Plant (Open Loop) Phase and Magnitude 0 − 20 − 126 Phase (Degree) Magnitude (dB) 40 − 40 − 60 − 153 − 80 − 100 0.1 1 10 100 − 180 3 1×10 Frequency (Hz) Figure 4-6: Open loop Bode plot for voltage loop LB = 400 µH, Co = 1000 µF, RL = 88 Ω, R2 = 75 kΩ, C2 = 1 µF and C3 = 100 nF. 4.3. Ripple Steering Techniques in PFC Application Although the control strategy of PFC stage is similar to that of a conventional boost converter, but their power stage transfer functions are different. And no modeling has been done to verify the effect of added coupled filter to the power stage transfer functions, and thus the design of feedback loop compensator. Figure 4-7: Modified PFC boost converter with coupled inductors 88 Application of PWM switch model discussed in section 1.3.6 is used for the circuit shown in Figure 4-7 for boost converter with coupled inductors in order to get the voltage and current plant transfer functions. 4.3.1. PWM Switch Model of Boost Converter with Coupled Inductor In order to model the boost converter with coupled inductors, the winding arrangements shown in Figure 4-8 can be described mathematically by: 5& 5H where: r q& qH 6 6+ +r +r s tq& qH 6+ 4-4 6 4-5 4-6 M is the mutual inductance and k is the coupling coefficient of the windings. Figure 4-8: Equivalent circuit of coupled inductors From above equations, Figure 4-8 can be replaced with its equivalent circuit shown in Figure 4-9 [71]. Figure 4-9: Equivalent circuit of coupled inductors 89 The equivalent circuit of coupled inductor can be shown according to: 5& 5H q& − r qH − r 6 6+ +r +r 6 6 + + 6+ 4-7 6+ 4-8 Figure 4-10: Modified boost converter with PWM switch Figure 4-10 shows the equivalent circuit of boost converter with a coupled inductor, where active and passive switches are replaced by the averaged model of PWM switch. By application of superposition and other circuit theory, the power stage transfer functions can be derived: ;A $ 5 $ >? . h .[= :-& .[[= A+ ,v : +-&] 0 : 4-9 0 : 4-10 >? .h &,0 .[[= A+ ,v : +-&] Where: )< $ ) H [xA . y . y: . r qH − 3r ] $ Z + [y: . r 2qH − 3r + xA . y . qH ] $ D + xA [y: . qH 1 − + 2y: . r 1 − ) H + 2y . r] $ H + 2r $ + xA 1 − ) H 4-11 The voltage loop transfer function approximation is valid here as well, so equation (4-5) can be simplified as equation (4-3). Figure 4-11 shows the Bode plot of current loop transfer functions for modified boost converter with coupled inductor and a conventional boost converter. 90 As it can be noted, the phase margins for both plants are the same and steady at -90 °. But the main difference is the magnitude of current plant transfer functions, where for conventional boost has lower cross over frequency and lower gain. If the same feedback compensation network is used for both converters, the steady state and large signal responses would not be 75 0 50 − 30 Modified Boost Plant Transfer Function Phase and Magnitude 25 − 60 − 90 0 − 25 − 120 Conventional Boost Plant Transfer Function Phase and Magnitude − 50 − 75 3 1×10 Phase (Degree) Magnitude (dB) adequate. − 150 4 1×10 5 1×10 6 1×10 − 180 7 1×10 Frequency (Hz) Figure 4-11: Current loop plant Bode plots LB = 400 µH, Co = 1000 µF, RL = 88 Ω, Ldc = 400 µH, Lac = 250 µH, M = 260 µH, Cs = 1 µF. 91 4.3.1. Feedback Compensation Design of Boost Converter with Coupled Inductor Figure 4-12 shows the current loop plant and a type II compensator Bode plots. The current loop power stage has a cross over frequency at 20 KHz, so a type II compensator was chosen with the following pole and zero: pole at m and one zero at mp & HP+ [+ nE n+ o nE & HP+ [+ . The values for compensator are: R2 = 22.1 KΩ, C2 = 1 nF and C3 = 150 pF. The open loop Bode plot is given in Figure 4-13. As it can be seen, at the cross over frequency, the open loop has a phase margin of 50° which is a stable design. 0 75 − 30 50 − 60 − 90 25 Controller Type II Phase and Magnitude 0 Modified boost Plant Transfer Function Phase and Magnitude − 120 − 25 − 50 10 Phase (Degree) Magnitude (dB) 100 − 150 100 3 1×10 1×10 4 1×10 5 6 1×10 − 180 7 1×10 Frequency (Hz) Figure 4-12: Current loop plant and compensator Bode plots for boos converter with coupled inductor RL = 88 Ω, Ldc = 400 µH, Lac = 250 µH, M = 260 µH, Cs = 1 µF, R2 = 22.1 kΩ, C2 = 1 nF and C3 = 150 pF. 92 − 100 150 125 Magnitude (dB) 75 − 140 50 25 − 160 0 Phase (Degree) − 120 100 − 25 Compensated Current Loop Plant (Open Loop) Phase and Magnitude − 50 − 75 − 100 10 100 1×10 3 1×10 4 − 180 5 1×10 − 200 6 1×10 Frequency (Hz) Figure 4-13: Open loop plant Bode plot for boost converter with coupled inductor RL = 88 Ω, Ldc = 400 µH, Lac = 250 µH, M = 260 µH, Cs = 1 µF, R2 = 22.1 kΩ, C2 = 1 nF and C3 = 150 pF. 4.4. Simulation Results PSIM simulation was used to verify the feedback loop design for steady state and large signal perturbation. Figure 4-14 shows the simulation results of boost converter with coupled inductors, converting universal AC input voltage to 400 V DC at 240 V input and 1.6 kW. Figure 4-15 illustrates: Top: The boost inductor current in a conventional boost converter - No Filtering technique. Middle: The boost inductor current with ripple steering technique applied, and Bottom: Series capacitor current in the coupled inductor filter. 93 Figure 4-14: PSIM simulation circuit for ripple steering technique applied to PFC boost converter 12 10 8 6 4 2 0 -2 I(M1_1) 12 10 8 6 4 2 0 -2 I(M1_2) 4 2 0 -2 -4 0.16 0.17 0.18 0.19 0.2 Figure 4-15: Inductor Current - no filtering technique (Top) – ripple steering technique (Middle), series capacitor current (Bottom): Vin = 240 V, Vo = 400 V, Po = 1600 W, fsw = 70 kHz 94 4.5. Experimental Results Prototypes of the boost converter with coupled inductors and conventional boost converter were built to verify the proof-of-concept and analytical work presented in this thesis and to benchmark the proposed compensation network design. Figure 4-16 shows a coupled inductor used for this experiment. The inductor values are: Ldc = 400 uH, Lac = 250 uH, M = 265 uH and the series capacitor Cs = 1 uF. Figure 4-17 and Figure 4-18 show the experimental and simulation ripple current waveform in dc inductor under the following operating conditions: Vin = 120 V, Iin = 15 A, Po = 800 W, Vo = 400 V and 70 kHz. Figure 4-16: Coupled inductors used in experimental circuit - Top left: DC inductor, Top right: AC inductor, Bottom: Coupled inductor 95 Figure 4-17: Inductor current Idc ripple at 120 V input and 800 W output - Experimental Figure 4-18: Inductor current Idc ripple at 120 V input and 800 W output - Simulation 96 Figure 4-19: Inductor current Iac at 120 V input and 800 W output - Experimental ac Inductor Current 8 6 4 2 0 -2 -4 -6 -8 0.17 0.175 0.18 0.185 Time (s) 0.19 0.195 0.2 Figure 4-20: Inductor current Iac at 120 V input and 800 W output - Simulation 97 Figure 4-21: Peak inductor current Iac ripple at 120 V input and 800 W output - Experimental Figure 4-22: Peak inductor current Iac ripple at 120 V input and 800 W output - Simulation 98 Figure 4-23: Inductor current Idc ripple at 240 V input and 1600 W output - Experimental Figure 4-24: Inductor current Idc ripple at 240 V input and 1600 W output - Simulation 99 Figure 4-25: Inductor current Iac at 240 V input and 1600 W output - Experimental Figure 4-26: Inductor current Iac at 240 V input and 1600 W output - Simulation 100 Figure 4-27: Peak inductor current Iac ripple at 240 V input and 1600 W output - Experimental Figure 4-28: Peak inductor current Iac ripple at 240 V input and 1600 W output - Simulation 101 Figure 4-19 and Figure 4-20 show the experimental and simulation current waveforms in ac inductor under the following operating conditions: Vin = 120 V, Iin = 15 A, Po = 800 W, Vo = 400 V and 70 kHz. Figure 4-21 and Figure 4-22 show the experimental and simulation ripple current waveforms in ac inductor under the same operating conditions as above. Figure 23 through Figure 28 repeats the same waveforms for different test condition: Vin = 240 V, Iin = 15 A, Po = 1600 W, Vo = 400 V and 70 kHz. 4.6. Conclusion An averaged PWM model for boost converters with ripple steering technique has been developed and effect of ripple steering technique on properly design of compensation network for the current loop has been studied. The analytical and simulation results were compared with experimental results of a breadboard converter circuit converting universal AC input voltage to 400 V DC at 240 V input and 1.6 kW. The proposed model shows an accurate prediction of steady state and large transient behavior of boost converter with coupled inductors. 102 CHAPTER 5. Conclusions and Future Work 5.1. Conclusions As the adaptation rate of PHEVs increases, the stress on the utility grid is projected to increase significantly at times of peak demand. Therefore, efficient and high power factor charging is critical in order to minimize the utility load stress, and reduce the charging time. In addition, a high power factor is needed to limit the input current harmonics drawn by these chargers and to meet regulatory standards. To meet the next generation requirements of these applications, two new topologies and two new modeling techniques have been proposed in this thesis. 5.1.1. Bridgeless Interleaved Boost PFC Converter The first contribution is a new bridgeless interleaved boots PFC converter for plug in hybrid electric vehicle level 2 charger applications. The proposed topology achieves a peak efficiency of 98.94% was reached at 265 V input and 1.2 kW output power. High efficiency over the entire load range is achieved in this topology, enabling fewer problems with heat dissipation and cooling systems. Furthermore, a higher efficiency means more power is available from the mains feed to charge the batteries, reducing charging times and electricity costs. 5.1.2. Phase Shifted Semi-Bridgeless Boost PFC Converter The second contribution is a new phase shifted semi-bridgeless boost PFC converter for plug in hybrid electric vehicle level 1 charger applications. Experimental results demonstrate that the mains current THD is smaller than 5% from 50% load to full load and the converter is compliant with the IEC 61000-3-2 standard. The converter power factor was also provided for 103 full power range at 120 and 240V input. The power factor is greater than 0.99 from 50% load to full load. The proposed converter achieves a peak efficiency of 98.6 % at 240 V input and 1 kW output power. 5.1.3. New Loss Modeling for PFC Boost Converters In order to properly select the power stage components of a converter and calculate the associated power losses, it is necessary to determine the RMS and average values of their currents. In a typical boost converter, the MOSFET and diode current waveforms are pulsedwidth modulated, with both the duty cycle and peak amplitude varying with the AC input. And without an effective mathematical method for computing these RMS and average values, the proper design and selection of power stage components can be flawed. A new analytical model for the four different topologies was developed, enabling the calculation of power losses and efficiency calculation. 5.1.1. New Average Modeling for PFC Boost Converters with Coupled Inductors An averaged PWM model for boost converters with ripple steering technique has been developed and effect of ripple steering technique on properly design of compensation network for current loop has been studied. The analytical and simulation results were compared with experimental results of a breadboard converter circuit converting universal AC input voltage to 400 V DC at 240 V input and 1.8 kW. The proposed model shows an accurate prediction of the steady state and large transient behavior of boost converter with coupled inductor. 104 5.2. Future Work This sub-section outlines the possible future work for the thesis topics. 5.2.1. LLC Resonant Converter for DC/DC Stage The front end PFC section of a PHEV charger is followed by a DC/DC section to complete the charger system. This stage can be an LLC resonant converter. High efficiency (> 98%), high output voltage (~ 450 V), high power (~ 3.3 KW) and wide output voltage range (220 V to 450 V) makes it a challenging research topic. 5.2.2. Resonant PFC Converter Most of hard switching PFC boost topologies have been discussed in this dissertation, but a new class of topology for PFC is resonant PFC converters. Recent research activities have been conducted by Infineon Technology on Bridgeless Interleaved resonant PFC and also by Dr. Slobodan Ćuk from TESLAco on Bridgeless Resonant PFC. 5.2.3. Level 3 Chargers With cities having it mandatory to have a Level 3 charger installed in every high rise build in the lower main island in a near future, the business case is already justified. The ultimate in charge technology, but nobody has yet really settled on a standard, leaving manufacturers of cars and charging equipment to come up with their own. This type of charging is defined as any charging above 14.4 kilowatts. Also, most Level 3 stations are considered “fast chargers,” but they don’t have to be. The conventional wisdom is that if an average EV battery can be charged to full in about a half hour, then it is a fast charge. Interleaved resonant PFC and also by Dr. Slobodan Ćuk from TESLAco on Bridgeless 105 5.2.4. 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Dunford, "A Phase Shifted Semi-Bridgeless Boost Power Factor Corrected Converter for Plug in Hybrid Electric Vehicle Battery Chargers," in IEEE Applied Power Electronics Conference and Exposition, APEC Fort Worth, TX, 2011. 111 Appendix MathCAD Simulation: Power Component Selection, Magnetic Design and Compensator Network Design Part 1 - Power Component Selection Design Variables Definition fin := 50 Line Frequency Vinmin := 85 Minimum Input Voltage Vinmax:= 265 Maximum Input Voltage VinHL := 230 High Line Input Voltage VinLL := 115 Low Line Input Voltage η1 := 0.985 Converter Effeciency at High Line η2 := 0.965 Converter Effeciency at Low Line Iin := 16 Maximum RMS Input Current Pin1 := Iin⋅ VinHL Maximum Input Power at High Line Pin2 := Iin⋅ VinLL Maximum Input Power at Low Line Pout1 := Pin1⋅ η1 Maximum Output Power at High Line Pout1 = 3.625× 10 Pout2 := Pin2⋅ η2 Maximum Output Power at Low Linw Pout2 = 1.776× 10 Vboost := 400 Boost Output Voltage 3 fs := 175⋅ 10 Minimum Switching Frequency ∆Irp := 50 Maximum boost ripple current as a percentage 3 3 Boost FET duty Cycle: δF_LL( θ) := 1 − 2⋅ Vinmin ⋅ sin ( θ) Vboost δF_HL( θ) := 1 − 2⋅ Vinmax ⋅ sin ( θ) Vboost 112 Duty Ratio PFC Boost Converter Duty Ratio v.s. Angle 0.8 δF_LL ( θ ) 0.6 δF_HL ( θ )0.4 0.2 0 0 1.571 3.142 4.712 6.283 θ Angle (Rad.) π DminLL := δF_LL 2 π DminHL := δF_HL 2 Minimum Duty Cycle at High Line Voltage DminLL = 0.699 Minimum Duty Cycle at Low Line Voltage DminHL = 0.063 Maximum Duty Cycle Dmax := 0.95 Boost Diode Duty Cycle: Duty Ratio δD_LL( θ) := 2⋅ Vinmin ⋅ sin ( θ) Vboost δD_HL( θ) := 1 − δF_HL( θ) 0.8 δD_LL( θ ) 0.6 δD_HL( θ )0.4 0.2 0 0 1.571 3.142 4.712 6.283 θ Angle (Rad.) PFC Boost Inductor Selection: Iin IL := =8 2 Maximum RMS Inductor Current per Phase Ipk := Peak Inductor Current per Phase 2⋅ IL = 11.314 ∆I Lb_LL := ∆Irp 100 Ipkabs := Ipk + 1 2 ⋅ Ipk = 5.657 Boost Inductor Current Ripple ⋅ ∆I Lb_LL = 14.142 Absolute Peak Inductor Current 113 ∆I Lb_HL := Vinmax⋅ 2⋅ DminHL fs ⋅ Lb Current Ripple at High Line = 3.142 2 Since Inductor in each Phase will be divided by two equal inductors, so the final indcutor value is 43 uH. Boost FET Selection: iL( θ) := Instantaneous Boost Inductor Current per Phase 2⋅ IL⋅ sin ( θ) isw ( θ) := iL( θ) ⋅ δF_LL( θ) Instantaneous Boost Switch Current iD( θ) := iL( θ) ⋅ δD_HL( θ) Instantaneous Boost Diode Current isw_rms := 1 ⌠ 2π ⌡ 2π ( isw(θ) )2 dθ RMS Boost Switch Current isw_rms = 5.974 0 The FET intrinsic body diode current is exactly the same as boost diode current. 1 ⌠ iD_ave := 2π ⌡ 2π (iD(θ) ) dθ RMS Boost Diode Current 0 iD_ave = 0 Boost FET Loss Calculation: Pcond ( Rds ) := Rds ⋅ isw_rms Pcoss ( Coss ) := 1 2 2 ⋅ Coss ⋅ Vboost ⋅ fs Pgate ( Qg , Vg) := Qg⋅ Vg⋅ fs Psw ( Tr , Tf ) := 1 i ⋅ ( Tr + Tf ) ⋅ Vboost ⋅ fs 2 sw_rms Pid ( Vid) := Vid⋅ iD_ave Conduction Loss Die Output Capacitor Loss Gating Losse Switching Loss Intrinsic body diode Loss 114 Boost Inductor Core Selection: Lb −5 = 4.3 × 10 Lb1 := 2 3 Lb1⋅ 10 ⋅ Ipkabs 2 = 8.6 mH - Amperes^2 −5 Lb = 8.6 × 10 From the following chart, Kool Mu cores with 60u permability can be used. Kool Mu Core Selector Chart Selected CORE 77894 Core Data: µ per := 60 Permability of the core A L := 61 Aw := 1.56 Window Area cm^2 le := 6.35 Path length cm Ve := 4.15 Volume cm^3 MLT := 5.23 Winding Factor 100% cm/Turn 115 Determining Number of Turns: Assuming 50% roll off, winding for: 1 −5 ⋅ L = 8.6 × 10 0.5 b1 1 Nt := ⋅ L ⋅ 1000 0.5 b1 −6 = 37.548 Calculated Number of Turns A L⋅ 10 Final Number of Turns Ntf := 25 Hor := 0.4⋅ π ⋅ Ntf ⋅ Ipkabs = 69.967 le DC Magnetizing force From the above curve, the permeability vs DC bias showes a 61% initial permeability at full load. −3 2 Lb_FL := 0.61⋅ Ntf ⋅ A L⋅ 10 2 −3 Lb_NL := Ntf ⋅ A L⋅ 10 = 23.256 = 38.125 uH Inductance at FL uH Inductance at NL Calculating the Core and Copper Losses: The wire used is AWG#16: 116 Rpu := 0.1319 mΩ/cm Wa := 0.0152 Wire Area cm^2 Ff := Ntf ⋅ Wa Aw = 0.244 Ipk_max := Ipk + 0.5⋅ ∆I Lb_LL = 14.142 Peak Inductor Current Ipk_min := Ipk − 0.5⋅ ∆I Lb_LL = 8.485 Min Inductor Current Hor_max := Hor_min := 0.4⋅ π ⋅ Ntf ⋅ Ipk_max le 0.4⋅ π ⋅ Ntf ⋅ Ipk_min le = 69.967 Max DC Magnetizing force = 41.98 Min DC Magnetizing force From the following curve for 60u Material: Bmax := 3800 Maximum Flux Density Gauss Bmin := 2500 Minimum Flux Density Gauss ∆B := Bmax − Bmin 1 2 ⋅ ∆B = 650 117 Pc_pu := 1 ∆B 2⋅ 3 10 Pc := Pc_pu ⋅ Ve 1000 2 ⋅ 1000 fs 1.46 Core losses mW/cm^3 = 795.54 Core losses W = 3.301 Coil resistivity mΩ Rdc := MLT ⋅ Ntf ⋅ Rpu = 17.246 Pcu := Rdc 2 DC Copper Losses W ⋅ I = 1.104 1000 L Wire Gage factor for AWG#16 Kf := 14.25 Rac := Rdc 100 fs ⋅ Kf ⋅ 6 AC Resistance at Switching frequency = 1.028 10 2 ∆I Lb_LL Pac := Rac⋅ = 2.742 2⋅ 3 AC Copper Losses Ptot := Pc + Pcu + Pac = 7.147 Total Inductor losses per phase PFC Bus Capacitor Selection: Capacitance Value Tholdup := 1 1 −3 ⋅ = 2.5 × 10 8 50 Minimum Hold Up time for 50Hz Maximum Allowable PFC Voltage Droop Vpfc_droop := 20 Io := Pout1 Vboost PFC DC Current = 9.062 2⋅ Pout1 ⋅ Tholdup Cpfc := 2 Vboost − ( Vboost − Vpfc_droop ) −3 Cpfc = 1.162× 10 2 Minimum Required Bus Capacitance 118 Capacitors RMS Current: 16⋅ Vboost ichf_t := Io⋅ 6⋅ π ⋅ 2⋅ Vinmin⋅ η1 ichf_lf := Io 2 − 1 = 12.528 Total rms ripple current of PFC Caps 2 x Line Frequency rms ripple current = 6.408 2 16⋅ Vboost ichf_hf := Io⋅ 6⋅ π ⋅ 2⋅ Vinmin⋅ η1 2 − 1.5 = 10.765 High Frequency Ripple current of PFC Caps Part 2 - Control Component Selction Definition Device Parameters Refrence and Internal Bias Voltage VREF := 6 −6 Current Amplifier Ox Transconductance g mv := 70⋅ 10 −6 Voltage Amplifier O Transconductance Vramp := 4 PWM ramp amplitude VRDM := 3 Timing regulation Voltage at RDM VRT := 3 Timing regulation Voltage at RT VDMAX := 3 Timing regulation Voltage at DMAX g mi := 100⋅ 10 1 - Component Selection for RT and DMX: Calculate RRT for target switching frequency: 3 RRT := 7500⋅ 10 fsw 3 RRT := 43.2⋅ 10 target fsw := 175 KHz 4 = 4.286 × 10 Standard Value chosen 119 Actual nominal switching frequency (KHz) will be: 3 fsw := 7500⋅ 10 fsw = 173.611 RRT KHz Power dissipation is negligible. Select a suitable size and power rating for the resistor. Calculate RRDM for desired maximum duty cycle: Target Dmax := 0.95 ( 4 ) RDMX = 3.888× 10 RDMX := RRT⋅ 2Dmax − 1 3 RDMX := 39.2⋅ 10 Standard Value chosen 1 RDMX Dmax := + 1 2 RRT Actual Dmax = 0.954 2 - Component Selection for CDR and RDM frequency Dithering: KHz Frequency Dithering Range fdm := 0.07⋅ fsw = 12.153 3 RRDM := 937.510 ⋅ fdm 4 = 7.714× 10 3 RRDM := 76.8⋅ 10 Standard Value chosen fdr := 1 KHz Frequency Dithering Range CRDM := 66.7⋅ CRDM := 5100 RRDM 1000 fdr = 5.123× 103 pF pF 3 - Component Selection - Vsense snd Vinac Resistor Configuration: Choose the voltage feedback resistor divider network: VREFfb = 3 VREFfb := 3 desired target VOUT_AVG := 400 120 3 Voltage divider resistor B RB := 24.9⋅ 10 VOUT_AVG RA := RB⋅ VREFfb 6 − 1 = 3.295 × 10 6 RA = R1 + R2 + R3 Voltage divier resistors A R1 := 1.1⋅ 10 RA := 3⋅ R1 2 PRdiv := VOUT_AVG RA + RB VOUT := PRdiv = 0.048 Total power dissipation RA R + 1 ⋅ VREFfb B VOUT = 400.59 Select Components for Vinac Divider: Vinac sense resistors must be the same divider-ratio as that of the Vout sense resistors. Perferably, they would be the same values and ratings. 4 - Size current-sense resistor for a 3.0V dynamic range: Choose current sense transformer turns ratio NCT: Rs := 3⋅ NCT Ipk ⋅ 1.20 NCT := 100 (Use 120% of Ipk to allow for a little extra current/power to recover from large load-step or line-step changes, or line drop-outs.) Rs = 22.097 Choose a standard value: Rs := 22.1 2 Ipk Rs P_rs := ⋅ 2 N 2 CT (Power will actually be somewhat lower than this because of less than 100% duty-cycle of Ipk.) P_rs = 0.141 Choose a 1/4-watt resistor. 121 5 - Select components for Pulse by Pulse Current limiting: a. Calculate the peak limit voltage: Peak current limit is a cycle-by-cycle limit which determines the absolute peak value of MOSFET drain current allowable (per phase). If this limit is exceeded, the current PWM cycle is shut down. The peak value of the inductor ripple current must be included, as well as any multipling factor used in the average current limit calculation. Vpklmt := ( Ipk + 0.5⋅ ∆I Lb_LL) ⋅1.20 ⋅ Rs NCT Vpklmt = 3.75 b. Calculate Ipeak Resistor Divider Limit current out of Vref to less than 0.5mA to minimize loading, since these resistors will be off of the Vref. VREF ( 4 −3 = 1.2 × 10 −3 VREF⋅ 0.5⋅ 10 ) = 3 × 10− 3 0.5⋅ 10 ( ) Vpklmt 4 3 ⋅ 1.2 × 10 = 7.501× 10 VREF Select standard bottom divider-resistor value Rpklmt1 := 3 Rpklmt2 := 7.5⋅ 10 VpklmtRpklmt2 ⋅ VREF 3 Rpklmt1 = 4.688× 10 Select standard top divider-resistor value 3 Rpklmt1 := 4.53⋅ 10 Resistor power dissipation is negligible; choose any suitable size and rating. Actual peak limit voltage: Rpklmt2 Vpklmt := VREF⋅ Rpklmt1 + Rpklmt2 Actual peak MOSFET current per phase: Ipeak := Vpklmt ⋅ NCT Rs Vpklmt = 3.741 Ipeak = 16.926 122 Make sure PKLMT voltage is set properly to: a) co-ordinate with low-line power limit, b) account for PKLMT detection propagation delay, and c) allow for additional saturation swing at extra power levels. 6 - Current Synthesizer: RB kR := RA + RB The Resistor-Divider Atenuation at the Vsense and Vinac 6 Boost Inductor uH LB := Lb ⋅ 10 = 86 RSYN := 10⋅ NCT⋅ LB⋅ kR = 29.143 Rs KΩ Selected Standard Resistor RSYN := 30.1 7 - Multiplier Set up: VAOmax := 5 Output of Voltage-error Amplifier kVFF := 0.398 Min Voltage Feed Forward Factor VVINAC := 0.76 Lowest Max Power Limit −6 17⋅ 10 Imo_max := VINAC := 73 Iin_pk := 2⋅ ( ) ⋅ VVINAC⋅ VAOmax − 1 −4 Imo_max = 1.298 × 10 kVFF Accounting for 2-V bridge drop Pout1 ⋅ 1.1 η1 ⋅ 1 Iin_pk = 78.421 VINAC Ipkabs ⋅ Rs RIMO := NCT⋅ Imo_max Choose standard value: 4 RIMO = 2.407× 10 3 RIMO := 24.9⋅ 10 123 8 - Current Loop Compensation Design: i := 1 .. 2000 ( i − 200) f := 100 500 i w := f i i s := 2⋅ π ⋅ j ⋅ w i i Vout := 400 For I-loop compensation calculations, set Lboost to mid-way between max and min swing values (50uH to ~100uH). −6 Lboost := 86⋅ 10 3 select target cut off frequency fcxo := 16⋅ 10 Vout⋅ Gip := i Rs NCT Current-Loop Plant Transfer Function: s ⋅ Lboost ⋅ Vramp i ( ) dBGip := 20⋅ log Gip i i ( ( i) ( i)) φGip := atan2 Re Gip , Im Gip ⋅ i 180 π Initial I-loop compensation target values: 1 Rz := 3 Vout⋅ gmi Cz := = 3.912× 10 Rs NCT 2⋅ π ⋅ fcxo ⋅ Lboost ⋅ Vramp 1 2⋅ π ⋅ fcxo −9 = 7.628× 10 ⋅ Rz 3 Cp := 1 2⋅ π ⋅ ( 2⋅ fcxo) ⋅ Rz −9 = 1.271× 10 124 3 Rz = 3.912× 10 −9 Cz = 7.628× 10 −9 Cp = 1.271× 10 (Note: values for Rzi, Czi, Cpi for the following GCEA equation are entered as global variables defined on the next page, below.) 3 Rzi ≡ 4⋅ 10 − 12 Czi ≡ 7500⋅ 10 − 12 Cpi ≡ 1200⋅ 10 GEA := i ( ) g mi⋅ 1 + s ⋅ Rzi⋅ Czi i Compensator Transfer Function Cpi⋅ Czi s ⋅ ( Czi + Cpi) ⋅ 1 + s ⋅ Rzi⋅ i i Cpi + Czi dBGEA := 20⋅ log GEA i i φG EA := atan2 Re GEA , Im GEA ⋅ i i 180 i π Current Loop Plant & Compensator Transfer Function 40 0 26.667 − 30 13.333 − 60 φGip i dBGip i i − 90 φG EAi − 13.333 − 120 − 26.667 − 150 dBGEA 0 − 40 10 100 3 1×10 1×10 4 5 1×10 6 1×10 − 180 7 1×10 fi 125 Current Loop Gain (open-loop transfer function): Vary parameters to see how crossover is affected: Rs Vout⋅ g ⋅ 1 + s ⋅ Rzi ⋅ Czi NCT ) mi ( i ⋅ GOL := s ⋅ Lboost ⋅ V i Cpi ⋅ Czi ramp s ⋅ ( Czi + Cpi) ⋅ 1 + s ⋅ Rzi⋅ i i i Cpi + Czi dBGOL := 20⋅ log GOL i i φG OL := atan2 Re GOL , Im GOL ⋅ i i 180 i π PM_GOL := φG OL + 180 i i Current Open Loop Bode Plot 100 90 80 60 36 40 − 18 φG OL 20 dBGOL i 0 i PM_G OL − 20 − 72 i − 40 − 60 − 126 − 80 − 100 100 3 1×10 4 1×10 1×10 5 − 180 6 1×10 fi Phase margin (PM) should be ~ 45deg at the cross-over frequency fcxo as parameters vary. 126 9 - Voltage Loop Compensation Design: a. Evaluate Modulator and Power Stage Gain Pin := Pout1 η1 3 = 3.68 × 10 ∆Vea := 3.3 −6 Cpfc := 1200⋅ 10 Pin1 Gvp := ∆Vea ⋅ Cpfc⋅ Vout⋅ s i PFC Voltage-Loop Boost Plant Transfer Function: i This approximation is valid for frequencies well below the LC resonance frequency. The low voltage-loop crossover freq helps ensure this condition is true. dBGvp := 20⋅ log Gvp i i φG vp := atan2 Re Gvp , Im Gvp i i 180 ⋅ i π Voltage Loop Plant Transfer Function − 45 100 80 − 63 60 40 − 81 20 dBGvp i φG vp 0 − 20 i − 99 − 40 − 60 − 117 − 80 − 100 0.1 1 10 100 − 135 3 1×10 fi 127 b. Evaluate Compensation Scheme H1 is "gain" of the VSENSE resistor-divider network. VREFfb := 3 H1 := Vout = 400 VREFfb 1 −3 H1 = 7.5 × 10 Vout H1 = 133.333 From above, Bulk Capacitor zero-peak ripple voltage: v Clf := Pin1 v Clf = 12.202 2⋅ π ⋅ 100Cpfc⋅ Vout −3 for Cpfc = 1.2 × 10 Voltage error amplifier gain is adjusted with compensation components to attenuate the twice-line frequency ripple (vClf) on the output capacitor to obtain a desired reduction of 3rd harmonic THD. Cpv sets the reduction level, Rzv sets the phase margin to 45degrees at fvxo, and Czv sets the beginning of phase boost. %k3rd := 1.5 f2LF := 100 −3 ( ) Zov f2LF := Cpv := %k3rd⋅ 66 × 10 ⋅ Vout⋅ 2π f2LF⋅ Cpfc g mv⋅ H1⋅ Pin1 1 −7 ( ) 2π f2LF⋅ Zov f2LF g mv⋅ H1⋅ Pin1 fvxo := −6 g mv := 70⋅ 10 2 Cpv = 1.03 × 10 fvxo = 17.321 ∆Vea ⋅ Vout⋅ ( 2π ) ⋅ Cpv⋅ Cpfc Rzv := Czv := 1 2π fvxo ⋅ Cpv 10 2π fvxo ⋅ Rzv 4 Rzv = 8.923× 10 −6 Czv = 1.03 × 10 Vary standard values and parameters to check fvxo and Phase Margin (PM) under different conditions. Typically: 0.00 < kPin < 1.0, 0.8 < kCout < 1.2, 0.8 < kgmv < 1.2 128 −7 Calculated values: Cpv = 1.03 × 10 Select standard values: Cpv := 0.05⋅ 10 −6 Pin1 GVL := ⋅H ⋅ i ∆Vea ⋅ Cpfc⋅ Vout⋅ s 1 i Czv = 1.03 × 10 −6 3 Rzv := 91⋅ 10 Czv := 0.5⋅ 10 (01.0⋅ gmv)⋅(s i⋅ Rzv⋅ Czv + 1) s i⋅ Rzv⋅ Czv⋅ Cpv s ⋅ ( Czv + Cpv ) ⋅ Czv + Cpv i dBGVL := 20⋅ log GVL i i −6 4 Rzv = 8.923 × 10 + 1 φG VL := atan2 Re GVL , Im GVL ⋅ i i 180 i π PM_G VL := φG VL + 180 i i Voltage Open Loop Bode Plot 100 90 80 60 36 40 − 18 φG VL 20 dBG VL i i 0 PM_G VL − 20 − 72 i − 40 − 60 − 126 − 80 − 100 0.1 1 10 100 − 180 3 1×10 fi 129 Part 3 - Load Step Change Transient Response Load Step Change Transient Response from 5% to 95% of Full Load Top: Output Voltage Bottom: Input Current Load Step Change Transient Response from 95% to 5% of Full Load Top: Output Voltage Bottom: Input Current 130
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Investigation of high performance single-phase solutions for AC-DC power factor corrected boost converters Musavi, Fariborz 2011
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Title | Investigation of high performance single-phase solutions for AC-DC power factor corrected boost converters |
Creator |
Musavi, Fariborz |
Publisher | University of British Columbia |
Date Issued | 2011 |
Description | Plug-in Hybrid Electric Vehicles (PHEVs) and Electric Vehicles (EVs) are an emerging trend in automotive circles, and consumer’s interest is growing rapidly. With the development of PHEVs, battery chargers for automotive applications are becoming a large market for the power supply industries. The improvement of overall charger efficiency is critical for the emergence and acceptance of these vehicular technologies, as the charger efficiency increases, the charge time and utility cost decreases. Additionally, to meet the efficiency and power factor requirements and regulatory standards for the AC supply mains, power factor correction is essential. Due to limited space in vehicle and increasing power consumption, chargers are required to deliver more power with smaller volume. As a key component of a charger system, the frontend AC-DC converter must achieve high efficiency and high power density. In this dissertation, several conventional plug in hybrid electric vehicle charger front end AC-DC converter topologies are investigated and a new bridgeless interleaved and a phase shifted semi-bridgeless power factor corrected converter are proposed to improve the efficiency and performance, which is critical to minimize the charger size, charging time, and the amount and cost of electricity drawn from the utility. A detailed analytical model for these topologies is developed, enabling the calculation of power losses and efficiency. Experimental and simulation results of several prototype boost converter converting universal AC input voltage to 400 V DC at 3.4 kW are given to verify the proof of concept, and analytical work reported in this thesis. The results show a power factor greater than 0.99 from 750 W to 3.4 kW, THD less than 5% from half load to full load and a peak efficiency of 98.94% at 265 V input and 1200 W load. |
Genre |
Thesis/Dissertation |
Type |
Text |
Language | eng |
Date Available | 2011-06-02 |
Provider | Vancouver : University of British Columbia Library |
Rights | Attribution-NonCommercial-NoDerivatives 4.0 International |
DOI | 10.14288/1.0071876 |
URI | http://hdl.handle.net/2429/35082 |
Degree |
Doctor of Philosophy - PhD |
Program |
Electrical and Computer Engineering |
Affiliation |
Applied Science, Faculty of Electrical and Computer Engineering, Department of |
Degree Grantor | University of British Columbia |
GraduationDate | 2011-11 |
Campus |
UBCV |
Scholarly Level | Graduate |
Rights URI | http://creativecommons.org/licenses/by-nc-nd/4.0/ |
AggregatedSourceRepository | DSpace |
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